The Resonant Commutated Twin Pole Inverter

F. Hinrichsen* and W.-R. Canders** * Converteam GmbH, Culemeyerstr. 1, Berlin, Germany, frank.hinrichsen@converteam.com

** Technical University Braunschweig, Institute for Electrical Machines, Traction and Drives, Hans-Sommer-Str. 66, Braunschweig, Germany, w.canders@tu-bs.de

Abstract - The Resonant Commutated Twin Pole Inverter

(RCTPI), a novel resonant pole inverter topology using only

main switches, is introduced. One phase of a voltage source

RCTPI consists of four switches arranged in two symmetric

half-bridges called the twin poles. Each pole carries half of

the phase current. The switches are turned off as zero

voltage switch (ZVS) and zero current switch (ZCS)

respectively and turned on under ZCS both. Switching

losses in the semiconductor devices can be reduced in all

operating points by soft switching, while overall losses are

reduced only at high load currents. The concept can be

realised using IGBTs in standard packages. Single switch

modules, half- and full-bridge modules and some six-packs

are suitable - this is an advantage compared to many other

resonant topologies.

I. INTRODUCTION

Many three phase resonant inverter topologies have been presented during the last two decades. While in the beginning, resonant link topologies were favoured, a trend towards resonant transition topologies can be observed in the last years. A comprehensive overview is given for example in [1].

Resonant inverters make use of ZCS and/or ZVS to reduce switching losses compared to their hard-switched counterparts. Reducing the current and voltage stress for the power devices was and is still an important objective, but modem semiconductors are fast and robust enough to be employed in most three phase applications without any kind of snubber. Indeed it remains a strong motivation to construct resonant inverters to increase the switching frequency or to save energy. Today, practical inverter concepts have to fulfil most of the following requirements be competitive with hard-switched inverters: modularity, scalability, PWM control, compatibility to common DC­link/busbar concepts, simple control without additional sensors, use of standard driver stages and switches in standard packages.

One topology that features all the above mentioned properties is the Auxiliary Resonant Commutated Pole (ARCP) Converter proposed in [2, 3]. Practical realisations are reported in [4, 5, 6]. Figure 1 shows its commutation cell. The main switches Sib SlI function as Zero Voltage Switch. The auxiliary switches SaU, Sall can initiate a resonance cycle to commutate the load current from a conducting diode to a switch and support the capacitive commutation from a conducting switch to a diode, if the load current is not sufficiently high. The only drawback is the poor utilisation of the auxiliary switches which are only used during commutation intervals for a few microseconds per switching period. On the other

hand, other concepts that work without (active) auxiliary switches like [7, 8, 9] suffer from poor controllability.

Figure 1. Network N1: ARCP commutation cell

Figure 2. Graphical method to perform the duality transformation Nl --> N2

Lr22

Figure 3. Duality transformed ARCP cell N2

The aim of this work is to find a converter topology that avoids these drawbacks and is nevertheless as simple to realise as the ARCP.

II. DERIVATION OF THE RESONANT COMMUTATED

TWIN POLE INVERTER

Starting from the ARCP commutation cell considered as network N I (Figure 1), the novel topology can be derived using the rules of duality theory [10] which are valid for planar networks. The transformation can be done mathematically, as a matrix transformation, or via a graphical method as shown in Figure 2. Each network element is replaced by its dual element (e.g. capacitors by inductors, voltage sources by current sources etc.) and each node turns into a mesh, e.g. CD ---+ I.

978-1-4244-1668-4/08/$25.00 ©2008 IEEE 1414

1,12

Figure 4. Commutation cell of the current source RCTPI

Figure 5. Commutation cell of the voltage source RCTPI

The result is depicted in Figure 3: all fonnerly reverse­conducting switches have turned into reverse-blocking ones. Therefore all control signals have to be inverted. A resonant commutation for example is now initiated by blocking one of the auxiliary switches, allowing the capacitor to be charged with one specific polarity. This capacitor takes the role of the inductor in the original ARCP cell. Consequently the main switches operate as ZCS and the auxiliary switches as ZVS.

To avoid that half of the current has to pass two semiconductor devices in series, which would lead to undesirable voltage drop and losses, the modified structure in Figure 4 is proposed: here the fonner auxiliary switches S2, S3 bypass the main switches S], S4. The load was split and the current sources were tied together to fonn a single current path as it is common in current source inverters.

In Figure 5 this circuit was redrawn to get a more familiar kind of depiction. In this case it is equipped with reverse conducting devices to fonn a commutation cell for voltage source inverters. In steady state, always both upper switches S], S2 or both lower switches S3, S4 share the load current. Capacitor Crl is shorted. The most obvious change referred to Figure 1-3 is the absence of any auxiliary circuitry, leading to a type of resonant inverter using only main switches. This novel commutation cell will be further on referred to as a Resonant Commutated Twin Pole (RCTP), because one inverter pole consists of two equal half-poles.

III. FUNCTIONALITY

The commutations in the RCTP cell can be divided in different modes that are described in the following sections.

A. Commutation/rom Reverse Conducting Switches to Forward Conducting Switches (Mode 1)

For example S], S2 are carrying a negative load current h < O. The switches S3, S4 can be turned on instantly. A simple inductive commutation takes place. All devices switch under ZCS-condition. The duration of this commutation is calculated as follows:

(1)

A supported commutation for small load currents, as known from the ARCPI, is not necessary, because the commutation time gets even smaller with decreasing load current, as can be seen in (1).

B. Commutation/rom Forward Conducting Switches to Reverse Conducting Switches (Mode 3)

Again h should be negative. To switch off S3 and S4, the strategy depicted in Figure 6 is followed. It starts with the ZVS turn-off of S3. As a result, S4 takes over the whole load current h at to. In the following boost phase (to-tl) the resonant capacitor is pre-charged.

Figure 6. Resonant commutation scheme for Mode 3a

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The difference between Vd and VCr is the boost voltage

(Ia)

This boost method is dual to the one described in [11] for the PWM current stiff converter with resonant snubbers (current source ARCPC). After the boost phase, SI is switched on, leading to a resonance between Lrb Lr2 and Cr!. After the zero-crossing of iS4, switch S4 is turned off as ZCS. When VCr! reaches zero again at t4, switch S2 takes over half of the current of S3. Assuming ideal switches and lossless passive components, the waveforms of voltage and currents of the resonant elements between tl and t3 are given by the following equations:

VCrJ(t) = Vd +Zr' I; ·sin(lVr(t-tJ))­

-Vb 'COS(lVr(t-tJ)),

iLr(t)=-I; . COS(lVr(t-tJ)) + �b . sin(lVr(t-tJ)), r

iLr2 (t) = iLrJ (t) + I L'

(2)

(3)

(4)

The smaller the load current, the higher is the necessary boost voltage Vb' At very small load currents the fall time of VCr during t3 and t4 and therefore the total commutation time gets unacceptable long. One possibility to avoid this is to turn on SI fIrst, leading to a current boost in Lr! and Lr2 through SI and S3 that is stopped by turning off S3 (Mode 3b, [12]). Unfortunately, this method leads to disproportional losses with respect to the small load current. Hence hard turn-off of small load currents should be preferred.

IV. DESIGN CONSIDER ATIONS

In contrast to the ARCP inverter, where the auxiliary switches can be thyristors, for the RCTP inverter, four fully controllable reverse conducting switches per phase are needed for example IGBTs with anti-parallel diodes. Because of the H-bridge structure of the RCTP cell, single, half-bridge or full-bridge IGBT modules can be engaged. A three-phase inverter can even be realised with two six-pack modules that feature open collector and open emitter. Only the connections between the switches of each phase and the connections to the resonant capacitor have to be of low inductance. That leads to a high degree of freedom for the designer.

(a) u, V, w, u, v, w, (b)

Figure 7. Alternative concepts for the current divider: (a) special sine wave filter. (b) motor with separate coils

The current sources at the output can be realised by two coupled inductors per phase as can be seen in Figure 11. If a sine wave fIlter is implemented anyway, the inductor of the fIlter can be split. Alternatively special machines with divided stator windings might be used to balance the currents (Figure 7).

For the test inverter, discrete IGBTs with and without integrated freewheeling diodes were chosen because the power of the test setup is rather low and because it was an aim to test the new topology also with SiC diodes, which were not available in modules at that time.

A. Choice of the Components

Previous to the construction of the test converter, different IGBT technologies (NPT, Short-Tail NPT and Trench Fieldstop) were tested in a simple half-bridge arrangement with capacitors Cr and inductors Lr of different values to choose the right combination for minimum losses and short commutation times. Table I sunmIarises the tested semiconductors and their typical values.

TABLE r. DATA SHEET VALUES OF THE EXAMINED 1200-V IGBTs

Technology Type

IGBT Standard NPT BUP314D

Short-Tail NPT lXER35N120DI

Trench Fieldstop IKW40T120 IGW40T120

Diode

FR ED BUP314D

HiPerFR ED lXER35N120DI

EmCon IKW40T120

SiC Schottky 2 x CSD20120

T, =90 DC

Ic,n 33 A 32 A

40 A'1

IF,n 28 A

25 A

40 A'1

2 x 20 A'2 * I, T, JOO C *2: T, � J60 'C

1j = 125 DC

UCE (30 A) 3.53 V

2.31 V

1.92 V'3

VF(30A) 1.92 V

1.67 V

1.58 V

2.56 V *3 T. J50 'C

As can be seen in Figure 8, at capacitive turn-off, the Trench Fieldstop IGBT has the best performance. The gate resistor has no influence on the switch-off behaviour. For the inductive commutations sunmIarised in Figure 9, the gate resistor should be as small as possible. The standard NPT-IGBT shows the lowest turn-on losses but because of its high conduction losses it is not the best choice. In addition, the chip technology is of minor importance for high inductance values.

While the modem IGBT types have the lower overall losses, the new diode types HiPerFRED and EmCon �ead to much higher switching losses (Figure lO), especIally with increasing commutation inductance. The reason is that these chips were optimised for soft recovery in low inductive circuits.

Because of the outstanding low conduction losses the Trench Fieldstop IGBTs were chosen. To check how much lower the losses can be with a better diode, silicon carbide Schottky diodes without any reverse recovery charge were tested additionally. All components and values of the test inverter are listed in Table II.

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t 4.0 mJ+ ··················································· .................. .

Ev 3.0 2.5 2.0 1.5 1.0 0.5

NPT, RG = 27 n ShortTall NPT, RG = 15 n Short Tail NPT, RG = 27 n

.....- Trench Fleldstop, RG = 10 n

0.0 +---�--�---�--_I_-----< 20

40 60 80 nF 100

Cr-

Figure 8, Tum-off losses as a function of parallel capacitance (Ie = 30 A, Ud = 600 V, T, = 25 DC)

t 4.0 mJ .f"II. ............................................. I ··········· NPT, RG = 10 n

E v 3.0 -A- Short Tail NPT, Ro = 15 n Short Tail NPT, Ro = 27 (l 2.5 -.i!- Trench Fieldstop, RCj = 10 n

2.0 +-----'��--+---'------'---+------l 1.5 l ··········· \ ... "" '''"I·····················j ··········· j..................... +.......... I 1.0 + ...................... '" ' 0.5+-----0.0 +--�--_I_-

0.0 0.5 1.0 1.5 2.0 �H Lr_

3.0

Figure 9, Tum-on losses as a function of connnutation inductance (Ie = 30 A, Ud = 600 V, T, = 25 DC)

t 30

mJ Ev 20

1.5 1 0 0.6

0.0

-4- FR�D �HiPerFRED

--- Em

V � r- 0-'� ..-

0.0 0.5 1.0 1.5 2.0 �H Lr ___

3.0

Figure 10, Diode tum-off losses as a function of connnutation inductance (IF = 30 A, Ud = 600 V, T, = 25 DC)

TABLE II, COMPONENTS OF THE TEST SETUP

Part/Dimension Values Type

IGBTs with EmCon 40 Al1200 V lnfineon IKW40T120 diode

IGBTs without diode 40 Al1200 V lnfineon IGW40T120

SiC Schottky diodes 20 Al1200 V CREE CSD20120

Resonance capacitance C, =451lF WIMAFKPI

Resonance inductance L,=2.4IlH Ferrite core: EDT 44, material: N87

Coupled inductor Coil resistance: Ferrite core: EC70, 2 x 4,6 mQ material: N27

B. Control

Compared to a standard hard-switched inverter an additional instance is needed to control the resonance cycle. This has been realised by one small programmable logic device per phase (CPLD with 64 macrocells). Figure 11 shows a block diagram of the control. The phase control logic gets the PWM signals normally sent directly to the driver stages of the upper and the lower switch of a phase-leg. Only the dead time was extended to about 6.25 f.!S. Depending on the height and polarity of the measured phase current, the logic decides whether Mode 1 or Mode 3 should be used for the demanded

commutation. With load currents between -8 A and +8 A the IGBTs are switched hard. Both modes work with a fixed timing scheme, load current depending switching times as known from the ARCP inverter [ l3] are not needed.

All standard driver stages can be used because no additional sensing or feedback is necessary. As a drawback, active clamping is needed to manage the overvoltage spikes appearing at diode turn-off in Mode 1. The driver stage of the test inverter is based on an integrated gate driver chip (HCPL-316J). An active clamping level of Vel = 950 V is realised by three series connected suppressor diodes.

Figure 11, Block diagram of the control of one phase (Module V)

300 1200

t A vt 200 800

150 600

100 400

50 200

0 0

-50 -200 a 2 4 6 8 �s 10

t __

t __

Figure 12, Short-circuit test with IKW40T120, Ld,2 = 2,IIlH, Ro = 10 Q, Vel = 950 V

The benefit of the active clamping is that even in case of a short circuit in the bridge, which is detected by the driver chip, the IGBTs can be switched off safely (Figure 12).

C. Power Section

Figure l3 shows the schematic of the test inverter: phase module U is hard-switched, first with EmCon diodes, later on with SiC Schottky diodes. Phase Modules V and W are soft-switched (Module W features silicone carbide Schottky diodes). Figure 14 shows a photo of the test setup. Each phase module has its own

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heat sink of the same type (Figure 15) so that thennal measurements of the semiconductor losses of the different realisations are possible. For that purpose, NTC resistors were embedded in the heat sinks. The IGBTs with integrated freewheeling diodes or external silicone carbide Schottky diodes (both in T0247 package) were mounted with screws and isolating, thennal conductive pads.

"1= alternative: * Figure 13. Schematic of the test inverter

Figure 14. Experimental setup (Module W unfinished)

Figure 15. Mounting of the switches on the heat sink (Module V)

V. EXPERIMENTAL RESULTS

Experiments have been carried out to proof the functionality of the new topology. The inverter worked fine under any load condition (also open circuit) without any destruction of semiconductors in the soft-switched modules.

A. Switching Behaviour

The switching wavefonns (Figure 16 to Figure 21) are measured under the maximum peak load current of h =

-48 A and a DC-link voltage of Vd = 513 V in Module V, IGBT S8. These measurements validate the theoretical wavefonns given in Figure 6. The oscillations in the current wavefonns in Figure 17 and Figure 19 are caused by parasitic inductances. The losses in Figure 18 are caused by the current tail of the blocking IGBT. Mode 3 wavefonns in Module W look nearly identical.

50 1000 t A vt veE 30 600 20 400 10 200 0 0 -10 -200 0 2 4 6 8 �s 10

t--

,1,�1114+H+ III o 2 4 6 8 �s 10 1_

1'lgUH: 10. L V � U11 alto �lVIUUt: .), lVlUUUlt;: V)

200 o +--+--1--+-++--1--1+-+-+--+--+ -200

-600 -f---I--'-'+--+ -800

+--+--1--+--1--1--1-+-+--+--+-1000 2 4 6 8 �s 10 t __

Figure 17. ZCS on at tJ (Mode 3. Module V)

�+-+--+--I--+--+-I- 100�t 600 400 200 o

+--+--1--+--+-1--1-+-+--+--+ -200

2 4 6

1_

8 �s 10 t--

Figure 18. Near ZCS off att3 (Mode 3. Module V)

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(� 20 10

o

-10 -20 -30 -40 -50

·· ···c·

o

-VF

ift � WI

2 4

-

IVV WV

6

(V 1\-iF

8 �s 10 t_

80�t 400 200 o

-200 -400 -600 -800

-1000

Figure 19. ZVS on at t4 (Mode 3, Module V)

( 2

0 A VeE A

I a I a

0 0 2 4

I . /e

j\ � ir\

6 8 �s 10 t_

8°�i 400 200

o

-200

Figure 20. Turn-on of Sg (Mode 1, Module V)

(� 40 30 20 10

o -10 -20 -30

15

tkW Porfo 5

o

-5

o

o

-/

2 4

J

2 4

-V;:

.• ···c .. c·· t .. .. / ! ... . ....

6 8 �s 10 t--

I I I Eoff,D = 2.00 mJ

111 I I I 6 8 �s 10

t_

120�i 800 600 400 200

o

-200 -400 -600

Figure 21. Turn-off of Dg (Mode 1, Module V)

In Mode 1 the turn-on is nearly lossless (Figure 20) while at diode turn-off the above mentioned overvoltage spikes appear. In case of the silicon diode (Module V), the overvoltage is clamping by the parallel IGBT (Figure 21).

B. Power Losses

The inverter was loaded with a squirrel-cage induction machine (380 V, 43 A, 22 kW, 1460 min-

I) coupled to a

DC generator brake. Figure 22 shows the load current iw and voltage VW-N at full inverter load (measured with an RC-lowpass filter between phase output terminal W and the negative pole N of the DC-link).

The semiconductor power losses were derived from measurements of heat sink temperature T hs and ambient temperature T amb according to (5).

+__-1-,,-----+ '=»FF�-I---+--+ =iFF1!!;-f--f----:!- 6D� t +---+--����+--�,+__-+__7��-�������200

t __

Figure 22. Phase current and output voltage of phase W (measured with filter/g = 2.1 kHz) of module W atfs = 10.6 kHz

f: PI,semi

hard Si soft

160 _SiC hard SiC soft

120+--�-------��-�-�'=--�

80+---+---��=�-+--����===�

40 +---+---3-���-�--+----�

10 15 20 25 A 35 'L -

Figure 23. Semiconductor losses per module atfs = 15.9 kHz

t240

PI.� 160

120

80

40

0

97 96 95 94 93 92

-Si hard Si soft

_SiC hard SiC sal!

5 10 15 20 25 A I 35

Figure 24. Total losses per module atfs = 15.9 kHz

o

r ",·"··""·

/.'C'" x

r- ".".' Si hard Si soft if SiC hard / c' SiC soft il. .

5 10 15 20

1 I

"", llc' icc .... "

,-

25 A 35 IL-

Figure 25. Efficiency of module atfs = 15.9 kHz

p. _Ths-Tamb l,semi - R , 'th.hs-amb (5)

The thermal resistance Rth,hs-amb of the heatsinks was determined before in an experiment: the heatsink of Module V was warmed up by a DC current floating from N2 to P2 through the freewheeling diodes of all four switches. The current, the voltage drop across the diodes and the temperature difference were measured to calculate Rth,hs-amb' Figure 23 shows the results for different operating points.

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38:JV'�

Figure 26. Schematic of the test setup for measurements with the power analyser

For each measurement, the motor was reved up to its nominal speed by the inverter and then loaded step by step with increasing torque. Between two duty points the heat sink temperatures had enough time to settle. The semiconductor losses in the soft switching modules were always lower or equal than in the hard-switched modules, but obviously only at higher load currents a reasonable save in power loss can be achieved.

The total power losses in Figure 24 were measured by wide band power analyser (NORMA D6200) according to Figure 26. At about 25 A the power analyser changed over its measuring range, thus the measured values left and right of this point differ more from each other than the real values did.

Figure 25 shows the efficiencies of the different modules at a switching frequency ofls = 15.9 kHz. The efficiencies of both modules with SiC Schottky Diodes are nearly equal at high load, because the small reduction of switching losses is about the same as the additional losses in the resonant components and the coupled inductor.

VI. CONCLUSION

In this paper the RCTP commutation cell is derived from the commutation cell of the well known Auxiliary Resonant Commutated Pole Inverter (ARCPI) via duality transformation and modification of the dual topology. The novel ZCS/ZVS commutation cell features only main switches. It is suitable to VSIs and CSIs. The simple full­bridge structure of the phases makes the RCTP VSI easy to be implemented with standard IGBT modules.

Function and control strategy of the novel inverter are discussed. A three phase test inverter was built. Measurements of current and voltage waveforms during commutation intervals as well as power loss measurements are presented. Hard- and soft-switching losses were directly compared while the inverter was loaded with an induction motor.

The experiments have confmned the functionality of the new inverter concept and the switching loss reduction in the semiconductors by soft switching as well as SiC Schottky diodes. Because the soft switching topology generates some additional losses in the auxiliary elements,

the measured overall losses in the resonant topology with SiC diodes are not smaller than in a hard switching inverter with SiC diodes. If SiC Schottky diodes are not available, for example for high power inverters, the RCTP topology is advantageous, especially at full load.

REFERENCES

[1] M. D. Bellar. T-S. Wu. A Tchamdjou. I Mahdavi. M. Ehsani. A Review of Soft-Switched DC-AC Converters. IEEE Transactions on Industry Applications. Vol. 34. No. 4. July/Aug. 1998. pp. 847-860

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[4] F.-F. Protiwa. I KoB. A 20 kVA Auxiliary Resonant Commutated Pole Converter - Design and Practical Experiences -. Proceedings of the 6th European Conference on Power Electronics and Applications (EPE '95), Sevilla, Spain, 19-21 Sept 1995, pp. 2.111-2.116

[5] H.-I Pfisterer, Der Auxiliary-Resonant-Commutated-Pole-Strom­richter, ein Resonanzstromrichter mit Spannungszwischenkreis, am Niederspannungsnetz, Dissertation, Universitat Karlsruhe, Fortschritt-Berichte VDI, Reihe 21, Nr. 309, VDI Verlag, Dusseldorf, 2001

[6] F. Himichsen, G. Tareilus, W-R. Canders, 1 MVA-ARCPI with High Voltage IGBT-Modules - Design and Practical Experience, Proceedings of the 10th European Conference on Power Electronics and Applications (EPE '03), Toulouse, France, 2-4 Sept 2003

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[8] A Cheriti, K. AI-Haddad, 1. A Dessaint, T A Meynard, D. Mukhedkar, A rugged soft commutated PWM inverter for AC drives, IEEE Transactions on Power Electronics, Vol. 7, No. 2, April 1992, pp. 385-392

[9] X. Zhang, Y. Zou, I Zhang, I Hu, Investigation of AUXiliary Diode Resonant Pole Inverter, Conference Record of the 4th IEEE International Conference on Power Electronics and Drive Systems (PEDS), 22-25 Oct 2001, Denpasar, Bali, Indonesia, Vol. 2, pp. 643-646

[10] F. lenni, D. Wuest, Steuerverfahren fur selbstgefuhrte Strom­richter, vdf, ZUrich; Teubner, Stuttgart, 1995, pp. 70-71

[11] B. I Cardoso Filho, T A Lipo, A Reduced Parts Count Realization of the Resonant Snubbers for High Power Current Stiff Converters, Conference Proceedings of the 13th Annual Applied Power Electronics Conference and Exposition 1998 (APEC '98), Anaheim, California, USA, 15-19 Feb. 1998, Vol. 2, pp. 558-564

[12] F. Himichsen, Untersuchungen zu Resonant Commutated Pole Kommutierungszellen in Spannung- und Stromzwischenkreis­Umrichtern, Dissertation, Technische Universitat Braunschweig, 2008, Cuvillier-Verlag, Gottingen, 2008

[13] G. Tareilus, F. Himichsen, W.-R. Canders, Simple Design and Control of ARCP-Inverter for Universal Power Range, Proceedings of the Symposium on Power Electronics, Electrical Drives, Automation & Motion (SPEEDAM), Ravello, Italy, 11-14 June 2002, pp. A2-7-A2-11

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