High Linearity and High Efficiency Power Amplifiers in GaN HEMT Technology

Shouxuan Xie

Department of Electrical and Computer Engineering,

University of California, Santa Barbara June 30, 2003

Outline

1. Introduction and motivation

- Why GaN HEMTs

- Objectives of the GaN HEMTs PA design

2. Class B for high efficiency and high linearity

- Why single-ended Class B

- Circuit design and measurement result

3. Identify and model nonlinear sources of GaN HEMTs

- Nonlinear gm

- Nonlinear Cgs

- Nonlinear Gds

4. Proposed new designs to further improve linearity

- Common drain Class B (to improve gm nonlinearity)

- Pre-linearization diode (to improve Cgs nonlinearity)

5. Problems and future works

• Advantages of GaN

- High breakdown field: 3 MV/cm

- High Vsat @ 2.5 x 107 cm/s

- Thermal conductivity: 3x GaAs

- Large channel charge: > 1x1013 cm-2

- Good electron mobility: >1200 cm2/V-s

• Advantages of GaN HEMTs

- High power density: 12W/mm for X-band (8-12GHz)

- High Ft (50GHz) and fmax (80GHz) for 0.25um device

- Linear I-V characteristics

Why GaN HEMTs

Standard AlGaN/GaN HEMT structure

SiC substrate ~400 um

1.4 um GaN buffer

25 nm Al0.3Ga0.7N barrier

60nm AlN Nucleation layer

1nm AlN layer

400 nm Silicon Nitride

Plated Airbridge

Silicon Nitride Passivation Layer

Ti/ Al/ Ni/ Au ohmic Contact

Ni/Au Schottky Contact

GaN HEMT process and device structure

0.25um T-gate for 50GHz ft

SiC substrate for high heat conducting

MIM capacitors

SiN passivation for High RF output power

Air-bridge for ground connection of CPW

Idss = 1 A/mm @Vgs=0V

0

5

10

15

20

25

30

35

40

1 10 100

h 21 (d

B)

Frequency (GHz)

fT = 50GHz

I-V Curve for 600m SG device

• Lg ~ 0.25um,

• Idss ~ 1A/mm

• ft ~ 55GHz (50GHz for DG)

• Vbr ~ 40V (55V for DG)

RF Performance 150m DG device

Device performance

0

200

400

600

800

1000

-5 -4 -3 -2 -1 0 1

Dra

in C

urre

nt, m

A

Gate Bias,V

Linear Id-Vgs characteristic on SiC

Device performance summary:

Objectives of GaN HEMT PA design

Design RF MMIC power amplifier in GaN HEMT technology to achieve:

1. High linearity (low IMD3 distortion)

2. High efficiency

3. High output power

4. Broad bandwidth

(High linearity and high efficiency are primarily concerned here)

Class A: Very high linearity and wide bandwidth; but very low efficiency (Ideal PAE 50%, feasible PAE 20-30%).

Switch mode Amplifiers (Class D, E): Very high efficiency (Ideal PAE 100%, feasible PAE 60-70 %); but poor linearity and poor bandwidth.

Class B: Good efficiency (Ideal PAE 78.6%; feasible PAE 40-50% ) and good bandwidth, and potentially low distortion.

RL

VG

Vin

Cbias

Cbias

VD

1:1

1:1

Vin-

Vin+

ID+

ID-

Vout

VDS+

VDS-

• Even harmonics are suppressed by symmetry => wide bandwidth • Half-sinusoidal current is needed at each drain. This requires an even-

harmonic short. It can be achieved at HF/VHF frequencies with transformers or bandpass filters. However,

1. Most wideband microwave baluns can not provide effective short for even-mode. Efficiency is then poor.

2. They occupy a lot of expensive die area on MMIC.

Push-pull Class B

ID1

Vin

-Vin

Vin

+-ID2

= ID

33

2211 inininD VaVaVaI

3

32

221

21

2)(2 ininin

DDD

VaVaaVa

III

33

2212 )( ininininD VaVaVaVI

][ 33

221 inininLLDout VaVaVaRRIVZero Z at 2f0

RLvi

band passfilter @ fo

voutID

Push-pull Class B

Single-ended Class B with bandpass filterEven harmonics suppressed by symmetry

Conclusion: From linearity point of view, push-pull and single-ended Class B with bandpass filter B are equivalent – same transfer function.

Even harmonics suppressed by filter

Single-ended = push-pull

Bandwidth restriction < 2:1

0

180

0

180

+Vin

-Vin ID2

ID1

Class B bias for high linearity

Ideal Class B Bias too low: Class C Bias too high: Class AB

ID1

Vin

Vin

Vin

+

ID2

= ID

Vp

ID1

Vin

Vin

Vin

+ID2

= ID

Vp

ID1

Vin

Vin

Vin

+ID2

= ID

Vp

RLTLIN

R1 L1

L2

C1

RF IN

Vg

Vd

BIASTEE Input

matchingnetwork

Outputmatchingnetwork

Cds BIASTEE

(short at 2fo, 3fo...)

Gate 2

Gate 1

Lossy input matching - section lowpass filter

Single-ended Class B Power Amplifier

• Dual gate device is used since it has higher Vbr, higher MSG (smaller S12)

and higher output resistance Rds

• Lossy input matching network to widen the bandwidth

• Cds is absorbed into output matching network (Low pass filter)

Signalgenerator_1

Signalgenerator_2

Poweramplifier_1

Poweramplifier_2

Powercombiner Coupler

Powermeter

Bias T DUT

Coupler

Bias T

- 20 dB - 20 dB

Coupler 50 OhmLoad

Spectrum Analyzer

CH_A

CH_B

- 20 dB- 20 dB - 20 dB

• Single tone from 4 GHz to 12 GHz;

• Two-tone measurement at f1 = 8 GHz, f2 = 8.001 GHz;

• Bias sweep: Class A (Vgs = -3.1V), Class B (Vgs = -5.1V), Class C (Vgs = - 5.5 V) and AB (Vgs = -4.5 V).

Measurements:

Measurement setup

Gain and bandwidth

-5

0

5

10

15

20

2 4 6 8 10 12 14 16

Gain

(dB

)

Frequency (GHz)

Class AB

Class B

3 dB bandwidth for Class B: 7GHz - 10GHz

Class B PA measurement results

Class B bias @Vgs = - 5.1V

Single tone performance @ f0 = 8GHz:

Two tone performance @ f1=8GHz, f2=8.001GHz :

15

20

25

30

35

40

0

0.1

0.2

0.3

0.4

0.5

0 5 10 15 20 25 30

Ou

tpu

t po

wer,

db

m

PA

E

Input power, dbm

PAE (saturated) ~ 34%

Saturated output power 36 dBm

Good IM3 performance: • 40dBc at Pin = 15 dBm• > 35 dBc for Pin < 17.5 dBm-50

-40

-30

-20

-10

0

10

20

30

0

0.05

0.1

0.15

0.2

0.25

-15 -10 -5 0 5 10 15 20

Outp

ut

pow

er,

dB

m

PA

E

Input power, dBm

f1,f2

2f1-f2, 2f2-f1

Two tone performance @ f1=8GHz, and f2=8.001GHz :

Good IM3 performance at low power level but becomes bad rapidly at high power levels

-10

0

10

20

30

40

0

0.05

0.1

0.15

0.2

0.25

0.3

0.35

0.4

-20 -10 0 10 20 30

Out

put

pow

er,

dBm

PA

E

Input power, dBm

-30

-20

-10

0

10

20

30

40

0

0.05

0.1

0.15

0.2

0.25

0.3

0.35

-20 -15 -10 -5 0 5 10 15 20

Outp

ut

pow

er,

dB

m

PA

E

Input power, dBm

f1,f2

2f1-f2, 2f2-f1

Class A bias @Vgs = - 3.1V

Single tone performance @ f0 = 8GHz:

PAE (saturated) ~ 34%

Saturated output power 36 dBm

10

20

30

40

50

60

0 5 10 15 20 25 30 35

IM3 c

om

pre

ssio

n,

dB

c

Pout, dBm

Class BClass A

Class C

Class ABPsat

1. Low output power levels (Pout < 24 dBm), Class A and Class B both exhibit good linearity (Class B > 36 dBc, Class A > 45 dBc).

2. Higher output power levels, Class A behaves almost the same as Class B.3. Class AB and C exhibit more distortion compared to Class A and B.

IM3 suppressions of all Classes

10

20

30

40

50

0

0.05

0.1

0.15

0.2

0.25

0.3

-5 0 5 10 15 20 25 30 35

IM3

supp

resi

on,

dBc P

AE

, twoto

ne

Output power, dBm

Class B

Class A

IM3 suppression and PAE of two-tone

0

0.05

0.1

0.15

0.2

0.25

0.3

0.35

10 15 20 25 30 35 40

PA

E,

sing

le

Output power, dBm

PAE of single tone

Class B

Class A

Class B vs. Class A

Maintaining good IM3 suppression, Class B can get 10% PAE improvement over Class A during low distortion operation.

Nonlinear sources of GaN HEMT

Three major sources have been investigated: 1. Nonlinear gm ( or Ids -Vgs characteristic) 2. Nonlinear Cgs 3. Nonlinear Gds

0

0.05

0.1

0.15

-12 -10 -8 -6 -4 -2 0

gm (

S)

Vgs (V)

1. gm vs. Vgs of 600um SG device

100

200

300

400

500

600

700

-8 -7 -6 -5 -4 -3 -2 -1 0

Cgs

(fF

/mm

)

Vgs (V)

2. Cgs vs. Vgs of SG device

0

0.005

0.01

0.015

0.02

-12 -10 -8 -6 -4 -2 0 2

Gds

(m

ho)

Vgs (V)

3. Gds vs. Vgs and Vds of 600um SG device

Vds=20V

Vds=15V

Vds=10V

Goal: Try to investigate nonlinear sources of the GaN HEMT device and understand how they affect the linearity on circuit

Input MN (linear, Zs)

tVVin 000 cos)(

CgsRL

Leqoutout RIV

CdsGds

gsVgsmout VgI

Nonlinear sources of GaN HEMT

...33

2210

gsgsgs

gsmout

VIVIVII

VgI

Input MN (linear, Zs)

tVVin 000 cos)(

CgsRL

Leqoutout RIV

CdsGds

)( 0gsV

00 3,

Nonlinear gm

0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

-5 -4 -3 -2 -1 0

Experimental (A/mm)Modeled (A/mm)

Dra

in C

urr

en

t I d

s (

A/m

m)

Vgs

(V)

Nonlinear gm

Vp

Dominate at low output power levels

Dominate at high output power levels – more interesting

.........*** 33

2210 gsgsgs VIVIVIIIModeled as:

This term creates IM3 distortion

Nonlinear Cgs

Input MN (linear, Zs)

t

VQ gsi

)()1()0(

CgsRL

tVVin 000 cos)(

Q(Vgs)

33

2210)( gsgsgsgs VqVqVqqVQ

Leqoutout RIV i

CdsGds

)( 0)0( gsV

+ -sZiV *

t

VQZVV gssings

)( )0()1(

Directly effect of Cgs

000 3,2,

000 3,2,

100

200

300

400

500

600

700

-8 -7 -6 -5 -4 -3 -2 -1 0

Experimental (fF/mm)

modeled (fF/mm)

Cg

s (fF

/mm

)

Vgs

(V)

Cgs vs Vgs of GaN HEMTs on SiC

2210

2321 32)( VcVccVqVqq

V

QVC

Therefore even order component of Cgs(Vgs) creates IM3 distortion

Vc Vp

)(10 VcVTanHCCC gsgs

If modeled as:

Nonlinear Cgs

This term creates IM3 distortion

33

2210 VqVqVqqQ

Anti-symmetric about V=Vc

then should be no distortion

direct

Input MN (linear, Zs)

CgsRL

tVVin 000 cos)(

+ - Q(Vgs)sZiV *

*)1( VVV ings Leqoutout RIV )1(i

CdsGds

00 2, t

VQZVV gssings

)( )0()1(

33

2210

)1( )( gsgsgsgs VqVqVqqVQ

Nonlinear Cgs – Indirect effect

Input MN (linear, Zs)

CgsRL

tVVin 000 cos)(

+ - Q(Vgs)sZiV *

*)1( VVV ings Leqoutout RIV )1(i

t

VQZVV gssings

)( )1()2(

00 3,

CdsGds

00 2, t

VQZVV gssings

)( )0()1(

33

2210

)1( )( gsgsgsgs VqVqVqqVQ

3)1(

3

2)1(2

)1(10

)2( )( gsgsgsgs VqVqVqqVQ

Nonlinear Cgs – Indirect effect

)2,( 00 gsV

...33

2210

gsgsgs

gsmout

VIVIVII

VgI

Input MN (linear, Zs)

tVVin 000 cos)(

Cgs

Leqoutout RIV

CdsGds

00 3,

directIndirect

Nonlinear Cgs + nonlinear gm

Input MN (linear, Zs)

tVVin 000 cos)(

CgsRL

dsLLeq RRR //

CdsRds

gsVgsmout VgI

0

0.005

0.01

0.015

0.02

-12 -10 -8 -6 -4 -2 0 2

Gds

(m

ho)

Vgs (V)

Vds=20V

Vds=15V

Vds=10V

Leqoutout RIV

Nonlinear Gds

Gds vs. Vgs of 600um SG device

Nonlinear Gds

DC I-V curve of 600um device on SiC

Vgs = 0 V

Vgs = -7 VVds = 15V

Current through GaN buffer, need more gate voltage to pinch off

Short channel effect

Vgs = -7VVds = 8V

0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

-12 -10 -8 -6 -4 -2 0

Ids

(A)

Vgs (V)

Vds=20V

Vds=15V

Vds=10V

Vp shift

Vp shift due to short channel effect

1.2mm SG device DC I-V curve at different drain bias

Nonlinear Cgs + Vp shift

Vb=Vp=Vc Vb<Vc Vb>Vc Vb>>Vc

Vc Vin

Vin

Vin Vin Vin

Vin Vin

Cgs

-Cgs

Cgs

Cgs

Cgs

Cgs

Cgs

Vin Vin

Vin

Cgs

Cgs

Cgs

Vin

Vin

2C02C0 2C0 2C0

Cgs

Cgs

C0 C0 C C0

C0C0C0C0

0

Vc

Vb

Vc

Vb

Vb

Vb Vb

VbVb

Vc

Vc

Vc

Vc

Vc

Vb

Cgs(Vin)

Cgs(-Vin)

DC

Even order component

vs

Vg vd

VARVAR14

vp12(v)=vp-0.0*(v-8)vp11(v)=vpvp1(v)=if (v<8) then vp11(v) else vp12(v) endif;

EqnVar

CC2C=Cgd pF

VARVAR7

I2(v)=0I4(v)=0.00I3(v)=0.011I1(v1,v2)=gm*(v1-vp1(v2))

EqnVar

RR9R=2 Ohm

I_ProbeI_Probe1

SDD2PSDD2P1

Cport[1]=C[1]=I[2,0]=if ((_v1)>vp1(_v2)) then I1(_v1,_v2)*tanh((_v2)/2) else I2(_v1) endif;I[1,0]=0

VARVAR6Q(v)=cgs*0.6*v+cgs*0.4*ln(cosh(v+5))

EqnVar

SDD1PSDD1P1

Cport[1]=C[1]=I[1,0]=0I[1,1]=Q(_v1)

RR6R=Rin Ohm

CC3C=Cds pF

Paidi’s nonlinear model

Cgs is ideal tanHI-V characteristic currently is linear

Nonlinear Gds currently is modeled by shift in Vp;

Vd

TLIN

R1

C1L2 C2

RF IN

Vg

BIASTEE Input

matchingnetwork

Band-Pass Filter( Short at 2fo,3fo..)

RFC

Further improve linearity

CD circuit schematic

)(1

)(

gsmLoad

gsmLoad

in

outCD VgR

VgR

V

VGain

)( gsmLoadin

outCS VgR

V

VGain

Disadvantage -- Stability problem: Since the MSG is less, the circuit is not unconditionally stable in order to keep reasonable high efficiency. Therefore, extra requirement for the source and load impedance is needed.

1. Common drain Class B to improve gm linearity

RL

RL also functions as series-series feedback resistor, which increase gm linearity.

Linearization factor

10

15

20

25

30

35

40

0

0.1

0.2

0.3

0.4

0.5

5 10 15 20 25 30 35

Ou

tpu

t P

ower

, dB

mP

AE

input power, dbm

Pout

PAE PAE(sat) ~ 38%

Simulation result of CD @5GHz

Pout ~ 38dBmPout and PAE in single tone

0

10

20

30

40

50

60

70

5 10 15 20 25 30 35

IM3 S

up

pre

ssio

n, d

Bc

pout, dBm

10 dB

Common Drain

Common Source: with 37.6dBm Pout and 42% PAE(sat)

Two-tone simulation result of CD vs. CS

Simulation result of CD vs. CS – cont.

12 dB

Simulation result of IM3 suppression at 1W total output power as a function of bias point

0

10

20

30

40

50

60

-8 -7 -6 -5 -4 -3 -2

IM3 s

uppre

ssio

n (

dB

c)

Vgs (V)

Class ABClass C

Class B Class A

Common Source

Common Drain

Common Drain vs. Common Source – cont.

100

200

300

400

500

600

700

800

-8 -7 -6 -5 -4 -3 -2 -1 0

Cgs

(fF

/mm

)

Gate Bias Vgs (V)

Cgs

C_total

C_pd

2. Pre-linearization diode to improve Cgs linearity

Further improve linearity – cont.

Vc

C gd

C gsC gs_pd

Vb1=Vp=-4V

Vb1=2*Vp=-8V

0.25umx100umx12

0.75umx100umx4

Can be very easily implemented on chip and occupy very small area

Gate length can be varied and optimum value can be found since write using E-beam-lithography

Pre-linearization diode

Simulation result of PD

20

30

40

50

60

70

10 15 20 25 30 35 40

IM3 s

uppre

ssio

n (

dB

c)

Output power (dBm)

With PD

Without PD

At least 4dBc improvement in IMD3

IM3 simulation result the designed dual gate CS Class B with pre-linearization diode @10GHz

!! Problem: Short channel effect for 0.25um device !!

• Nonlinear Gds will affect linearity performance directly;• It creates Vp shift, hence generate nonlinear Cgs distortion;• Increases DC bias current, hence decreases PAE; • Decreases breakdown voltage, hence decreases the output power

and also PAE …

Problems and future works

0.75umx100um device on Sapphire 0.25umx100um device on Sapphire

Vgs=0V

Vgs=-10VVds=16V

Vgs=0V

Vgs=-7V

Short channel effect

Currently dual gate device is used: - Nearly no Vp shift - Lower Gds (higher Rds) - Higher maximum stable gain (MSG)

- Number of gates get doubled, hard to yield all- Little bit lower ft, and higher Vknee, hence lower PAE - Not easy to model the nonlinear effect

0

0.1

0.2

0.3

0.4

0.5

0.6

-12 -10 -8 -6 -4 -2 0 2

Ids

(A)

Vgs (V)

0

0.002

0.004

0.006

0.008

0.01

-12 -10 -8 -6 -4 -2 0

Gds

(m

ho)

Vgs (V)

I-V curve of 600um DG device

Vds =15V

Vds =20V

Gds of 600um devices at Vds=20V

Dual gate

Single gate

CD SG Class B @5GHz CS SG Class B @5GHz

CS DG Class B @10GHz CS DG Class B @10GHz with PD

Layouts of the new designed circuits

SiC substrate ~400 um

1.4 um GaN buffer

25 nm Al0.3Ga0.7N barrier

60nm AlN Nucleation layer

1nm AlN layer

400 nm Silicon Nitride

Plated Airbridge

Silicon Nitride Passivation Layer

Ti/ Al/ Ni/ Au ohmic Contact

Ni/Au Schottky Contact

Add Fe doping layer to decrease leakage current through the buffer

New device structures to improve linearity

Improve short channel effect by: - Make the Fe doping layer closer to the channel - Gate recess to increase aspect ratio ??? Question: How about decrease Al% in AlGaN -Increase breakdown and decrease gm?How about P-type doping GaN buffer layer?

??? Other ideas to increase breakdown???

Summary

• Class B bias is good for high linearity and high efficiency;

• Three main nonlinear sources of the GaN HEMT device have been investigated with a new idea of nonlinear model;

• According to simulation, common drain class B can improve linearity by 10dB over CS, and pre-linearization diode can improve linearity by 4dB. Four more circuits are designed and being fabricated to prove them;

• Short channel effect for 0.25um device has been observed. New device structure is proposed to solve the problem and better linearity performance is expected.

1. Fabricate and measure the new designed circuits (CD and PD)

- Need to stabilize the PECVD passivation process

2. Complete the new model to understand all the nonlinear effects

- Add gm nonlinearity

- More accurate model for dual gate device

3. Further improve linearity by new device structures

- Work with Mishra’s group to improve the short channel effect

4. Publish paper and write thesis

5. New ideas on device structure and model to further increase linearity and efficiency

Proposed future works

summer

Fall

Fall

summer

Publications and references

1. Vamsi Paidi, Shouxuan Xie, R. Coffie, U. Mishra, M J W Rodwell, S. Long, “Simulations of High linearity and high efficiency of Class B Power Amplifiers in GaN HEMT Technology.”  Lester Eastman Conference, Aug. 2002

2. Shouxuan Xie, Vamsi Paidi, R. Coffie, S. Keller, S. Heikman, A. Chini, U. Mishra, S. Long, M. Rodwell, “High Linearity Class B Power Amplifiers in GaN HEMT Technology.” Topical Workshop on Power Amplifiers, Sept. 2002

3. Shouxuan Xie, Vamsi Paidi, R. Coffie, S. Keller, S. Heikman, A. Chini, U. Mishra, S. Long, M.J.W. Rodwell, “High linearity of Class B Power Amplifiers in GaN HEMT technology.” Microwave and Wireless Components Letters, to be published

4. Vamsi Paidi, Shouxuan Xie, R. Coffie, B. Moran, S. Heikman, S. Keller, A. Chini, S. P. DenBaars, U. K. Mishra, S. Long and M. J.W. Rodwell, “High Linearity and High Efficiency of Class B Power Amplifiers in GaN HEMT Technology.”  IEEE Transactions on Microwave Theory and Techniques, Vol. 51, No. 2, Feb. 2003

Publications:

Other references:

1. K. Krishnamurthy, R. Vetury, S. Keller, U. Mishra, M. J. W. Rodwell and S. I. Long, “ Broadband GaAs MESFET and GaN HEMT Resistive Feedback Power Amplifiers.” IEEE Journal of Solid State Circuits, Vol. 35, No. 9, Sept. 2000.

2. K. Krishnamurthy, S. Keller, C. Chen, R. Coffie, M. Rodwell, U. K. Mishra, “Dual-gate AlGaN/GaN Modulation-doped Field-effect Transistors with Cut-Off Frequencies ƒT >60 GHz”, IEEE Electron Device Letters, Vol. 21, No. 12, Dec. 2000

3. Solid State Radio Engineering, Herbert L. Krauss, W. Bostian, Frederick H. Raab/ Wiley, John & Sons, Nov. 1980

4. Raab, F.H. Maximum efficiency and output of class-F power amplifiers. IEEE Transactions on Microwave Theory and Techniques, vol.49, (no.6, pt.2), IEEE, June 2001. p.1162-6.

5. Kobayashi, H.; Hinrichs, J.M.; Asbeck, P.M. “Current-mode class-D power amplifiers for high-efficiency RF applications”. IEEE Transactions on Microwave Theory and Techniques, vol.49, (no.12), IEEE, Dec. 2001. p.2480-5.

6. Eastman, L.F.; Green, B.; Smart, J.; Tilak, V.; Chumbes, E.; Hyungtak Kim; Prunty, T.; Weimann, N.; Dimitrov, R.; Ambacher, O.; Schaff, W.J.; Shealy, J.R. Power limits of polarization-induced AlGaN/GaN HEMT's. Proceedings 2000 IEEE/ Cornell Conference on High Performance Devices, Piscataway, NJ, USA: IEEE, 2000. p.242-6. 274 pp..

7. Wu, Y.-F.; Kapolnek, D.; Ibbetson, J.; Zhang, N.-Q.; Parikh, P.; Keller, B.P.; Mishra, U.K. “High Al-content AlGaN/GaN HEMTs on SiC substrates with very high power performance”. International Electron Devices Meeting 1999, Piscataway, NJ, USA: IEEE, 1999. p.925-7. 943 pp.

8. Joseph, J. Teaching design while constructing a 100-watt audio amplifier. Proceedings. Frontiers in Education 1997, 27th Annual Conference (vol.1)Pittsburgh, PA, USA, 5-8 Nov. 1997.) Champaign, IL, USA: Stipes Publishing, 1997. p.170-2 vol.1. 3 vol. xxxvi+1624 pp. 3

9. Shealey, V.; Tilak, V.; Prunty, T.; Smart, J.A.; Green, B.; Eastman, L.F.” An AlGaN/GaN high-electron-mobility transistor with an AlN sub-buffer layer”. Journal of Physics: Condensed Matter, vol.14, (no.13), IOP Publishing, 8 April 2002. p.3499-509.

10. W. R. Curtice and M. Ettenberg, "A nonlinear GaAsFET model for use in the design of output circuits for power amplifiers," IEEE Trans of Microwave Theory Tech, vol. MTT-33, pp. 1383-1394, Dec. 1985.

Publications and references- cont.

Cgs bias dependence

0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

-12 -10 -8 -6 -4 -2 0

Vgs, V

Cg

s, p

F

Does Vc change?

Cgs Vs Vgs (SG600um)

0

0.1

0.2

0.3

0.4

0.5

0.6

-12 -10 -8 -6 -4 -2 0

Vgs, V

Cg

s, p

F

Consider Cgs nonlinearity only simulate IM3 result at 1W output power level: Vp = -5V, Vc = -5V, without PD: 46.3dBc, with PD: 57.4dBcVp = -5.5V, Vc = -5V, without PD: 40.1dBc, with PD: 57.6dBc

Vc

Cgd Vs vgs

0

0.02

0.04

0.06

0.08

0.1

0.12

0.14

0.16

0.18

0.2

-12 -10 -8 -6 -4 -2 0

Vgs, V

Cg

d, p

F

Cds Vs Vgs

0

0.02

0.04

0.06

0.08

0.1

0.12

0.14

0.16

0.18

0.2

-12 -10 -8 -6 -4 -2 0

Vgs

Cds

2 4 6 8 10 12 14 16 18 20 22 240 26

0.20.40.60.81.01.21.4

0.0

1.6

Vds, V

Dra

in C

urre

nt, A

GaN HEMT Model – Vp shift

0 2 4 6 8 10 12 14 160.00

0.02

0.04

0.06

0.08

0.10

0.12

0.14

0.16

Ids(

A)

Vds(V)

Ids200ns Ids80us IdsDC

0 2 4 6 8 10 12 14 16 180.00

0.02

0.04

0.06

0.08

0.10

0.12

0.14

0.16

Ids(

A)

Vds(V)

Ids200ns Ids80us IdsDC

Advantages of GaN

0

1

2

3

4

Ele

ctro

n V

eloc

ity (

107

cm/s

ec) T = 300 K

3.0

Note scale change ( 10 larger)

GaN: ND = 1017 cm-3

GaN: ND = 1019 cm-3

0 0.5 1.51.0 2.0 2.5

Electric Field Strength (105 V/cm)

0.3

Ref: Gelmont et al., J. Applied Physics 74, August 1, 1993

GaAs

InP

InGaAs

Si

Class B two-tone output spectrum

-60

-40

-20

0

20

40

7.998 7.999 8 8.001 8.002 8.003

Outp

ut

spectr

um

, dB

m

Freq, GHz

-60

-40

-20

0

20

40

7.998 7.999 8 8.001 8.002 8.003

Out

put

spec

trum

, dB

m

Freq, GHz

-60

-40

-20

0

20

40

7.998 7.999 8 8.001 8.002 8.003

Out

put

spec

trum

, dB

m

Freq, GHz

-60

-40

-20

0

20

40

7.998 7.999 8 8.001 8.002 8.003

Out

put

spec

trum

, dB

m

Freq, GHz

Pout = 4 dBm

IM3 = 43 dBc

Low input power

Medium input power 1

Medium input power 2 High input power

Pout =18 dBm

IM3 = 39 dBc

Pout = 22 dBm

IM3 = 40 dBc

Pout = 26 dBm

IM3 = 25 dBc

Class A two-tone output spectrum

-60

-40

-20

0

20

40

7.998 7.999 8 8.001 8.002 8.003

Out

put

spec

trum

, dB

m

Freq, GHz

-60

-40

-20

0

20

40

7.998 7.999 8 8.001 8.002 8.003

Out

put

spec

trum

, dB

m

Freq, GHz

-60

-40

-20

0

20

40

7.998 7.999 8 8.001 8.002 8.003

Out

put

spec

trum

, dB

m

Freq, GHz

Pout = 10 dBm

IM3 > 50 dBc

Pout = 27 dBm

IM3 = 31 dBc Pout = 31 dBm

IM3 = 15 dBc

Low input power

Medium input power 2 High input power

-60

-40

-20

0

20

40

7.998 7.999 8 8.001 8.002 8.003

Out

put

spec

trum

, dB

m

Freq, GHz

Medium input power 2

Pout = 23 dBm

IM3 = 42 dBc