DesignCon 2010 Cost Effective Crosstalk Management & Decision Makings3.amazonaws.com/zanran_storage/ · 2012-04-30 · Cost Effective Crosstalk Management & Decision Making Mike White,

DesignCon 2010 Cost Effective Crosstalk Management & Decision Making Mike White, Intel Corporation [email protected] Richard Mellitz, Intel Corporation [email protected]

Dennis Miller, Intel Corporation [email protected] Virapandiane Ragavassamy, Intel Corporation [email protected] Ted Ballou, Intel Corporation [email protected] Michael Brownell, Intel Corporation [email protected]

Abstract This paper guides the board designer in making effective decisions concerning crosstalk management. It describes important coupling mechanisms in board designs, means of predicting crosstalk, and methods for determining routing length/performance impacts. To bring greater clarity to a complicated problem, it introduces the practice of crosstalk management based on a notion of equivalent routing length. The board designer often has limited scope for assessing crosstalk risk; design decisions aimed at layer count reduction, part selection and increasing system density often exacerbate coupling effects. The paper provides several case studies of problematic crosstalk scenarios. Author(s) Biography Mike White is an Analog Engineer in the Platform Validation Engineering Group with 29 years experience in integrated circuit validation. Since joining Intel in 1981 he has been a key member of the validation teams on 80286®, 80486®, P6®, Prescott® and Nehalem® processor families. Mike has two patents and one patent pending related to high speed interconnects. Mike is a member of the IEEE and AAAS. Richard Mellitz is a Principal Engineer in the Server Products Group at Intel. His expertise in platform signaling architecture has facilitated leadership of signal integrity for Intel servers. Since joining Intel in 1998 Richard has been key member of various key processor and I/O bus teams including Itanium®, Pentium® 4, and PCI Express®. In addition he has been a primary contributor for the channel sections of the IEEE802.3ap 10GB Ethernet backplane standard and IPC PCB loss test methods. Richard holds a number of signal integrity patents in time domain reflectometry, clocking architecture, and memory busing as well as delivering many signal integrity papers at PC industry design conferences. Richard wrote the original IPC TDR standard which is used throughout the PWB industry. Dennis Miller is a Senior Signal Integrity Engineer in the Server Products Group at Intel with 30 years experience, has spent the past 16 years at Intel where he has analyzed the signal integrity of interconnect circuitry starting in the era when data at 32MHz was thought to be fast, to now when microwave frequencies are standard. He holds seven patents related to signal transmission, and one relating to power supply design. He is the author of several papers and one book relating to high-speed interconnect design. Vira Ragavassamy is an Analog Engineer in the Embedded Communications Group at Intel with 16 years experience in Analog and RF system designs; he has spent the past 5 years at Intel solving signal integrity and power integrity problems in embedded products.

Ted Ballou is a Signal Integrity Engineer for Intel Server Products Group. Since joining Intel in 1998 he has been modeling and analyzing high speed differential interfaces, and has a background in test and measurement. Ted hold two patents related to clock routing and compensation. Michael Brownell is a Senior Package & Interconnect Engineer working on advanced interconnects technologies for Servers. Michael has 24 years experience in the field of interconnect technology development. He holds 19 patents related to various aspects of microelectronics packaging and interconnects.

Summary Over the last few years CPU signal frequencies have increased by a factor of 3X to 8X. Due to a corresponding increase for interconnect crosstalk risk, these performance gains add complexity to the signaling environment.

Several authors1, 2, 3 point out that passive elements in the circuit at low speed constitute nothing more than a wrapper around the active circuitry. At higher speeds the same passive elements have a profound effect on the behavior of the signal propagation as it traverses the interconnect. In addition to the task of controlling reflections and material loss, we see that managing crosstalk has now become a critical concern4. As a practical matter, system density and signal speed often make it impossible to improve one interconnect element without affecting another. Design success requires a systems approach with quantifiable cost/benefit estimates.

Introduction Interconnect Speed Signaling frequencies have moved into the microwave range, and most signals in today’s systems traverse electrically long structures. As speeds go above 1GHz, the interconnect is electrically long with respect to signal rise time, and multiple bits can be traversing the interconnect at any point in time. The channel is increasingly subject to signal integrity phenomena such as insertion loss, return loss, crosstalk, electromagnetic interference (EMI) and non-transverse electromagnetic mode (non-TEM) behavior.

Interconnect Type vs Signal Frequency (GHz)

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Figure 1. Relative increase in Signal Frequency by Interconnect Type At data rates of only a few hundred megahertz, transmission line losses are typically ignored. Above 1GHz, insertion loss starts to have a significant effect on the signal transfer. Return loss changes the effective impedance of the transmission line, and can cause resonances that severely distort the signal. The cascade of different system elements creates ripple in the insertion loss called insertion loss deviation (ILD). Together these factors serve to reduce the signal at the receiver to a fraction of that sent by the transmitter.

Bandwidth and Channel Capacity Shannon's Theorem gives an upper bound to the capacity of a link, in bits per second (bps), as a function of the available bandwidth and the signal-to-noise ratio of the link.5 The Theorem is stated in Formula 1.

C = B * log2(1+ S/N) Formula 1 Channel capacity limit as a function of available bandwidth and signal-to-noise ratio C is the achievable channel capacity, B is the bandwidth of the link, S is the average signal power and N is the average noise power. The signal-to-noise ratio (S/N) is usually expressed in decibels (dB). Crosstalk power is one component of noise; insertion loss reduces the power of the received signal. Hence channel capacity is reduced by I/O bandwidth and signal power limitations as well as loss and crosstalk. Crosstalk Crosstalk is an important noise factor which can degrade high bandwidth communication buses and is especially significant for single ended signals. It is one of many factors that must be simultaneously examined to optimize signal integrity. A good quantitative description of parallel line crosstalk is given by Hall & Heck6 using a terminated pair of traces running in parallel. Once a wave front hits a coupled structure, it begins coupling energy between the lines though mutual capacitance and inductance. Current that couples through mutual capacitance splits into forward-traveling and backward traveling components in the victim line. Current that couples though the mutual inductance travels back towards the source end. This results in a forward coupled wave that is a function of the difference between the capacitively coupled current and the inductively coupled current. The coupling of energy to the victim line continues as the aggressor propagates along the line until it reaches the far end. Arrival of the far end (or forward) crosstalk noise occurs simultaneously with arrival of the aggressor signal. Since the far end crosstalk travels along with the aggressor signal, the crosstalk noise pulse grows in amplitude but does not grow in width. Coupling occurs only during the signal transition, so the width of the forward coupled pulse will be approximately equal to the rise (or fall) time of the aggressor signal. For poorly terminated traces, there is additional energy coupling associated with the signal reflections.

Figure 2 Propagation of forward- and back- coupled noise halfway down the line Board designers are constrained by channel capacity requirements and concerned with feasible channel length, loading and component placement. Crosstalk is a channel impairment that can be expressed in terms of the impact it has on the channel margins. It limits what can be achieved in terms of routing density channel length, loading, and component placement. The magnitude of the impairment is a function of many factors that need to be considered simultaneously by modeling and solving the differential equations that represent the system dynamics. Typically these factors will include parallel trace coupling, non-linear interactions that can occur in I/O circuits as a result of common path induced simultaneous switching events, inter-symbol interference, non-TEM propagation in cavities between planes and resonant structures. In the time domain there is also the challenge of finding the worst cases of stimulus for stressing timing and voltage margins. Integrated insertion loss to crosstalk ratio (IICR) is a convenient method for estimating crosstalk in linear time invariant (LTI)7 systems so long as the designer knows the excitation frequency characteristics and the receiver specifications in frequency domain metrics.

Simultaneous Switching Noise When two or more switching circuits share a current path through a common power delivery structure, the impedance of that structure causes an interaction between them. This interaction is commonly referred to as simultaneous switching noise (SSN).

Figure 3 Schematic showing transmitters sharing a common current path for power delivery Consider the simple case in Figure 3 where a 1 nH inductor is used to isolate a number of transmitters from an ideal power source. The impedance of the structure (inductor) in this case results in significant coupling because of the common path from the transmitters to the ideal power rail. The coupling is significant enough to affect both the transmit and received signals Figure 4 Example of common path coupling through the power delivery path, transmission and receiver signals show a change in power-to-signal coupling with a change in impedance to common ground from (4a) 1 nH inductor to (4b) 0.05 Ohm resistor A change in the in the supply source impedance achieved by changing the value of the inductor to 1.0 nH to a 0.05 ohm resistor can significantly reduce simultaneous switching noise.

Fig 4b Fig 4a

Figure 5 Common path power delivery impact with data scrambling and reduced path impedance. Real power delivery structures can also support a number of resonant modes that affect the source impedance they supply to active devices. To limit the impact of rapid current demand associated with SSN there are two approaches that help. Lowering the plane impedance across the operating frequency range is the first. This can be done by adding decoupling and taking advantage of static loads (device leakage) which lower the Q. The second is to distribute the current demand in time. Data scrambling has the advantage of lowering the likelihood that all the bits of a bus will switch simultaneously in a coherent sequence and excite a power delivery resonance.

Non-ideal return path and non-TEM behavior High speed signals can propagate through structures other than normal traces and in modes beyond transverse electro-magnetic mode (TEM), with results such as unanticipated crosstalk sums. Consider a structure consisting of two conducting planes separated by a dielectric. In microwave terms this structure can be called a cavity waveguide. As in any waveguide, there are ways of introducing a signal into this structure. An excellent method is to drill a perpendicular hole through the two planes, run a wire through, and drive a current through that wire. The energy transmitted into this waveguide by this structure will be linearly proportional to the current in the wire and the distance between the planes. It is also proportional to the frequency of the current. Excitation of the Z-axis (via) structure launches waves in the dielectric that expand radially outward and scatter on other z-axis structures and plane edges resulting in non-TEM propagation modes (TEM, TEMmn, TEMnm where n and m are integers). A common metric that is often misused is the signal to ground ratio. It is useful in understanding capacitive coupling potential of a signal and return assignment (pin out). But it has little value as a metric for inductive coupling. Connectors, sockets and vias are Z-axis structures which can form signal paths that couple and resonate within a parallel plane cavity. To demonstrate the coupling potential of a Z-axis non-ideal return path, a 100-pin model is pinned out using a known pathological pin selection for single ended signals. Eight groups of signals are arranged around the victim pin forming an array of aggressors that act on the center pin of the array. Although each aggressor is adjacent to a ground pin, collectively they form an inductive implosion engulfing the victim when switched in phase.

Red=aggressorBlue =Victim Green=GND

Figure 6a Pathological Pin out with a 1:2.5 signal to ground ratio and figure 6b the resultant imploding wave front moving towards the victim pin in the center of the array with aggressors driven in phase through the structure Figure 7a. Coupling coefficient greater than 1 above 5.0 GHz and Figure 7b corresponding time domain simulation (victim near end – blue, victim far end – red) The results are plotted in Figure 7a showing a coupling coefficient greater than 1 above 5.0 GHz for in-phase excitation. This is confirmed in a time domain simulation as shown in Figure 7b. A simple signal to ground pin ratio does not guarantee performance of the interface. Vertical and return path effects need to be considered in terms of field theory rather than circuit theory.

Parallel Trace Proximity Interaction Trace proximity is another crosstalk parameter which is typically managed by a set of minimum spacing rules between traces. Although these constraint distances are rather inflexible (e.g. 3X multiple of dielectric height), they can be managed more efficiently using a tradeoff algorithm such as an ICR metric. ICR is the ratio of the insertion loss, measured from transmit to the receive ends, to the total crosstalk measured at the receiver. The system impact due to crosstalk is dependent on the insertion loss, since the signal to noise ratio is what matters. As an

Surface current induced in plane by H-Field imploding on the victim.

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example, a channel can tolerate more crosstalk when it has lower insertion loss. Hence insertion loss to crosstalk ratio (ICR)8 is a good metric to use for comparing different trace scenarios during early design analysis. A simple way of combining the effect of the transmitter and receiver bandwidth and ICR is integrated insertion loss to cross talk ratio (IICR). IICR is given by the following relationship:

IICRi = i

i ii

FfXSTS∑ Δ− *|))21log(|*20|)21log(|*20(

Formula 2 Integrated insertion loss to crosstalk ratio (IICR)

S21Ti is the through end loss, S21Xi is the crosstalk (NEXT or FEXT); Fi is the max frequency of the transmitter and receiver (assuming they are the same) and Δƒ is the spacing between frequency points (Fn – Fn-1) uniformly spaced across the spectrum. Near end crosstalk (NEXT) also reflects back to the receiver due to impedance mismatch at the Transmit end. Hence both FEXT and NEXT can degrade signal quality at the far end receiver.

Figure 8 NEXT reflecting back to Receiver

The effect of both IICR FEXT and IICR NEXT is examined for different trace geometries using a pair of strip line traces with the stack up shown in Figure 9.

Figure 9 Cross section of a pair of stripline traces

Figure 10 shows the relationship of trace spacing on the IICR (SNR), for different Tx/Rx bandwidths. As IICR is the equivalent to signal to noise ratio, a larger IICR is better. The insertion loss is kept constant by fixing both the length (10”) and the stack up properties (dielectric thickness and trace width).

Figure 10 Trace Spacing/Dielectric height ratios vs. IICR (SNR) for stripline and microstrip structures Figure 10 indicates the coupling in strip line trace is sensitive to spacing until approximately 5x the dielectric thickness, where the curve begins to flatten out. Each of the curves is weighted by integrating over the frequency bandwidths of 1, 3, 6 and 10GHz to show the effect of spectral content of the transmitted signal and receiver bandwidth on the coupling properties.

Figure 11 Relationships of Trace Width and Dielectric Thickness on IICR (SNR) Figure 11 shows the effect of trace width and dielectric thickness on SNR. Note that the relevant metric is the ratio rather than the absolute crosstalk noise value. IICR relationships can help avoid over-engineering a design. Microstrip has 2x to 3x lower IICR FEXT (2x to 3x higher cross talk impact) when compared to symmetric stripline. Hence microstrip demands much larger spacing than stripline traces to support the same interface and/or reduced length. Stripline traces have much higher IICR FEXT (lower FEXT) and less sensitivity to trace width and dielectric thickness. Reducing dielectric thickness helps to increase IICR NEXT (lowers NEXT) and hence improves SNR.

ISI and Board/Connector Interactions Inter-symbol interference (ISI) is the result of residual energy propagation or storage in a circuit and results from excitations occurring in previous unit intervals (a unit interval (UI) is the time duration that a single bit of information is valid). This stored energy causes non-zero initial conditions at the start of the next unit interval. Crosstalk amplitudes can be either enhanced or diminished by impedance mismatches in the transmission path. These mismatches cause reflections that can sustain propagation (i.e. resonate) as opposed to absorbing it. The conditions required to produce resonance (a form of ISI) in a structure are:

– At least two reflection coefficients with each approaching a magnitude of plus or minus one and separated in time by propagation delay.

– Relatively low energy loss in the signal path and its return path. – A high impedance source (like crosstalk) to excite the signal at the resonant

frequency of the structure. The resonant frequency of a structure is related to the propagation delay between the aforementioned impedance mismatches. If the signs of the reflection coefficients differ, then the lowest resonant frequency the structure will support is the quarter wave resonance. If the signs are the same the lowest resonant frequency is the half wave resonance. Higher resonant frequencies can occur in half wave increments. The loss in the circuit is a significant factor in determining the Q of the oscillator and thus the duration of the resonance after the energy source is quieted. If this exceeds a unit interval or crosses a unit interval boundary the result is a form of inter-symbol interference. ISI on aggressors can enhance or diminish the crosstalk energy depending on whether the interference is constructive or destructive. In general ISI is a behavior that is the result of both circuit structure and stimuli which change more rapidly than the rate at which steady state quiescence occurs due to losses. It can occur on power delivery, signal, and packaging structures due to impedance mismatches, delays [RC, L/R and (LC)-1/2], and excitation. ISI is often discussed as a characteristic of the signal transmission path, but it also applies to crosstalk. Strictly speaking, ISI results when the settled response of a digital signal transition extends beyond the adjacent unit interval. ISI can be created by dispersion and reflections. The simple example in Figure 12 illustrates one aggressor bit, the resultant crosstalk response at the receiver when the victim is held quiet, and a single bit response at the receiver without the crosstalk. Crosstalk from a noise source is superimposed on a victim. In the case where the noise source (the aggressor) is a data signal as in Figure 13 the duration of the noise signal (crosstalk) may extend beyond 1 UI. A pathological case occurs when the maximum crosstalk peaks coincide with the bit width. The signal would be reduced by the sum of absolute values of crosstalk (a+b+c+d+e+f+g+h), a total of 20 mV. Combinations of aggressor bits superimpose on the victim signal and add up in time.

Figure 12 Aggressor, crosstalk and victim response Certain board features exhibit another interesting behavior. Figure 13 is a three segment differential topology that can be used to explore crosstalk effects. For demonstration, crosstalk is limited to occurring in the vertical segment which in this case is a 750mil via. This vertical segment is analogous to a connector or more complex via configuration. The wave fronts are travelling in opposite directions in the Z axis, and they are travelling in the same direction in the horizontal axis forming a structure not unlike a LaGrange coupler. Figure 13 A three segment differential topology

Figure 14 Crosstalk Pulse Responses (UI=125ps), Victim Quiet The time between crosstalk peaks (Figure 14) in this configuration is proportional to the height of the vertical structure. For this example all the transmission lines are identical and have 75 ohm differential impedance. For constant total length, a sweep of Length A shows a small effect on ISI. Note that the eye opening in Figure 15 does not change much for Length A between 5 and 12 inches at 0 UI. Figure 15 Eye widths for crosstalk delay vs. connector separation The vertical structures are the dominant factor in producing coupling for this overall structure. Sweeping the phase relationship between the aggressor and victim is illustrated on the x-axis in Figure 15. The eye closes for delays between aggressor and victim skews that are greater than 3/8 of a UI. In a system with asynchronous signals, all possible skews between aggressor and victim would need to be considered. In summary, strongly coupled crosstalk from vertical features whose pulse responses have a periodicity anywhere near the UI can be problematic. These problems may not be detected in all systems since there can be system variation in crosstalk skew, so

screening can be ineffective. The best advice is to design for low crosstalk and include all crosstalk skew possibilities in a system analysis.

Crosstalk in Resonant Vertical Structures Via fields, connectors and sockets are usually perpendicular to the plane of circuit boards and so are referred to as vertical structures. The emphasis in the following paragraphs is primarily on via fields. Where there is a connector or a socket there will also be via fields. A structure that works well to transmit a signal also works well to extract a signal. The energy arriving at the extraction point will reduce as the square of the distance from the insertion point. The energy will also be reduced by mechanisms such as dielectric loss and resistive loss in the conductors. For best transmission efficiency, use a low-loss dielectric, high conductivity conductor, and maximize the current in the insertion device. If the input structure can be made a high-Q resonator, the current at the injection point is greatly enhanced. The current in a resonant structure is multiplied by the Q factor. At microwave frequencies impedance shifts cause signal reflections. Structures where multiple reflections take place, such as circuit boards with connectors, can produce a signal that bounces back and forth and thereby enhances the current in that particular region. Enhancing current enhances crosstalk. The structure of greatest concern is the high-Q resonator. A microwave engineer creates this structure by using a post that is tuned in length to be a quarter-wavelength at the frequency of choice. The circuit-board engineer calls this a resonant stub on a via. Unlike the microwave application, this is an unintended consequence of multi-layer board design – a case where a signal enters a via on one layer, exits on a nearby layer, and leaves a long stub with no intentional signals. If the length of this stub is such as to become resonant at a frequency near the operating bandwidth of the signals, the result is seldom good. An example of this type of structure is shown in Figure 16.

Figure 16 Resonant vertical interconnect structure in the PCB (under yellow BGA ball)

Figure 17 Lumped equivalent schematic for the structure in Figure 16

Figure 18a shows the effect of a via grounded on the top and un-terminated on the bottom resulting in a quarter wave resonator. Figure 18b shows the effect of also grounding at the bottom of the stub (half wave resonator). Note the frequency doubling. Had the stub been terminated on the bottom of the board the resonance would not have occurred.

Figure 18 Shift in resonance peak from (18a) at 3.7GHz to (18b) at 7.8GHz as a result of terminating the resonant vertical interconnect structure A 1GHz wavelength is 6 inches in FR-4 and a quarter wavelength is about 1.5 inches. At 5GHz, a quarter wavelength is about 0.3 inches in FR-4. At the current generation of data rates, board thickness greater than 62 mils should be closely examined for resonant stubs on the high-data-rate signal lines. These lines are often attached to connectors or sockets that extend their electrical length. Electrical length, defined by the geometric mean of the forward and return path delays, is the main factor in determining the resonant frequency. One means to shorten the electrical length in connection stacks is back drilling the vias.

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Figure 19 Illustration of backdrill technique used to minimize vertical stub length in through hole vias, typically of press fit connector vias

As frequencies have increased smaller structures begin to have the potential to resonate. Structures (including no-connects) need to be checked to ensure that they don’t have more than one reflection coefficient approaching 1.0.

S-parameters, Extraction and Modeling The system model used in the initial development of layout design rules is often based on a set of assumptions about the proposed topology. These assumptions can capture many of the important features that affect signal integrity but they rarely do justice to coupling mechanisms. To refine the coupling model it is desirable to base the model on the actual geometry and avoid making unsubstantiated assumptions about where, when and how much coupling exists. The use of scattering parameters as a basis for simulation models is a more appropriate method than RLGC models at frequencies greater than 1GHz. S-parameters can accurately represent not only the transmission and reflection but also the loss and coupling associated with passive interconnects. S-parameters introduce problems typical of sampled data structures. The spacing between frequency sampling points and the overall frequency range impact the accuracy of their modeling. In particular the issues of passivity, causality, truncation, aliasing, stability and accuracy must be handled with rigor to avoid misleading or incorrect results. Passivity and causality enforcement of high order coupled S-parameter models is often problematic and requires an iterative approach. Algorithms have been developed to ensure passivity and causality in measured or extracted S-parameters, but they have limitations and cannot be relied upon for repairing big problems. When a passivity or causality problem occurs it is best to solve the underlying problem in the S-parameter generation and not to rely on post-facto enforcement. Frequency domain measurements are usually more accurate than time domain measurements, and are increasingly used for high speed design and validation. As discrete-point models, S-parameters can lead to trouble when the placement of frequency points fails to align with important characteristics of the scattering behavior of the structure being modeled or measured. The use of adaptive frequency point selection

in a field solver can capture the important features. But chaining of discrete S-parameters from different structures with different frequency point distributions will lead to passivity and causality problems. The standard solution to this problem is to convert the separate series of frequency points into continuous functions and chain the continuous functions. Failure to avoid these types of problems will lead to numerical errors in time domain analysis that obscure and confuse the impact of coupling. The range of frequency represented in S-parameter models is also important when they are the basis for models used in the time domain. Often the original measurement will be extrapolated to higher frequencies, and this can lead to problems. It is wise to use platform design guides, when available, to greatly simplify the analysis required for a particular system topology. These guidelines are derived from significant amounts of analysis and contain detailed methods for avoiding coupling structures. The closer the actual system comes to matching the designs described in the platform design guide, the less risk incurred from coupled structures. While model extraction can be used to model new designs, it’s more cost effective when used mainly to examine special cases.

Summary & Conclusion Beyond the effects of parallel coupled lines, board designers need to develop an understanding of structures that are also the subject of study for microwave engineers. Crosstalk management is complex, but it’s helpful to review some important points. Switching noise can be managed by reducing power delivery impedance and data scrambling. Non-ideal return paths extend the range of coupling beyond what would be expected by just looking at signal to ground ratios. The relative proximity of the victim traces to the aggressors and return paths directly affect crosstalk. IICR is a convenient method to estimate crosstalk in systems where the LTI assumption is approximately valid. Resonant crosstalk interactions may be caused by connector and mounting choices as well as board routing choices. Vertical structures like vias in non through path circuits can affect crosstalk. And finally all of these components of crosstalk can be modeled and understood with S-parameters. Structures like the examples mentioned in this paper are problematic with respect to undesired coupling. Considerations are necessary for the application of S-parameters as a means to model undesired couplings. Many other structures can produce surprising coupling induced impairments.

References: 1. J. Clink, “Maximizing 10-Gbps Transmission Path length in Copper Backplanes with

and without Transceiver Technology”, in Design and Test for Multiple Gbps Communication Devices and Systems, International Engineering Consortium, Chicago, IL, 2005, pg 207.

2. H. Johnson, M. Graham, High Speed Digital Design: A Handbook of Black Magic, Prentice-Hall PTR, Upper Saddle River, NJ, 1993, pg 1.

3. B. Young, Digital Signal Integrity, Prentice-Hall PTR, Upper Saddle River, NJ, 2001, pg 42.

4. L. Chang, R. Mellitz, I. Ganga, “10G High Speed Serial Ethernet”, DesignCon 2007 5. L L Peterson and B S Davie, Computer Networks: a systems approach (Morgan

Kaufmann), 1996. ISBN: 1-55860-368-9 (Paperback ISBN: 1-55860-404-9 ), pp 94-95

6. S. Hall, H. Heck, High Speed Digital Designs, John Wiley & Sons, Inc, Hoboken, NJ, 2009, pg 166-171

7. R. Mellitz, M. Tsuk, T. Donisi, S. Pytel, Strategies for Coping with Non-linear and Time Variant Behavior for High Speed Serial Buffer Modeling, DesignCon 2008, pg 6-7.

8. IEEE Std 802.3ap™ - 2007, Annex 69B, IEEE Standard for Information technology Telecommunications and information exchange between systems Local and metropolitan area networks Specific requirements Part 3: Carrier Sense Multiple Access with Collision Detection (CSMA/CD) Access Method and Physical Layer Specifications Amendment: Ethernet Operation over Electrical Backplanes, pg192-194

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DesignCon 2010 Cost Effective Crosstalk Management & Decision Makings3.amazonaws.com/zanran_storage/ · 2012-04-30 · Cost Effective Crosstalk Management & Decision Making Mike White,