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DESIGN AND DEVELOPMENT OF A 3.6 GHZ DIELECTRIC RESONATOR OSCILLATOR WITH WIDE TUNING SENSITIVITY. By AMIR EFFENDY BIN MUHAMMAD AFIFI Thesis submitted in fulfillment of the requirements for the Degree of Master of Science September 2011

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Page 1: Design & Development of a 3.6 GHz Dielectric Resonator ...eprints.usm.my/43225/1/AMIR EFFENDY MUHAMMAD AFIFI.pdfDESIGN AND DEVELOPMENT OF A 3.6 GHZ DIELECTRIC RESONATOR OSCILLATOR

DESIGN AND DEVELOPMENT OF A 3.6 GHZ DIELECTRIC

RESONATOR OSCILLATOR WITH WIDE TUNING

SENSITIVITY.

By

AMIR EFFENDY BIN MUHAMMAD AFIFI

Thesis submitted in fulfillment of the requirements for the

Degree of Master of Science

September 2011

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ACKNOWLEDGEMENTS

All praises to Allah, unto Him belongs all the knowledge and understanding. I

would like to acknowledge and extend my heartily gratitude to the following people

without whom the completion of this research would not have been possible. I wish

to express my appreciation and thankfulness to my supervisor, Dr. Widad Ismail

who was very helpful and offered invaluable advice, support and guidance; not

forgetting my initial supervisor, Dr. Mandeep Singh for his encouragement and

advice that motivated me to take up this research. My appreciation to my manager,

Lim Kok Keong, who shares the same admiration for knowledge and ‘technology’;

for his persistent support in all my works. I would also like to convey thanks to

Agilent Technologies for funding the materials in this research. To my colleagues

Law Boon Wan, Toh Chee Leng and Ahmad Helmi Mokhtar for their relentless help

and technical contribution, let me express my sincere gratitude. I wish to express my

love and gratitude to my beloved wife, Alhan Farhanah, my children and family; for

their understanding & sacrifice, throughout the duration of my study. I thank you to

all of you and indeed Allah is the best for reward and the best for the final end.

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TABLE OF CONTENTS

ACKNOWLEDGEMENTS ..................................................................................... ii

Table of Contents ................................................................................................... iii

LIST OF TABLES ................................................................................................ vii

LIST OF FIGURES ................................................................................................ ix

LIST OF SYMBOLS .......................................................................................... xvii

ABBREVIATIONS .............................................................................................. xix

ABSTRACT.......................................................................................................... xx

ABSTRAK .......................................................................................................... xxii

1 INTRODUCTION ........................................................................................... 1

1.1 Motivation – A High Tuning Sensitivity Second Down-Converter Local

Oscillator ............................................................................................................. 1

1.2 Research Problem Statement ..................................................................... 1

1.3 Research Objectives .................................................................................. 3

1.4 Research Scope and Limitations ................................................................ 4

1.5 Research Contribution ............................................................................... 5

1.6 Thesis Organization .................................................................................. 5

2 LITERATURES REVIEW .............................................................................. 7

2.1 Microwave Frequencies Sources ............................................................... 7

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2.2 Active Device Consideration for Oscillator ............................................... 8

2.3 High Q-factor Resonator ......................................................................... 11

2.4 Dielectric Waveguide as a Microwave Resonator .................................... 13

2.5 Discussion on DR tuning techniques prior works..................................... 14

2.6 Electrically Driven Tuning Options ......................................................... 15

2.6.1 Magnetic and Optic Tuning DRs ...................................................... 15

2.6.2 Varactor Diode ................................................................................. 17

2.6.3 Varactor Tuned DRs ........................................................................ 19

2.7 Literature Review Summary .................................................................... 30

3 DESIGN OF A DIELECTRIC RESONATOR OSCILLATOR ...................... 33

3.1 Dielectric Resonator Oscillator Specification .......................................... 33

3.2 Development Flowchart .......................................................................... 34

3.3 Dielectric Resonator ................................................................................ 35

3.3.1 Resonant Mode ................................................................................ 36

3.4 Resonance Structure Construction ........................................................... 38

3.4.1 Metal Enclosure Effect on Resonant Mode and Frequency ............... 39

3.4.2 DR Coupling to Microstrip ............................................................... 43

3.4.3 Mechanical Adjustment of Resonant Frequency ............................... 46

3.4.4 Electronic Tuning Element – Varactor.............................................. 47

3.4.5 Microstrip Stub Coupled Varactor Tuning – Tunable Resonant Circuit

........................................................................................................ 53

3.5 Microwave Oscillator Fundamentals ....................................................... 56

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3.5.1 Positive Feedback Oscillator ............................................................ 57

3.5.2 Negative Resistance Oscillator ......................................................... 59

3.6 Phase Noise Consideration ...................................................................... 61

3.6.1 Active Device Selection – Bipolar Junction Transistor ..................... 63

3.7 Reference Dielectric Resonator Oscillator Design ................................... 64

3.7.1 Gain Block Design for Positive Feedback Oscillator ........................ 64

3.7.2 Integration of the Gain Block and the Feedback Block ..................... 66

3.8 Chapter 3 Summary................................................................................. 66

4 SIMULATIONS, MEASUREMENTS AND EXPERIMENTS TO IMPROVE

DRO KV ................................................................................................................ 67

4.1.1 Amplifier Matching for Gain and Open Loop Phase Shift ................ 67

4.1.2 Phase Noise...................................................................................... 72

4.1.3 Tunable Resonance Structure Coupling to Amplifier Block .............. 73

4.1.4 Signal Coupling to Output ................................................................ 75

4.2 Initial DRO Performance ......................................................................... 77

4.2.1 Fundamental Frequency and Tuning Sensitivity ............................... 77

4.2.2 Power & Harmonics ......................................................................... 80

4.2.3 Phase Noise...................................................................................... 82

4.3 Analysis and Experiments to Improve DR Tuning Bandwidth ................. 83

4.4 Investigating the Effect of Tuning Stub Dimensions on Resonant Structure

Tuning Bandwidth. ............................................................................................ 91

4.4.1 TE01 H-field Pattern in the Resonant Structure ................................ 92

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4.4.2 Tunable Resonant Structures Frequency Response Simulations ........ 97

4.5 Validating the New DRO Functionality ................................................... 98

4.5.1 Realization of a Negative Resistance Oscillator Alternative ............. 99

4.5.2 Modification of Tunable Resonant Structure (vi) ............................ 101

4.6 Chapter 4 Summary............................................................................... 103

5 RESULTS AND DISCUSSION OF THE PROPOSED DRO ...................... 104

5.1 Negative Resistance DRO Performance................................................. 104

5.2 Comparison with Prior Works and Researches ...................................... 108

5.3 Chapter 5 Summary............................................................................... 115

6 CONCLUSION AND FUTURE WORKS ................................................... 116

6.1 Conclusion on the High KV and Wide Tuning Range DRO Design ........ 116

6.2 Future Work on the High KV DRO ........................................................ 117

6.2.1 Phase Noise Improvement .............................................................. 117

6.2.2 Further KV or Tuning Range Improvement for Cost Reduction

Opportunity ................................................................................................. 118

LIST OF PUBLICATIONS ................................................................................. 120

APPENDIX A ..................................................................................................... 121

APPENDIX B ..................................................................................................... 122

APPENDIX C ..................................................................................................... 123

APPENDIX D ..................................................................................................... 124

APPENDIX E ..................................................................................................... 125

REFERENCES ................................................................................................... 126

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LIST OF TABLES

Page

Table 2.1 Summary of prior works that achieved wide tuning

bandwidth.

31

Table 3.1 Specification of the DRO. 33

Table 3.2 Overview of ceramic materials available for the

dielectric resonator

35

Table 3.3 Off-the-shelf metal enclosure dimensions 40

Table 3.4 Calculated waveguide resonant wavelength, d/2 for

available materials (Trans-Tech 2003).

41

Table 3.5 Electrical properties of 4500 series dielectric

resonator fabricated for this design, based on the

manufacturer measurement.

42

Table 3.6 Fabricated dielectric resonator physical dimensions. 42

Table 3.7 Materials used in the gain block or amplifier

construction.

65

Table 4.1 Antenna-waveguide structure simulation and

measurement data.

76

Table 4.2 Summary of proposed tuning structures performance. 91

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Page

Table 5.1 Comparison of harmonics level between negative

resistance oscillator and initial positive feedback

oscillator.

106

Table 5.2 High KV negative resistance DRO specifications 108

Table 5.3 Summarized comparison between proposed DR

tuning technique and related prior works.

114

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LIST OF FIGURES

Page

Figure 2.1 Q factors of some resonators used in microwave

frequency circuits (Carter and Flammer 1960; Day

1970; Gopinath 1981; Plourde and Chung-Li 1981;

Trans-Tech 2003; Gardner 2005; Mazierska, Krupka

et al. 2006).

12

Figure 2.2 Optical tuning dielectric resonator with

photoconductive semiconductor as part of tuning

stub.

17

Figure 2.3 Varactor ; (a) abrupt junction and (b) hyperabrupt

junction

18

Figure 2.4 Dielectric resonator resonant circuit with balance

loop varactor tuning (El-Moussaoui, Kazeminejad et

al. 1989).

20

Figure 2.5 Balanced loop varactor circuit (El-Moussaoui,

Kazeminejad et al. 1989)

21

Figure 2.6 Varactor tuning with the tuning stub electrical length

optimizable (Xiaoming and Sloan 1999).

24

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Page

Figure 2.7 Circular tuning stub proposed by Virdee (Virdee

1997), top view.

25

Figure 2.8 Popovic’s tunable DR and resonant structure

(Popovic 1990).

26

Figure 2.9 Multi-resonator DRO schematics, (a) parallel

feedback and (b) series feedback configurations.

28

Figure 2.10 Complication in using multi-resonator DRO for

wider tuning bandwidth.

29

Figure 3.1 A cylindrical DR with TE01δ mode, (b) top view and

(c) side view, in the case where L < OD (Cohn 1968).

37

Figure 3.2: (a) Microstrip, TEM mode (Gupta, Garg et al. 1996);

(b) dielectric resonator TE01 resonant mode coupling

to microstrip TEM mode; 2 >> 1

44

Figure 3.3: Electrical equivalent schematic circuit of the

dielectric resonator coupling to a microstrip line.

45

Figure 3.4: Initial stage breadboard circuit developed to

characterize microstrip lines coupling to the dielectric

resonator, dotted lines are indicating maximum H-

field coupling points.

45

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Page

Figure 3.5 Simplified assembly of the DR resonant structure 47

Figure 3.6 BB857 diode capacitance (CT) at 1 MHz versus the

reverse voltage applied or the vtune, taken from the

datasheet (Infineon 2006).

49

Figure 3.7 ADS model of packaged BB857 constructed from its

SPICE model and package model.

49

Figure 3.8 Simulation of packaged BB857 in ADS and EMDS at

1 MHz

52

Figure 3.9 Tunable resonant circuit assembly 55

Figure 3.10 Plot of the tunable resonant circuit simulated and

measured transmission response (S21)

56

Figure 3.11 Positive feedback oscillator block diagram 57

Figure 3.12 Schematic of a positive feedback dielectric resonator

oscillator

58

Figure 3.13 Equivalent circuit of a one port negative resistance

oscillator

59

Figure 3.14 Negative resistance transistor oscillator 61

Figure 3.15 Leeson's model of feedback oscillator noise. 62

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Page

Figure 3.16 Typical oscillator phase noise spectra (Gardner

2005).

63

Figure 4.1 Layout of the amplifier circuit, the schematic is in

APPENDIX D.

68

Figure 4.2 The amplifier gain and phase shift measurement

setup.

68

Figure 4.3 The amplifier gain, the markers are showing the gain

at 3.6 GHz.

69

Figure 4.4 The amplifier phase shift, the markers are phase

shifts at 3.6 GHz.

70

Figure 4.5 Simulated and measured noise figure of the designed

amplifier

71

Figure 4.6 The amplifier noise figure measurement setup. 71

Figure 4.7 Predicted phase noise for the designed oscillator 73

Figure 4.8 Measured response of the resonant structure

described in section 3.4.5.

74

Figure 4.9 Complete oscillator circuit resting in bottom

enclosure

75

Figure 4.10 Oscillator signal coupling to the output 77

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Page

Figure 4.11 Spectrum measurement set up. 77

Figure 4.12 DRO center frequency reading. 79

Figure 4.13 The positive feedback DRO f0 versus vtune. 80

Figure 4.14 Frequency spectrum of the dielectric resonator

oscillator

81

Figure 4.15 Phase noise measurement using Agilent E5052B

signal source analyzer

82

Figure 4.16 Dielectric resonator oscillator phase noise, measured

and simulated.

83

Figure 4.17 Proposed tunable resonant circuit designs. 86

Figure 4.18 Frequency sweep (solid lines) and insertion loss

(dotted lines) versus vtune measurements for structure

(i) – , (ii) – and (iii) – , tuning ranges shown

in percentage.

87

Figure 4.19 Frequency sweep (solid lines) and insertion loss

(dotted lines) versus vtune measurements for structure

(iii) – and (iv) – , tuning ranges shown in

percentage.

88

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Page

Figure 4.20 Frequency sweep (solid lines) and insertion loss

(dotted lines) versus vtune measurements for structure

(iv) – , (v) – and (vi) – , tuning ranges shown

in percentage.

89

Figure 4.21 Frequency sweep (solid lines) and insertion loss

(dotted lines) versus vtune measurements for structure

(vi) – and (vii) – , tuning ranges shown in

percentage.

90

Figure 4.22 Resonant structure model for EMDS simulations. 93

Figure 4.23 Magnetic field patterns of the tunable resonant

structures; (a), (b), (c) corresponds to structure (iv),

(v), (vi) respectively.

96

Figure 4.24 Frequency response simulations of the tunable

resonant structures; solid lines are resonant

frequencies and dotted lines are insertion loss.

97

Figure 4.25 Negative resistance source implementation 99

Figure 4.26 Simulated and measured negative resistance source

return loss

100

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Page

Figure 4.27 Smith chart showing impedances and reflection

coefficients of negative resistance source (ZnegR,

negR) and tunable resonant structure (ZDR, DR).

101

Figure 4.28 Modified tunable resonant structure (vi) – a one port

network

102

Figure 4.29 The negative resistance oscillator, housed in the same

package as the previous positive feedback oscillator.

103

Figure 5.1 Measurement of the designed negative resistance

dielectric resonator oscillator tuning range – solid

line and power – dotted line.

105

Figure 5.2 Spectrum of the negative resistance dielectric

resonator oscillator.

106

Figure 5.3 Phase noise of the two oscillators. 108

Figure 5.4 Tuning range of the negative resistance DRO (,

solid line) and the corresponding DR resonant circuit

(, dashed line), i.e. structure (vi).

109

Figure 5.5 Experimental results of the optically-tuned dielectric

resonator (Jianping and Deming 1998).

110

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Page

Figure 5.6 At distance of 4 mm from top cover, the balance-loop

varactor circuit performance, tuning range = 3 MHz;

resonant frequency (, solid line); Q factor (,

dashed line) (El-Moussaoui, Kazeminejad et al.

1989).

111

Figure 5.7 Tuning range as a function of the balance-loop

varactor circuit distance from the top cover; theory

(, dashed line); experiment (, solid line) (El-

Moussaoui, Kazeminejad et al. 1989).

112

Figure 5.8 - Resonant frequency, - unloaded Q obtained

using Virdee's tuning configuration (Virdee 1997).

113

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LIST OF SYMBOLS

Reflection coefficient

Wavelength in vacuum, free space or air

Frequency in radian

Transistor current gain

C Resonant frequency in radian

d Guide wavelength

r Dielectric constant

1/f Flicker noise

Cres Dielectric resonator equivalent capacitance

CT Varactor capacitance

F Noise figure

Varactor built-in potential

f0 Instantaneous oscillation frequency

fC Resonant frequency

fm Baseband frequency, offset frequency

f0 fundamental signal or carrier frequency

fT Transit frequency

GA Amplifier power gain

GaN Gallium nitride

IC Collector DC current

k Boltzmann constant, 1.3806503 × 10-23 JK-1

Ka Refering to 18 – 27 GHz frequency band

KV oscillator tuning sensitivity or oscillator gain

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Lres Dielectric resonator equivalent inductance

LT Tuning stub equivalent inductance

N Coupling coefficient (i.e. in transformer)

P1dB Output power 1 dB compression point

PO Output power

QL Loaded Q

QU Unloaded Q

T Temperature (in Kelvin)

tan Losses in the (dielectric) material

TE01 Transverse electric resonant mode in dielectric resonator

TE011 Transverse electric resonant mode in cavity resonator

TE111 Transverse electric resonant mode in cavity resonator

vb Base voltage

VCC Supply voltage, e.g. at collector

Vi Input signal, in volt

Vo Output signal, in volt

vtune Varactor bias, tuning voltage

vDC Instantaneous varactor tuning voltage

W Single sided noise spectral density

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ABBREVIATIONS

ADS Advanced Design System

BJT Bipolar junction transistor

CW Continuous wave

DC Direct current

DR Dielectric resonator

DRO dielectric resonator oscillator

E-field Electric field

EMDS ElectroMagnetic Design System

FET Field effect transistor

H-field Magnetic field

IL Insertion loss

LO Local oscillator

PCB Printed circuit board

PLL Phase-locked loop

SA Spectrum analyzer

SSA Signal source analyzer

TEM Transverse electric magnetic propagation mode

TWT Travelling wave tube

UHF Ultra high frequency

VCO Voltage controlled oscillator

YIG Yttrium-iron-garnet

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DESIGN & DEVELOPMENT OF A 3.6 GHZ DIELECTRIC

RESONATOR OSCILLATOR WITH WIDE TUNING

SENSITIVITY.

ABSTRACT

An oscillator is required as a second stage LO in a superheterodyne SA. The

oscillator operating frequency is a fixed 3.6 GHz, which is at the lower end of the

microwave frequency range. There are several options of active devices and

resonators that can be considered for the oscillator. A bipolar junction transistor

(BJT) is chosen for the amplifier block due to its low flicker noise corner frequency

and a dielectric resonator (DR) is chosen for its high Q factor. This combination

yields a low phase noise oscillator. Apart from its high Q factor, a DR is a high

dielectric constant ceramic thus enabling a miniaturized microwave oscillator design

compared to a cavity resonator. A varactor-tuned technique is adopted because it

results in a simple planar circuit design compared to optically and magnetically tune

DR.

This dielectric resonator oscillator (DRO) must have very high frequency accuracy.

The SA is specified to operate from 0C to 55C, thus among the design requirement

for the DRO is to be operable in a wide temperature range and, to last for many,

many years. Hence the DRO is controlled by a phase-locked loop (PLL). As the

DRO signal drifts with temperature as well as due to aging, a wide tuning range is

necessary to guarantee a reliable and repeatable performance over its operating life.

An existing DRO with a tuning range of 0.14% at of 3.6 GHz was used as a

benchmark. The development of the new DRO began with investigation on several

proposed varactor-tuned DR resonant structures. The resonant structures were

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observed for the resonant frequency tuning range, the linearity of resonant

frequencies versus tuning voltages and the tuning sensitivity. The promising DR

resonant structures – with wide tuning range, linear response and high tuning

sensitivity, were further analyzed to understand the resonant structures coupling

mechanism as well as the potential effect on the DRO performance like phase noise.

The successful DR resonant structure combined with the BJT amplifier circuit

formed the DRO. The first DRO is a positive feedback oscillator; however due to the

resonant structure high insertion loss compared with the available amplifier gain, the

DRO failed to work. An alternative negative resistance oscillator was then

developed, also using the same model BJT. It worked based on signal reflection

between the resonant structure port and the BJT emitter junction which posed a

negative real impedance; it proved successful. The newly developed negative

resistance DRO performance was measured and compared with the benchmark

DRO. The new DRO yields a tuning range of 23 MHz or 0.65% at 3.6 GHz and a

tuning sensitivity of 2.3 MHz/V. However, the new DRO phase noise degraded by

about 10 dB compared with the benchmark DRO.

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REKABENTUK & PEMBANGUNAN LITAR AYUNAN PENALA

DIELEKTRIK 3.6 GHZ DENGAN KEPEKAAN MENALA YANG

LEBAR.

ABSTRAK

Sebuah pengayun diperlukan sebagai pengayun setempat peringkat kedua di dalam

sebuah penganalisa spectrum superheterodyne. Frekuensi tunggal pengoperasian

pengayun ini ialah 3.6 GHz, yang mana terletak di hujung rendah julat frekuensi

gelombang mikro. Terdapat beberapa pilihan peranti aktif dan penala yang boleh

dipertimbangkan untuk pengayun tersebut. Transistor dwikutub (BJT) dipilih untuk

blok penguat kerana frekuensi sudut hingar kelipannya (1/f) yang rendah dan penala

dielektrik (DR) dipilih kerana faktor Q’nya yang tinggi. Gabungan ini menghasilkan

pengayun yang rendah hingar fasa. Selain dari faktor Q’nya yang tinggi, DR ialah

seramik berpemalar dielektrik tinggi, jadi ia membolehkan pengecilan rekabentuk

pengayun gelombang mikro berbanding dengan penala berongga. Teknik talaan-

varaktor digunakan supaya rekabentuk litar yang rata dan ringkas dapat dihasilkan

berbanding DR talaan optik atau magnet.

Pengayun penala dielektrik (DRO) ini wajib mempunyai ketepatan frekuensi yang

tinggi. Spesifikasi SA menetapkan pengoperasian dari 0C hingga 55C, oleh itu

antara keperluan rekacipta DRO ini ialah kebolehan beroperasi dalam julat suhu

yang lebar dan bertahan untuk betahun-tahun lamanya. Oleh sebab itulah DRO ini

dikawal oleh lingkaran fasa-berkunci (PLL). Oleh kerana isyarat DRO boleh tersasar

mengikut perubahan suhu dan juga akibat penuaan jangka hayat, maka, julat talaan

yang lebar adalah satu keperluan untuk memastikan prestasi yang terjamin setiap

kali, sepanjang hayat pengoperasiannya.

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Sebuah DRO sedia ada dengan julat talaan sebanyak 0.14% pada 3.6 GHz telah

digunakan sebagai penanda aras. Pembangunan sebuah DRO yang baru bermula

dengan kajian ke atas beberapa cadangan struktur talaan DR yang ditala-varaktor.

Pemerhatian dilakukan ke atas julat penalaan frekuensi, kelurusan frekuensi talaan

lawan voltan menala dan kepekaan menala. Struktur talaan DR yang berpotensi –

dengan julat menala yang lebar, respons yang lurus dan kepekaan menala yang

tinggi, dianalisa selanjutnya untuk memahami mekanisma gandingan struktur talaan

dan juga kesan yang berkemungkinan ke atas prestasi DRO seperti hingar fasa.

Struktur talaan DR yang terpilih digabungkan dengan litar penguat BJT, seterusnya

membentuk sebuah DRO. DRO yang pertama ialah sebuah pengayun suapbalik

positif, walau bagaimanapun, akibat kehilangan sisipan (IL) struktur talaan yang

tinggi berbanding dengan gandaan penguat yang ada, DRO tersebut gagal berfungsi.

Sebagai ganti, sebuah pengayun rintangan negative dibangunkan, menggunakan

model BJT yang sama. Ia berfungsi berasaskan kepada pemantulan isyarat di antara

liang struktur talaan dan simpang pemancar BJT yang menunjukkan galangan nyata

negatif; ia terbukti berjaya. Prestasi DRO rintangan negatif yang baru dibangunkan

ini diukur dan dibandingkan dengan DRO penanda aras. DRO yang baru

membuahkan julat menala sebanyak 23 MHz atau 0.65% pada 3.6 GHz dan 2.3

MHz/V kepekaan menala, dengan voltan menala dari 0 – 10 V. Walau

bagaimanapun hingar fasa DRO baru ini merosot lebih kurang 10 dB berbanding

DRO penanda aras.

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1 INTRODUCTION

There are many microwave frequency oscillators in the market serving the ever

growing electronic communications. Most of the shelf devices are tailored to the

commercial applications like mobile phones, wireless broadband data and

broadcasting industries. Although there are suppliers that do custom designs for

unique applications or low volume market, they come with high price tags; hence

the need for in-house designs. Secondly, proprietary technologies are best kept with

in-house design.

1.1 Motivation – A High Tuning Sensitivity Second Down-Converter Local

Oscillator

A swept signal spectrum analyzer (SA) requires a fix frequency oscillator operating

at 3.6 GHz for its second down-converter block. Although there is probably an

application out there that works at 3.6 GHz, a tight requirement on this oscillator, as

the SA is a test and measurement instrument, means the available of-the-shelf

oscillator may not meet the requirement – electrical and mechanical specifications as

well as physical size (form, fit and function). The SA is specified to operate over a

wide temperature range i.e. from 0C to 55C and over a long period of time. Thus

the oscillator frequency may drift over temperature as well as over time. Therefore,

the oscillator requires a tuning mechanism for the frequency drift compensation; the

frequency correction will be controlled by a PLL.

1.2 Research Problem Statement

In microwave fixed frequency oscillators as well as microwave filters, some limited

tuning capability is necessary for center frequency adjustment. Dielectric resonator

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materials have some fabrication tolerance which results in resonant frequency

variation. This problem is commonly addressed by means of mechanical

adjustments. However this is a onetime adjustment during manufacturing and it

cannot correct the resonant frequency should there be drifts throughout the device

operating life. A broad frequency range electronically tuned method as proposed in

this research offers a solution to alleviate this problem. It should be applicable either

in dielectric resonator oscillators and filters.

As for filters, particularly band pass filters, apart from the problem discussed above,

having an electronic tuning capability allows for bands selection. I.e. instead of

designing a broad band filter that covers the whole frequency channels, a narrower

band pass filter could be designed which may improves adjacent channel rejection.

The filter can be controlled to ‘pass’ only the channels it is operating.

The subsequent problem which will be the focus of this research is an example

where a broad, electronically tuned capability is required. The SA second local

oscillator specification requires a tuning bandwidth of 10 MHz and tuning

sensitivity, KV = 1 MHz/V) to enable the oscillator to provide a reliable and

consistent fixed 3.6 GHz signal to the down converter mixer in the SA. The output

power for the 3.6 GHz signal is +3 dBm 1 dB. Physically, the oscillator must be

small enough to fit in a typical bench top SA.

Looking at the frequency and output power requirement as well as the size, a solid

state oscillator is the most sensible option. With the choice of resonators available a

planar circuit is feasible. For a low phase noise oscillator design; the solid state

active device choice, e.g. a transistor or a diode must consider the flicker noise; and

secondly a high Q-factor resonator (Gardner 2005). The temperature coefficient of

the resonator resonance frequency must have a low part per million (p.p.m.) number

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for stability over wide temperature range. Materials selection will be further

discussed in the following sections.

1.3 Research Objectives

Based on the defined problem statement, the objectives of the research can be

defined as follows:

(i) To design a dielectric resonator oscillator (DRO) that must achieve a tuning

bandwidth of at least 10 MHz with tuning voltage of 0 V to +10 V.

The main objective of this design is to establish a circuit technique that can

achieve the wide tuning bandwidth; the choice of materials like transistor,

resonator and varactor are secondary. In this study, a method for achieving

wide tuning bandwidth and high tuning sensitivity, KV for DRO will be

investigated and verified with a final design of the DRO.

(ii) To fabricate and measure the proposed DRO design that considers the need

for it to suit manufacturing requirement

The solution is using low cost, off-the-shelf components; manufacturable with

existing industrial technology e.g. printed circuit board (PCB) in mass

production environment.

(iii) To analyze and validate the performance of the proposed DRO for practical

oscillator application and finalized the specification of such DRO in

which, the tuning sensitivity must be linear to ease PLL control.

This study only focuses on the design of the wide tuning bandwidth DRO.

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1.4 Research Scope and Limitations

The scope of this study is to find an electronically-tuned technique; that yields a

wide tuning bandwidth for a passive dielectric resonator (DR) circuit and

subsequently apply it to a DRO design and test the DRO functionality. The activities

include:

1. Literatures review on microwave signal sources, DR technologies, related

materials and devices to construct microwave oscillators and prior works and

techniques on DROs;

2. Characterize a DR circuit and the corresponding DRO (previously developed

by the author) to be used as a reference for the prospective oscillator,

3. Simulation of several initial proposed passive DR circuits, fabrication of the

designed circuits and measurements of the circuits.

4. Promising circuits will be further analyzed to find a trend that will give a

wide tuning bandwidth. Finally the passive circuits will be adopted into DRO

designs to prove the technique effectiveness.

This research does not intend to include in-depth study and improvement on phase

noise for the prospective DRO; however phase noise performance of the ‘reference’

and ‘prospective’ oscillators will be measured for completeness.

In the design, especially in simulation related to non-linearity parameters like the

amplifier gain and its open loop phase; reflection coefficient of negative resistance

port are not very correlated to their corresponding measurements. This is due to

limitation to components non-linearity parameters in the models in particular the

transistor.

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Circuit fabrication accuracy is as good as the prototyping machine capability, hence

some losses in measurements that are unaccounted for in simulations like insertion

loss and amplifier gain. The three dimensional electromagnetic field simulator,

EMDS, uses a lot of computer resources. Additional components used in supporting

measurement like RF SMA connectors are not included in the three dimensional

electromagnetic field simulation to reduce model complexity.

1.5 Research Contribution

This research proposes a technique to increase DRO tuning range with high tuning

sensitivity; suitable where only small vtune range is available. The technique offers a

low impedance varactor tuning stub that couples to the DR, exploiting the DR TE01

resonant mode. It is a practical approach where mass production is intended as most

of the components to build the circuit, are available off the shelf. Minimal custom-

made components are necessary, which are unique to the desired operating

frequency such as the metal enclosure, the DR and the PCB, of course. The

technique is also applicable for DR filter application as stated earlier in Section 1.2.

1.6 Thesis Organization

This introductory chapter presents the background of the problem that leads to this

research and development on the prospective DRO. This is followed by objectives of

the study and the expected end results.

The following literature review chapter begins with a brief review of microwave

sources technologies currently available, related materials and devices to construct

microwave oscillators and how the selection of materials relates to the theory, some

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fundamentals on DR applications followed by discussion on DROs prior works and

techniques.

In chapter 3, the research flowchart is presented. From here the proposed passive

tunable resonant circuits and experiments will be laid out to find out a tunable circuit

that is promising for a wide tuning bandwidth. Finally a brief notes on oscillator

fundamentals.

In Chapter 4 the analysis and results from all experiments are presented and

analyzed in determining a tunable circuit that can yield a wide tuning bandwidth.

The second part of this chapter elaborates how a potentially wide tuning bandwidth

resonant circuit is designed into a DRO. The potential wide tuning bandwidth DRO

design presented in Chapter 4 is further discussed in Chapter 5, in comparison with

existing and prior works or reports.

Finally the study’s findings is discussed and summed up; and in particular in the

justification of adopting the design in the said application as well as other potential

applications. Potential future works would also be discussed in which how to

possibly improves the phase noise, an important figure of merit for an oscillator and

secondly the possibilities to further increase the tuning range and/or tuning

sensitivity, KV.

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2 LITERATURES REVIEW

In general, microwave frequency signal sources are either wide band and tunable

like YIG-tuned oscillators, voltage controlled oscillators and frequency multipliers;

or fixed frequency like coaxial resonator oscillators and DROs. Although

categorized as fixed frequency, coaxials and DROs do have limited tunability for

frequency correction. The categories only imply the oscillators’ applications. A

microwave oscillator (wide band or fixed frequency) is basically a high Q-factor

resonator with an appropriate coupling structure, couples to an active device.

2.1 Microwave Frequencies Sources

Microwave frequencies range from 300 MHz to 300 GHz, i.e. equivalent to

wavelength of one meter down to one millimeter respectively (Pozar 1998). This is a

broad definition that includes ultra high frequencies (UHF) all the way to millimeter

waves. Commonly, microwave frequencies usually refer to 3 GHz to 30 GHz, or 10

cm to 1 cm wavelength at minimum, however in RF engineering the lower boundary

is usually at 1 GHz (30 cm), and the upper around 100 GHz (3 mm).

Microwave sources have different types of constructions and unlike lower

frequencies, are not limited to PCB. Vacuum tube devices include the magnetron,

klystron, traveling-wave tube (TWT), and gyrotron, and operate on the ballistic

motion of electrons in a vacuum under the influence of controlling electric or

magnetic fields. Vacuum tube devices are cumbersome but are capable of very high

output power, for example a continuous wave (CW) helix TWT yields an output

power in the range of several kilowatts at X band (Coaker and Challis October

2008). A ‘microwave amplification by stimulated emission of radiation’ (maser)

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produces coherent electromagnetic waves through amplification due to stimulated

emission; it has a very high frequency precision (Calosso, Levi et al. 2005). Solid-

state sources can be constructed from bipolar junction transistors (BJT), field-effect

transistor (FET) – at high microwave frequencies, tunnel diodes, Gunn diodes, and

IMPATT diodes. Solid state microwave oscillators are relatively smaller in size than

the above two sources; and can be either planar circuits i.e. on PCBs or combination

of planar circuits and physical structures, depending on the choice of resonators.

Until recently, solid state devices are capable of low to medium power handling in

microwave applications. With the advent of GaN devices, wideband capability

(ABI-Research 2009) and high power applications are possible all way into Ka band

(Shealy, Smart et al. 2002). Such prospect combined with lower cost gives

microwave solid state oscillators a promising future.

2.2 Active Device Consideration for Oscillator

An important parameter for any oscillators is its signal purity. As for harmonics, if

any of the harmonics signal is too high, it could cause the PLL to lock on that

harmonic instead of the fundamental signal, f0; secondly it could also create a

spurious signal at a mixer output like the one this oscillator is intended for. A low

pass filter can be added at the output of the oscillator to suppress the harmonics. The

harmonics arise due to the active device non-linearity. To reduce the harmonics

power level, the device must have much higher P1dB than the required output power

for the f0.

For phase noise, the active device must have low 1/f noise (or the flicker noise)

because this noise is up converted to the sideband of the oscillator output signals.

BJT has large parasitic capacitance to the ground, resulting in reduction of the 1/f

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noise. Also, 1/f noise is directly proportional to the current density in the transistor.

Large transistors with high maximum collector current, when used at low currents

will give the best 1/f performance. Harmonics can mix 1/f noise up and mix back to

the fundamental signal – this would cause higher phase noise. Choosing BJT with

low fT, means less harmonics generated and thus lower phase noise. A simple

guideline is the device fT should be two or three times the operating frequency (Muat

1984). FET devices although they too can offer high P1dB, unfortunately have higher

1/f corner frequency. A typical submicron MOSFET device, with a bias current of

several hundred microamperes would show a 1/f corner frequency around 1 MHz;

and this is worse than BJT (Razavi 1997; Zhaofeng and Jack 2001). FET would be

considered for oscillators operating beyond 10 GHz if the phase noise is not

demanding as there aren’t many BJTs available for operation above 10 GHz. Now

with the presence of heterojunction bipolar transistors (HBT), the bipolar transistor

technology is making its way even beyond 20 GHz as demonstrated by (Madhihian

and Takahashi 1991); the HBT phase noise performance is either better (Ali

Khatibzadeh and Bayraktaroglu 1990; Leier 2006) or comparable to BJT.

Gunn diodes are applied mostly in higher microwave frequencies i.e. 10 GHz

onwards because the Gunn diode oscillators require cavity resonators, many works

discussed oscillator applications in millimeter waves for example (Ruttan 1975;

Rahal and Bosisio 1995; Priestley, Newsome et al. 2002), and an oscillator at 35

GHz presented by (Strangeway, Ishii et al. 1988) has quite low phase noise too e.g. -

132 dBc/Hz to -125 dBc/Hz at 100-kHz offset from f0. Though theoretically it is

possible to use the Gunn diode in this work, the cavity resonator size at 3.6 GHz

makes it impractical for miniaturization. The cavity resonator will be further

explained in the following section. Until recently, for many Gunn diode oscillators,

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the frequency tuning is realized by the Gunn diode DC bias adjustment to create

frequency pushing, this will complicates the design (Priestley, Newsome et al.

2002). Nowadays though, there are varactors offered for Gunn diode oscillator

applications, like those offered by e2v Technologies (e2v 2008). IMPATT diodes

are similar to Gunn diodes though they are not interchangeable. Their DC operating

current is typically in hundreds of milliamps with output power ranges from tens to

hundreds of miliwatts.

There are microwave voltage controlled oscillator (VCO) designs (Sheng-Lyang,

Huang et al. 2008; Jhe-Jia, Zuo-Min et al. 2009) on integrated circuit (IC) for

tunable oscillator applications with tuning range in hundreds of Megahertz to a few

Gigahertz. Phase noise of tunable oscillator will suffer degradation due to the

presence of a tuning element like varactor; and the wider the tuning bandwidth the

poorer the phase noise (Carey 2011). Bulk acoustic wave (BAW) resonator

oscillator is also an IC based but it is for a fixed frequency or narrow band

application. For technological reasons BAW-based oscillators are limited up to 2

GHz (Li, Seok et al. 2011) though there are works (Aissi, Tournier et al. 2006; Li,

Seok et al. 2011) showing BAW-based oscillators at higher frequencies, achieved by

some sort of frequency multiplication. For a unique application i.e. this 3.6 GHz

oscillator, an integrated circuit VCO would require a substantial financial

investment.

Of all the choices of solid state devices mentioned above, discreet BJT is the most

suitable for low noise design especially for low microwave frequencies; in fact

Regis et al stated clearly their choice of BJT in (Regis, Llopis et al. 1998) for their

ultra low phase noise oscillator; and Warburton for his 3 GHz oscillator (Warburton

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2005).Considering cost, an off the shelf component is preferred over a dedicated IC

designed oscillator.

2.3 High Q-factor Resonator

There are many choices of microwave resonators available for application in the S-

band. Among others are cavity resonator (Condon 1941), yttrium-iron-garnet (YIG)

crystal (Carter and Flammer 1960), dielectric resonators (Plourde and Chung-Li

1981) – both transverse electric magnetic (TEM) mode (coaxial resonator) and TE01

mode, and microstrip resonators to name a few. Among these resonators, microstrip

resonators have the lowest Q factor which ranges in a few hundred (Gopinath 1981;

Mazierska, Krupka et al. 2006) and they suffer from radiation loss due to their

construction on low dielectric constant substrates (Belohoubek and Denlinger 1975).

The cavity resonator must be designed to resonate in TE011 mode instead of at the

dominant cylindrical cavity mode, TE111, at 3.6 GHz in order to achieve higher Q

(Pozar 1998). For this, its radius and height are approximately 5.5 cm and 11 cm

respectively, which are rather bulky and furthermore the structure is made of metal,

the dimensions tend to change with temperature.

YIG is a ferrimagnetic material that exhibits ferrimagnetic resonance. A YIG crystal

in sphere shape when subjects to a DC magnetic field (H-field) and placed in an RF

structure, exhibits a high Q resonance at a microwave frequency proportional to the

DC H-field. The resonance frequency is tunable a few octave range by varying the

DC H-field and there is a minimum DC H-field required depends on the YIG

saturation magnetization value. The YIG sphere is only a few tens of mils in size but

the solenoid structure that provides the DC H-field is cumbersome and that, this

design is focus on a fix frequency oscillator and does not require a few octave of

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tuning range. Figure 2.1 summarizes the Q factors of common and commercially

available resonators used in microwave frequency circuits.

Figure 2.1: Q factors of some resonators used in microwave frequency circuits

(Carter and Flammer 1960; Gopinath 1981; Trans-Tech 2003; Gardner 2005; Yom,

Shin et al. 2005; Mazierska, Krupka et al. 2006; Yom, Shin et al. 2007).

The two possible candidates left are either TEM mode coaxial resonator or TE01

mode DR. In terms of size they both are reasonably small though coaxial resonator

is smaller than the later. However, in general the TE01 mode DRs possesses a much

higher Q i.e. in the range of thousands whereas the coaxial resonators Q are in the

range of a few hundreds.

The TE01 mode DR material is a titanate based ceramic and will be used in the

design because of its excellent frequency stability over temperature and over time.

The high Q-factor is due to the ceramic material inherently very low loss i.e. tan

(Fiedziuszko, Hunter et al. 2002); an important feature for short term frequency

Microstrip

TEM mode DR

YIG sphere

TE01 mode DR

TE011 mode cavity

10000 20000 30000 40000 50000 60000 Q factor

4000 – 50000

10000 – 40000

100 – 400

150 – < 2000

500 – < 4000

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stability or phase noise ( is the loss angle from '''tan , and is the electrical

permittivity). Else the DRO would require a much wider tuning bandwidth and

possibly the PLL must have a relatively faster lock time or wider loop bandwidth

which could potentially degrade the far-out phase noise.

2.4 Dielectric Waveguide as a Microwave Resonator

Guided electromagnetic wave propagation in dielectric structures have been

discovered as earlier than 1920. In 1935 G. C. Southworth received U.S. Patent

2106769 describing, “The wave guiding structure may take a variety of forms: a

typical guide consists of a rod of dielectric material having high dielectric

coefficient relative to unity”. “A specific dielectric guide which may be considered

is a cylinder of ceramic material having rutile (titanium dioxide) as its principal

constituent” (Southworth 1935). In 1939 Richtmeyer showed that unmetallized

dielectric materials can perform as electrical resonators and he termed dielectric

resonators (Richtmyer 1939).

In their paper, Okaya and Barash presented the first analysis on high dielectric

materials resonant frequency, modes and DR circuit design (Okaya and Barash

1962). However due to temperature instability – variation of dielectric constant over

temperature, of the early high dielectric constant materials like rutile, they don’t find

practical usage (Fiedziuszko, Hunter et al. 2002). Finally in 1970s the first

temperature stable ceramic, barium tetratitanate was developed by Raytheon (Masse,

Pucel et al. 1971). This paved the way for DRs applications in microwave

frequencies. Later on, composite DRs with high dielectric constant and adjustable

temperature coefficients were introduced (Wakino, Nishikawa et al. 1975) allowing

for more commercial applications.

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Energy is confined in a dielectric waveguide by total internal reflection mechanism

whereby a core dielectric material with higher index of refraction is surrounded by

dielectric material with lower index of refraction (Snitzer 1961) e.g. air. The DR has

very small, finite loss tangent (tan ) and no wall losses in the dielectric, hence the

high unloaded Q, this is given by tan1uQ ; especially true for high dielectric

constant e.g. 100 or more. The electric field decays exponentially outside the

resonator. The resonant frequency depends on the dielectric constant (r), the

dimensions and the shape of the resonator (Okaya and Barash 1962; Cohn 1968).

Some common application of high dielectric constant materials are band-pass filters,

dielectric antenna, DROs and ferroelectric devices(Fiedziuszko, Hunter et al. 2002).

2.5 Discussion on DR tuning techniques prior works

Many papers have proposed various means of tuning the DR, and most papers (El-

Moussaoui, Kazeminejad et al. 1989; Krupka 1989; Buer and El-Sharawy 1995;

Virdee 1997; Xiaoming and Sloan 1999; Alford, Petrov et al. 2002) discuss their

proposals based on DR resonant circuits as examples rather than explicitly on DROs.

This is understandable due to the fact that microwave DROs constructions are

basically a DR resonant circuit coupling to an active device circuit like a transistor

or a Gunn diode. The resonant circuits can also be the construction blocks for band

pass filters.

With regards to the oscillator construction, a standalone resonant circuit that

promises the desired tuning specifications – Q factor, range and sensitivity, will not

necessarily achieve the same tuning characteristics when connected as an oscillator.

For example the Q factor, a standalone resonant circuit would have a high loaded Q

(QL) but when coupled to an active device circuit which would have different

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impedances at its ports; will create a mismatch and loss to the overall circuit. This

results in the QL being degraded.

Thus it is important too, to validate a promising DR resonant circuit with an

oscillator example. The literature review will nevertheless discuss the prior works

which are mostly on DR resonant circuit examples. Many DRO examples (Ishihara,

Mori et al. 1980; Kharkovsky, Kirichenko et al. 1996; Yom, Shin et al. 2007) don’t

emphasize on wide tuning range and high tuning sensitivity which on the contrary is

the main objective of this work.

2.6 Electrically Driven Tuning Options

DR is tuned effectively by perturbing its electromagnetic fields in the excited

resonant mode. The perturbation can be achieved mechanically and electrically. The

interest in this work is on electrically exited perturbation given the application of the

DR resonant circuit as an oscillator controlled by a PLL. Several widely known

electrical tuning techniques by means of magnetic, varactor, and optic are discussed

in these article (Virdee 1998) and paper (Farr, Blackie et al. 1983).

2.6.1 Magnetic and Optic Tuning DRs

Magnetically tuned DR is realized by attaching a microwave ferrite material to the

resonator. H-field which is usually provided by solenoid is applied to the ferrite to

control its magnetic property. This is a current controlled tuning system. Krupka, in

his paper (Krupka 1989), proposed putting a ferrite rod co-axis with the DR disc.

He achieved a very high tuning bandwidth of 120 MHz on his magnetically tuned

resonant system, considering the hysteresis in magnetic structure. Otherwise, with a

completely demagnetized ferrite to the maximum applied H-field, he achieved 405

MHz tuning range ( 4 %). The downside of this magnetically tuned system is the

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slow tuning speed, as Krupka also hope to obtain a faster tuning speed. Secondly the

electromagnetic structure that provides the H-field for his system is bulky, i.e. the

diameter of the electromagnetic poles is 250 mm. In fact this is typical drawback of

electromagnetic structures for similar applications. This negates the advantage of

DR for miniaturization.

Interest in optically tuned DR arises because the technique offers high isolation

between the tuning DC bias and the radio frequency (RF) signals, and potentially

very wide bandwidth. The first method, a photoconductive semiconductor slab is

attached on top of the DR cylinder. A light from a diode or a laser illuminates the

semiconductor slab thus creating a plasma layer on the slab illuminated surface. The

plasma behaves like a conductive tuning plate and the amount of frequency shift

depends on the plasma density, which is proportional to the illumination intensity

(Shen, Nickerson et al. 1993). Another optical method proposes putting a

photoconductive semiconductor patch as part of microstrip tuning stubs as shown in

Figure 2.2. The tuning stubs add reactance to the DR, whereby the photoconductive

patch connects the two stubs. By varying the light intensity illuminating the

photoconductive patch, the overall reactance of the resonator circuit is altered. Thus

the resonant frequency of the DR is tuned (Jianping and Deming 1998). This method

achieves a very much wider tuning range as oppose to the former (Shen, Nickerson

et al. 1993); both using Silicon (Si) as the photoconductive material for tuning at X-

band in their experiments.

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Figure 2.2: Optical tuning dielectric resonator with photoconductive semiconductor

as part of tuning stub.

Despite the achievement and advantages of the optical tuning techniques, the fact

that a light source has to be inserted means the DR metal enclosure has to have an

opening for the light source. This will potentially cause some loss to the radio

frequency signal as well as external interference. For this research, having to add a

light source is not favorable as it is more complex as compared to the varactor

tuning techniques that will be discussed in the following section.

2.6.2 Varactor Diode

Varactor or variable capacitance diode enables fast electronic tuning and it achieves

this without any physical movement. There are two type of varactors based on the

doping structure introduced inside the semiconductor. An abrupt diode has the

doping structure such that its gamma, is about 0.47, for a practical diode

(Klymyshyn, Kumar et al. 1997); where is the slope of the curve log (varactor

capacitance, CT) versus log (vtune + ); vtune is the varactor bias voltage and is the

varactor built-in potential, this is depicted in Figure 2.3. A hyperabrupt diode has the

Dielectric resonator Microstrip line

Tuning stubs

Substrate Ground plane

Photoconductive patch

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doping structure such that its > 0.5 (Frazier 1959; Covington and Hicklin 1978).

For varactors considered for this research, the abrupt junction diodes capacitance

ratio (capacitance at minimum vtune/capacitance at maximum vtune) is typically about

4 (Khanna 1987) and the hyperabrupt junction diodes, is > 10 (Infineon 2006). In

other words, the hyperabrupt junction varactor will give bigger tuning range

compared to the former. Al-Charchafchi and Dawson developed a varactor-tuned

ring resonator with a wide tuning bandwidth, of 31% at 3 GHz using a hyperabrupt

junction varactor (Al-Charchafchi and Dawson 1989). Hence for this research, the

hyperabrupt junction varactor will used. However, for hyperabrupt junction

varactors may not be constant over the vtune range. For that reason, for a limited

range of vtune the can be assumed constant.

(a) (b)

Figure 2.3: Varactor ; (a) abrupt junction and (b) hyperabrupt junction

Adding an electronic tuning element like varactor is not without a cost. When the

varactor is biased at a particular vtune, the resulting RF signal in the resonant circuit

causes a jittery vtune. The jittery vtune translate into phase noise on the RF signal

Log (vtune + )

> 0.5

Log

(CT)

Log (vtune + )

47.0

tune

T

vLogCLog

Log

(CT)

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(Carey 2011). Consider an oscillator with a tuning sensitivity KV and tuning voltage

vtune. The oscillation frequency, f0 is

tuneV vKff 000 2.1

Where f00 is the oscillation frequency at vtune = 0 V.

At a particular vtune = VDC, the presence of RF signal in the resonant circuit causes

jitter to vtune and is assumed to vary as such;

2/maxmin DCDCtune VVv 2.2

And thus the range of the instantaneous oscillation frequency, f0 is

minmax0 DCDCV VVKf . 2.3

The f0 appears as the sideband spectrum of the oscillation frequency and is

observed as the phase noise due to the varactor function.

2.6.3 Varactor Tuned DRs

In varactor tuned resonant circuit, looking from lump elements circuit model, the

varactor capacitance (CT) is the variable reactance that is added to the resonator

(parallel Rres, Lres ,Cres) and thus varies the resonant frequency. The papers discussed

in this section explain how the varactor is coupled to the DR and thus form the

tunable DR circuit. In summary, the varactor reactance coupling to the DR is by

means of the H-field when the DR is excited in TE01 resonant mode as depicted in

Figure 3.1.

2.6.3.1 Balanced Loop Varactor Circuit

El-Moussaoui et al proposed a balanced loop varactor circuit (El-Moussaoui,

Kazeminejad et al. 1989) where the varactor loop is mounted above the DR as

shown in Figure 2.4 below. The loop uses two varactors and is wounded in opposite

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direction to cancel the loop inductance and consequently prevents the varactor loop

from resonating with the DR. The varactor loop detail is depicted in Figure 2.5.

However the physical loop construction is not described in the paper but one can

expect that for a reliable and consistent performance, a firm mounting is required.

Otherwise the ‘dangling’ loop would give inconsistence coupling to the DR. Even

without the varactors being biased, the fluctuation of the loop itself is sufficient to

perturb the H-field.

Figure 2.4: Dielectric resonator resonant circuit with balance loop varactor tuning

(El-Moussaoui, Kazeminejad et al. 1989).

RF in

RF out

Varactor loop

DR

Low r DR support

+ -

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Figure 2.5: Balanced loop varactor circuit (El-Moussaoui, Kazeminejad et al. 1989)

Moussaoui’s results however, are interesting for this work as they explain the

performance of the DR circuit under weak and strong varactor couplings. Under

weak coupling, the tuning range is small, low insertion loss (IL) at resonant

frequency and high unloaded Q (QU) factor is maintained. Whereas under strong

coupling, the tuning range achieved is big but at a cost of high IL at resonant as well

as QU factor degradation.

2.6.3.2 Tuning by Coupling to a Non-Resonant Microstrip Line

Practically the manufactured DR resonant frequency accuracy comes with a certain

tolerance. The one used in this work is specified with 18 MHz. With a tuning

voltage limited to 10 V for this work, it will require such a high tuning sensitivity to

cover more than 36 MHz only to compensate for the tolerance, not to mention the

oscillator tuning in phase-locked operation. A passive tuning or adjustment using

+ -

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three-quarter wavelength microstrip line (Buer and El-Sharawy 1995) can be used to

compensate for the DR frequency tolerance. This technique would be convenient

because it allows for a planar circuit layout. By adjusting the position of the three-

quarter wavelength microstrip line, the amount of H-field perturbation can be

controlled and hence the nominal resonant frequency of the DR can be set. The

technique is further developed to include varactor for electrical tunability whereby

the microstrip line length is reduced to half wavelength with the varactor

incorporation. For this method however, since the position of the microstrip line

would be different from one circuit to the other, the wiring of the varactor bias

cannot be fix too, i.e. it has to follow the position of the microstrip line. This will

possibly require fine manual wiring and is undesirable for mass production

environment.

As can be seen, DR tuned by varying external reactance (e.g. a photoconductive

patch, a varactor or a certain wavelength microstrip line) requires the device to be

coupled to a tuning line or stub (El-Moussaoui, Kazeminejad et al. 1989; Buer and

El-Sharawy 1995; Virdee 1997; Jianping and Deming 1998). The reactance

translates into the TEM mode H-field along the tuning stub that effectively couples

to the DR. The tuning stub length determines the tuning range of the DR (Virdee

1997; Xiaoming and Sloan 1999). However the stub length is fixed or patches (very

short, e.g. square shape stubs) are added at the one end of the line for limited length

adjustment. This is because, due to materials variations, the predetermined tuning

stub length may not be optimum to some circuits compared to the others; and thus

yields varying results.

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2.6.3.3 Secondary Varactor to Optimize Tuning Stub Electrical Length

Xu and Sloan (Xiaoming and Sloan 1999) proposed a more flexible tuning stub

adjustment by adding a secondary varactor. The primary varactor, as with other

methods discussed earlier is to vary the reactance of the DR. The secondary varactor

is to adjust the tuning stub electrical length and hence ensuring the optimum results

for each circuit.

Figure 2.6(a) shows the equivalent electrical schematic of Xu’s proposed varactor

tuned DR circuit, showing only the tuning stub schematic. The primary varactor for

resonator tuning is on the right hand side and the secondary varactor for electrical

length, i.e. open circuit (OC) adjustment in on the left hand side. Figure 2.6(b)

shows the actual circuits with the DR, the two varactors are in Mesa diode package.

By measuring the DR maximum tuning range at different electrical lengths; the

tuning stub electrical length that gives the widest resonator tuning range can be

determined, the detailed results of this experiment is presented in the paper

(Xiaoming and Sloan 1999). However, the loading on the DR Q factor must also be

considered, therefore a trade-off between tuning range and unloaded Q factor is

usually necessary. With this technique, Xu achieved 89 MHz of tuning range at 8

GHz or 1.1%. The addition of the secondary varactor for electrical length

optimization on itself is simple; however the complexity is on the control circuit that

will set the bias voltage for the varactor and to ‘lock’ the bias voltage at the set value

consistently.

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(a) (b)

Figure 2.6: Varactor tuning with the tuning stub electrical length optimizable

(Xiaoming and Sloan 1999).

2.6.3.4 Circular Tuning Stub

Virdee (Virdee 1997) proposed a rather compact varactor-microstrip tuning stub

coupling to the DR by making the microstrip stub circular as shown in Figure 2.7.

The loop and the transmission lines dimensions must be determined for optimum

tuning range as well as not to degrade the DR QU too much. The curvy lines

maximize the field couplings. Virdee does not mention whether the loop and the

transmission lines dimensions were obtained empirically or by some calculations.

The DR is tunable by more than 1.6% or 53.7 MHz at 3.318 GHz with tuning

voltage from 0 V to 30 V. The resonant frequency relation with the tuning voltage

(or the varactor capacitance) is approximately linear, with this technique. This

compact and practical method has a potential and will be further investigated in this

work.

DR

Adjusting varactor

Tuning varactor

Optimum OC

point