1

Abstract—This paper studies the use of a voltage source

converter (VSC) with DC capacitors as energy storage medium

for the compensation of pulsating active and reactive power of

CERN’s proton synchrotron (PS) particle accelerator. The PS

accelerator load demands periodic, active and reactive power

pulses of about 2s duration and a magnitude of up to 45MW and

65Mvar. The proposed compensator is able to control both

reactive and active power exchange with the network in order to

eliminate network disturbances. The controllability study reveals

that the best control strategy is to use q-axis converter input for

active power and d-axis input for reactive power regulation. An

analytical system model is created to study the system dynamics

and to aid the controller design. The eigenvalue study with the

MATLAB model reveals that, with large energy storage units and

small converter losses, there is only a small interaction between

the control channels. The final testing is done with a detailed

non-linear model in PSCAD/EMTDC. The simulation results

show that it is possible to fully compensate the active power

exchange with the network during typical accelerator cycles, and

at the same time to achieve excellent AC voltage control.

Index Terms— State space methods, Root Loci, Eigenvalues.

Accelerator power supply, Proton accelerators, Energy storage,

Pulse width modulated power converters.

I. INTRODUCTION

CERN, the European Organization for Nuclear Research, is

an international organisation with 20 Member States. It’s

objective is to provide for collaboration among the European

States in the field of high-energy particle physics research.

CERN designs, constructs and operates the necessary particle

accelerators and the associated support equipment.

For a power system, particle accelerators represent heavily

pulsating electric loads with a variable power factor, which is

mainly caused by the twelve-pulse and six-pulse thyristor

converters. Because of the large magnitudes and short rise

times of the pulsating power, rapid reactive power

compensation and voltage control is necessary. In addition,

dedicated filtering is required to eliminate the harmonics

generated by the power converters. For this purpose, CERN is

currently operating nine 18kV Static Var Compensators (SVC)

with an installed total power of more than 500Mvar [1].

The Proton Synchrotron (PS) is the oldest and the most

versatile of CERN's accelerators, which has diameter of 200

meters and reaches a final energy of 28GeV. At present, the PS

D. Jovcic is with University of Aberdeen, Aberdeen, AB24 3UE, UK d.jovcic@abdn.ac.uk

K. Kahle is with CERN’s Electric Power Systems Group, CH-1211 Geneva

23, Switzerland. karsten.kahle@cern.ch

complex can accelerate all stable and electrically charged

particles (electrons, protons), their antiparticles (positrons,

antiprotons), and different kinds of heavy ions (oxygen, sulfur,

lead), which are then injected into the larger rings for further

acceleration.

The PS accelerator is continuously pulsating with a cycle

time of about 2s. The largest cycles (pulse 1) require power of

up to 45MW and 65Mvar at the cycle peak, and have a rise

time of 600ms, as seen in Figure 1.

In order to limit disturbances to other loads, the PS is

decoupled from the network using a motor-generator supply.

An integrated large rotating mass serves as a storage medium

for smoothing the power pulses. The 6MW supply motor

represents a stable load to the 18kV CERN network.

By now, the rotating machines have been in service for over

34 years, and CERN has initiated an investigation of

compensation options based on FACTS [2] technology.

Initially, the option of a conventional thyristor-based SVC

and a more modern STATCOM [2] for reactive power

compensation was explored, confirming that good 18kV AC

voltage regulation is possible. However, the remaining steep

pulses of active power gave concern related to periodic torque

variations at the small generators in the local supply network.

This initiated the study of a technical solution with active and

reactive power compensation.

This paper presents the study of the use of a Voltage Source

Converter (VSC) with DC energy storage elements for the

compensation of pulsating active and reactive power.

A STATCOM is also able to regulate active power

exchange if an energy storage medium is connected to the DC

side. Various options exist for the storage media, including

batteries, fuel cells, capacitors, supercapacitors and Super-

conducting Magnetic Energy Storage (SMES) [3].

Considering the fast power dynamics and high number of the

PS accelerator operating cycles, batteries or fuel cells would

not provide optimal dynamics and long-term reliability [3],

and the adequate medium would be only SMES,

supercapacitors or DC capacitors. STATCOM with SMES is

suggested [4],[5] as a possible solution for larger energy

storage (mainly over 50MJ) with fast dynamic response. On

the downside, such an element incorporates the main converter

with a bi-directional DC-DC chopper which increases

compensator losses. Also, because of the SMES technology

and the converter capital investment, at present such an

element may not be an economically viable solution for the

required energy storage level. As an outcome, DC capacitors

are selected as the storage medium.

A STATCOM with large DC capacitors will require

different design approaches and much different control

Compensation of Particle Accelerator Load

Using Converter Controlled Pulse Compensator D. Jovcic, Member IEEE and K. Kahle

2

strategies from the conventional STATCOM. This

compensation element is labeled Converter Controlled Pulse

Compensator (CCPC) to distinguish from the name

STATCOM with energy storage, which is frequently used

with SMES. The aim of this study is to investigate using a

CCPC for the PS particle accelerator compensation.

The paper firstly describes the modeling of the PS particle

accelerator. The design section briefly describes calculations

of main compensator parameters and the control strategy in

more detail. Simulation results with a non-linear digital

simulation are presented in the last section.

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25

35

45

55

65

0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2 2.2 2.4 2.6 2.8 3 3.2 3.4 3.6

Time [sec]

Active P. [MW], Reactive P. [MVAr] Pulse 1

Pulse 2

P -[MW]

P -PSCAD

Q -[MVAr]

Q -PSCAD

Figure 1 PSCAD/EMTDC model verification for two typical Proton

Synchrotron pulses. The P and Q curves are obtained by measurements on the

actual system.

II. PROTON SYNCHROTRON MODELING

The PS accelerator supply consists of two twelve-pulse

thyristor converters that supply DC power to the accelerator

magnets. The main electrical circuit is shown in Figure 2,

together with the proposed future supply configuration. The

PS control system consists of two control levels. The inner,

fast level has two DC voltage feedback control loops, one for

each twelve-pulse group. At the outer control level, there is a

DC current control loop, which has the role of keeping the

firing angle within the operating range. It also prevents a

commutation failure in the inversion operating mode. The

external input signal for PS control is the DC current

reference, which has the shape of square pulses. These

reference pulses are pre-calculated in the technical control

room on the basis of the accelerator cycle demand.

The system model is developed with a PSCAD/EMTDC [6]

simulator. Some control circuit parameters are not known;

thus they are estimated to match the measured power curves.

Figure 1 shows that the PSCAD/EMTDC model achieves

excellent matching against the power measurements.

In addition, a MATLAB-based small signal linear model is

developed to enable dynamic studies, as discussed below.

III. CCPC DESIGN

The simplified circuit of the CCPC compensator is

presented in Figure 2. The element consists of a two-level

VSC and large DC capacitors. It is connected to the 18kV

network using a reactor Ls, without the need for a transformer.

Based on the load power demand in Figure 1, a rating of

35MVA for the harmonic filters and ±55MVA for the converter is chosen to enable adequate control margin.

The reactor Ls is selected on the basis of the maximum

harmonic current and the maximum DC voltage deviations.

Sinusoidal PWM control is assumed with the pulse number of

27 (ratio between triangular career frequency and the sine

wave frequency). According to converter theory [7] the 25th

harmonic will have the largest magnitude. The harmonic

current through the converter can be determined using basic

circuit equations and assuming AC filters (zf) are tuned to this

harmonic frequency. Figure 3 shows the 25th harmonic current

as a function of reactor size. A reactor size of Ls=6mH is

chosen to keep the harmonic current below 7% of the

fundamental value. Note that this current is not injected in the

network since it is eliminated by the filters. Higher value for

Ls would improve the harmonic profile but it would also lead

to larger DC voltage variations. A higher DC voltage

requirement, on the other hand, implies reduced voltage range

available for the energy storage, resulting in increased

capacitor rating.

The DC capacitor size is determined considering the

energy storage requirements. With reference to Figure 1, the

positive and negative energy requirement for the largest power

pulse is determined as the surface under the active power

curve:

zac400

~

18/2.12kV

Y/Y

Y/Y DC

filters

main

magnets

Proton synchrotron

Supply network

Compensator

DC

filters

18/2.12kV400/18kV

18kV

4km cable CB

Y/

Y/

Vdc1

Vdc2

z f

ac

ps

s

dc

LR

I

I

I ,

I

α

α

P ,Q

I

I

V

I

V

V

VSC

C

C

ss

s

dc

ls ls

psdc

ps2

ps1

ls

f

c

s

s

Figure 2 Proton synchrotron with the proposed supply configuration.

3

MJE

MJE

pulse

pulse

112/55.040

5.19154.02/6.045

1

1

=×=

=×+×=

−

+ (1)

The aim is to develop a compensator that can fully supply

and absorb the pulse energy and in this way eliminate any

power variation from the supply network. In an ideal

compensation case, the network supplies only the PS power

consumed and losses over a cycle. Taking the difference

between Epulse1+ and Epulse1- we obtain the energy consumed

during one pulse. The average power supplied from the

network can be calculated as Pav=5.5MW, which does not

include the converter losses. Subtracting the energy supplied

from the network, and assuming some margin for converter

losses we obtain the maximum energy demand for the

compensator of Emax=16.0MJ.

DC capacitors exchange energy on the basis of their

voltage variations as:

( )2min

2max

2DCDC

totCCPC VV

CE −= (2)

At present, commercially available VSC’s have maximum

DC voltages of about VDCmax=60kV. The minimum DC voltage

is limited by the requirement for reactive power control. At

any level of stored energy, full reactive power exchange must

be possible to enable AC voltage control. Therefore, the

minimum voltage VDCmin must be greater than the maximum

DC voltage required for reactive power control. Figure 4

shows the reactive power control operating curve for the

CCPC assuming the reactor size as discussed above. The

lowest Vdc curve is the DC voltage variation across reactive

power exchange range assuming that control magnitude is

Mm=1. The PWM magnitude input (Mm) regulates the AC

voltage as presented below. The maximum voltage in the

curve with maximum Mm determines the minimum DC

voltage for storage range, which is VDCmin=36kV.

The converter AC voltage curve in Figure 4 (Vs) is

uniquely determined on the basis of the reactive power

exchange requirement. The AC voltage regulation at higher

DC voltage levels is achieved by reducing the control signal

magnitude Mm as shown with the curves in the storage region.

As the basic energy storage units we are considering the

commercially available DC capacitors C1=2.4mF and

V1dc=6kV. A total of 580 of these units (10 in series by 58

parallel) would have Ctot=Cs/2=13.92mF and according to (2)

their stored energy is just above the required value of 16MJ.

IV. CONTROLLABILITY ANALYSIS

A. Controller based on dynamic decoupling

This section investigates the control strategy for the

CCPC. Since there is a requirement for two control loops, one

for AC voltage control and the other for active power control,

the main design challenge is to develop high gain controllers

without cross-coupling between the control channels.

A decoupled control method has been suggested as the

control strategy for VSC converters in [5] and [8], which is

discussed firstly. The basic electrical equations for the CCPC

Ls

Figure 3. Series inductance Ls selection on the basis of harmonic current.

Vdc (Mm

=1)

Vs

storage

region

Vdc (Mm

=0.8)Vd

c (Mm

=0.67

)

Vdc (Mm

=0.4)

Figure 4. Operating curve for reactive power compensation.

in Figure 2 can be obtained by applying the rules for d-q

transformation [5],[9], since the AC voltage with a PWM

controlled converter is Vsd=0.5MdVdc, Vsq=0.5MqVdc,:

acddcdsqosdssd

s VVMIIRdt

dIL +−−−= 5.0ω (3)

acqdcqsdosqs

sq

s VVMIIRdt

dIL +−+−= 5.0ω (4)

where ωo=2π50[rad/s], and subscripts d and q denote corresponding components. Md and Mq are the rectangular

frame components of the control signal M (Mm and Mϕ are polar components) for the two-level PWM controller

converter. Rs represents the internal losses in the converter.

Assuming that the co-ordinate frame is attached to the AC

voltage (Vacm=Vacd), active and reactive power are controlled

using the respective converter currents:

sdacmIVP 3= , sqacmIVQ 3= (5)

where Vacm is the magnitude of the 18kV bus voltage.

It is evident that a control signal (Md or Mq) will affect both

Id and Iq in either of the equations. If however, the control

signal is made to linearise and decouple the currents in

equations (3) and (4) we have [5],[8]:

dcdxsqod VMIM /)(2 += ω (6)

dcqxsdoq VMIM /)(2 +−= ω (7)

4

where Mdx and Mqx are the external inputs for PI control.

Substituting (5-6) in (3-4) we obtain:

acddxsdssd

s VMIRdt

dIL +−−= (8)

acqqxsqs

sq

s VMIRdt

dIL +−−= (9)

In (8-9) it is seen that we have a decoupled controller that

enables independent control along the two channels with no

cross-interactions. In [8], a more advanced version is proposed

with internal predictive control that relies to a lesser degree on

the external current measurements. Although this controller

showed good results in our PSCAD/EMTDC simulation, it

gave inferior performance to the controller presented in

section B below. The weak performance is attributed to the

following:

• The controller requires the variables Id and Iq at a high

bandwidth in order to enable effective linearization. In

practice this is difficult to achieve because of the high

noise content on the AC signals as well as the PLL

dynamics, which affect the accuracy of measurements.

The predictive controller [8] improves responses by

calculating currents internally using the controller

outputs, but in case of external disturbances (Vac changes)

the currents can still have large transient peaks. As

simulation results below show, the Vac disturbances can

be large because of presence of PS converters with pulsed

operation.

• The coefficients (ωo) with the terms that we are canceling are much larger than the terms Rs and Ls. Even

small error (or delay) in the decoupling circuit will cause

large interactions.

• The controller has three control levels just for current

control, and additional two levels are required for DC

voltage and power control which would make it overly

complex.

The above issues limit the magnitude of feedback gains and

this deteriorates performance.

B. Controller based on static equations

In the alternative strategy, we analyse the steady-state

controllability by neglecting the dynamic terms. The CCPC

current and voltage equations are:

( ) ssacms zVVI −= (10)

jVVV sqsds += , ( soss LjRz ω+= ) (11)

Substituting (11) in (10) the converter current components

are obtained as:

( )

( )22sos

sqsossdacm

sdLR

VLRVVI

ω

ω

+

−−= (12)

( )( )( )22

sos

acmsdsosqs

sqLR

VVLVRI

ω

ω

+

−+−= (13)

In the above equations we have cross-coupling, i.e. one

control voltage (Vsd or Vsq) affects both current components.

Assuming now 0=sR , from (12,13) we obtain:

sosqsd LVI ω/−= , ( ) soacmsdsq LVVI ω/−= (14)

Substituting (14) in the basic equations for VSC power (5)

we obtain:

sosqacm LVVP ω/3−= , ( ) sosdacm LVVVQ ω/3 18−= (15)

Equation (15) shows that it is possible to independently

control P or Q by controlling q or d components of the VSC

voltage in the following manner:

Active power control using q voltage component,

Reactive power control using d voltage component.

The interactions are therefore low for small Rs. In addition,

the gain is larger in the desired direction in (15) if Ls is

smaller.

The above-considered ideal case (Rs=0) would lead to

completely decoupled control, implying no interactions

between the two control channels. However, in practice there

will be a small resistance Rs, which depends on the converter

topology, the pulse number, type of switches and the snubber

circuit. For the selected topology in our system a conservative

value for the resistance is Rs=0.24Ω (assuming 650kW losses at the average current of 0.95kA). This resistance value is now

used to analyze the realistic steady-state control and the extent

of the unwanted cross-coupling between the channels.

Assuming a typical operating point at Q=-25Mvar, P=-

2MW (results are similar for other operating points), we

analyse interactions between the channels as shown in Figure

5. In Figure 5 a), which is obtained for a ±20% change in the d-component of the CCPC AC voltage Vsd, it is seen that Q

changes approximately 60% and P changes 10% of its full

range, and therefore the interaction between channels is small.

Ideally P should remain at zero for this input. Similar analysis

is performed with Vsq changed by a factor of 5 (the gain is

lower along q axis), and the results are shown in Figure 5 b).

We observe that the active power P will change 25% of the

full range while reactive power Q changes only by 3-4%. It is

concluded that there is no need to use control compensation

for interactions of such small magnitude.

The above study only covers the steady-state operation,

whereas during transient conditions there will still be cross-

coupling between the control channels as seen in (3) and (4).

The dynamic control analysis is presented in the next section.

P

Q

P

Q

Figure 5. CCPC active and reactive power change a) as Vsd changes by

±20%., b) as Vsq changes by a factor of 5.

5

ki1*1/s

kp1

KPfPload

Qload

d(Qload)/dt

KQf

KQdf

+ +

+

+

+

+

++

Vacmref

feedforward

signal

control signal

components

PS load

measurements

18kV AC voltage controller

Power controller DC voltage controller

Vacm

MdMd Mm

Mq

Mϕ

ki2*1/s

kp2

++

+Vdcref

Vdcrefm

Plref

Vdc

Vdc

Vdc

Pls

Mq

ki3*1/s

P/Vdc

kp3

+

+1/s+

Hzn 180

1

=ω

Hzn 80

1

=ω

Hzn 20

1

=ω

a/b

b

a

a/b

b

a

Figure 6. Converter Controlled Pulse Compensator control system.

V. CONTROLLER STRUCTURE

The controller structure is shown in Figure 6. The

controller consists of two independent units: the AC voltage

controller and the active power controller.

The AC voltage controller is a conventional PI type

controller with a feedback of the 18kV AC voltage magnitude

Vacm. Because of the fast power variations of the PS

accelerator, we use additional direct compensation of the

disturbances by measuring PS reactive power Qload, reactive

power differential dQload/dt and the active power Pload.

These three signals could be directly measured on the AC side,

at the 18kV substation, however the following difficulties

might arise:

The measurement of AC variables has a low bandwidth

since vector transformation is employed and resulting

harmonics must be filtered,

The actual disturbance originates on the DC side of the

load (the DC voltage reference for PS), and measuring

the AC variables gives only a filtered disturbance.

The basic converter equations [9] are used to achieve

estimation of the above AC side variables by measuring the

DC side variables of the PS. Figure 7 outlines the principle of

this estimation, enabling much wider bandwidth feed-forward

signals. The input signals in Figure 7 are the PS converter

firing angle (αps) and the PS DC current (Ipsdc), whereas it is assumed that the AC voltage is constant since DC variables

undergo larger changes.

The active power controller consists of two stages: DC

voltage controller and power controller. The DC voltage

controller is required for two main reasons:

In case of no load, the VSC operates as an AC voltage

controller only, and DC voltage is kept at the constant

minimum value to minimize losses.

The DC voltage controller safeguards the DC voltage

limits and overrides the power controller if these are

violated.

The power controller is of PI type regulating the total

power exchange with the network (Pls). This controller also

includes an additional integrator to enable tracking of DC

voltage ramps.

It should be noted that special anti wind-up feedback

circuits (not shown in Figure 6) are also required to eliminate

winding up of the series connected integrators.

VI. CONTROLLER DESIGN

A. Small signal linear model

In order to analyze the dynamics of the above system and to

determine the controller parameters, a small-signal analytical

model is created. This is a state-space linearized model with

the structure presented in Figure 8. The model consists of two

state-space represented subunits, which are coupled together

using the interaction matrices and variables as presented in

[10]. The AC system model includes the cable and filters, and

the state-space VSC converter modeling follows the modeling

approaches presented in [11]. The load is not included in this

model (it is a disturbance in Figure 8), because of complexity.

Also the feed-forward control gains are not considered; they

are introduced at the latest testing stage in PSCAD/EMTDC.

α α

Ipsdc IpsdcPload

Qload

dQ/dtP=F1(Ipsdc,α ) Q=F2(Ipsdc,α )

ps ps

psps

Figure 7. The principle of estimation of PS active and reactive power by measuring the DC side variables. The AC voltage Vac is assumed constant.

DC voltage

controller

AC voltage

controller

Power

controller

CCPC AC

systemVdcref

VdcMq

Vac

Is

Ips

disturbance

Md

Pls

Plsref

Vacmref

Vacm

DC co-ordinate frame

D-Q rotating

co-ordinate

frame Figure 8. Structure of the simplified linear model.

6

PSCAD

MATLAB

time [s] time [s]

AC voltage [pu]

DC voltage [kV]

a) b)

PSCAD

MATLAB

Figure 9. System model response after a) +5% on AC voltage reference

(Vacmref), and b) -10% step input on DC voltage reference (Vdcref). (The gains are those used in final design with PS load and PSCAD simulation).

The model is coded in MATLAB and it enables the

eigenvalue and frequency domain design techniques, which do

not exist in PSCAD/EMTDC. Figure 9 is the verification test

of the analytical model against non-linear PSCAD/EMTDC

simulation, where very good accuracy is observed.

B. AC voltage and DC voltage control

This section studies the system dynamics using the root

locus method and assuming that only the DC voltage

controller is active on q-control input. In the study, the gains

of one controller are varied whereas the gains on the other

controller(s) are fixed at the optimum values. Also, only the

dominant eigenvalues are observed considering that the model

is of high order.

In all root-locus Figures diamonds represents the open loop

system. The selected location is the optimal value of the gains,

determined in the final simulations in PSCAD/EMTDC.

Studying solely the root locus much larger gains could be

adopted in many cases. However the noise and non-linearities

in the practical system will impose lower limits on the gains.

Figure 10 shows the root locus for the AC voltage

controller, assuming that the controller zero (-ki1/kp1) is pre-

selected. A larger controller zero will move the locus branch

to the left increasing the response speed, but wide-bandwidth

control is difficult in a practical system because of the

converter harmonics. The open loop eigenvalues on branch B

are those selected with q-input controller, as presented later in

Figure 11. It can be observed that the AC voltage controller

affects predominantly the eigenvalue branch A.

Figure 11 shows a similar root-locus analysis with the DC

voltage controller. The open-loop eigenvalues on branch A are

those selected in the design stage in Figure 10. It is seen that

the DC voltage controller moves only the eigenvalues on

branches B and C, where the C branch limits the achievable

speed of response.

From Figures 10 and 11 it is deduced that the two control

channels (along d and q axis) have largely decoupled

dynamics. These are important conclusions since as the

consequence:

• It is possible to use independent and sequential

controller design,

• If one control loop is broken, the other loop is not

much affected.

Figure 11 also shows the root locus when the CCPC

reactance Ls increases 5 times (B’,C’ branches), and evidently

in this case the system is more difficult to control. Therefore a

balance must be achieved since large Ls would be beneficial in

reducing the harmonic level.

Re

Im

C

A

Bselected

location

Figure 10. Root locus for the AC voltage controller. The open loop system

consists of the DC voltage controller.

Re

Im

A

CC’

B’B

selected

location

B’, C’ locus with Ls increased 5 times

Figure 11. Root locus for the DC voltage controller. The open loop system consists of the AC voltage controller.

C. AC voltage and Power controller

This section studies the dynamics in case that a power

controller is used in addition to the DC voltage controller with

the q axis control. During normal operation, the CCPC is

expected to frequently change between DC voltage and power

control modes and therefore both modes should have

satisfactory performance.

Figure 12 shows the root locus for the power controller

where the open loop system has DC voltage and AC voltage

controller gains as selected in Figures 10 and 11. It is seen that

the eigenvalues on branches B and C change position whereas

branch A is very short and very little interactions will result

with the d-axis controller. Comparing with the Figure 11, it is

seen that adding the power control mode has similar effect as

increasing the DC voltage gains.

Figure 13 shows the AC voltage controller root locus with

the power controller gains from Figure 12. There is only

marginal interaction with d-axis control as evident by the short

branch C. It is also seen that AC voltage control has faster

dynamics than the DC voltage control (as also confirmed in

Figure 9), which is the result of the large DC capacitance with

CCPC. The final controller gains are obtained by tuning in

PSCAD/EMTDC and their values are shown in the Appendix.

Note that the feed-forward gains have relatively small values

and minimal influence on the system dynamics, however they

contribute reducing the AC voltage peaks during PS cycles.

The converter losses Rs in our model cannot be accurately

determined. Since this parameter depends also on the

converter topology it would be meaningful to investigate

7

Re

Im

A B

C

selected

location

Figure 12. Root locus for the power controller. The open loop system consists of the DC voltage controller and AC voltage controller.

Im

C’ C

A

A’ B’

B

Re

A’,B’,C’ locus with 3 times increased Rs

Figure 13. Root locus for the AC voltage controller. The open loop system

consists of the DC voltage controller and the power controller.

the dynamic cross-coupling for different Rs. Figure 13 also

shows the AC voltage controller root locus with converter

resistance increased by a factor of 3. It is observed that branch

B’ is significantly longer implying stronger interactions

between the two control channels. In addition, the root locus

for the AC voltage controller (A’) moves towards Im axis and

it is less favorable in terms of stability and performance. This

indicates potential control problems with high-loss converter

concepts or with higher switching frequencies.

In our studies the following factors are found to negatively

influence the system stability:

• Increased converter losses Rs,

• Increased connecting reactance Ls,

• Reduced converter DC capacitance Cs.

• The power control mode shows slightly more

interactions than DC voltage control.

In case of the above system parameters which result in

more control interactions, the decoupling method from section

IV A) might offer the best performance. It is also mentioned

that even wide variations in the AC system impedance show

only minimal changes in the system dynamics.

VII. SIMULATION RESULTS

Figure 14 shows the PSCAD/EMTDC simulation of a

single PS cycle (pulse 1) when compensated with CCPC. The

first part of the simulation (between 0.7s and 1.9s), is the

capacitor charging period which is shown for completeness.

Figure 14a) shows that the active power exchange with the

network (Pls) remains approximately constant at 7MW. Note

that the power reference of 7MW accounts for the PS

accelerator and converter losses over the particular cycle. For

ideal compensation it should be adjusted individually for each

PS cycle. The reactive power exchange (Qls) is almost zero.

In Figure 14b) we observe excellent control of the AC

voltage. There is a 2.5% peak-to-peak deviation at the instant

of power reversal, which is very difficult to eliminate.

-45-40-35-30-25-20-15-10-505101520253035404550

0.7 0.9 1.1 1.3 1.5 1.7 1.9 2.1 2.3 2.5 2.7 2.9 3.1 3.3 3.5 3.7 3.9

Time [s]

Active Power [MW]

P -PSCAD

Pls Plsref

Charging period Pulse operation

P -PSCAD

Plsref

Pls

Qls

a) Active and reactive power exchange.

0.98

0.985

0.99

0.995

1

1.005

1.01

1.015

1.02

0.7 0.9 1.1 1.3 1.5 1.7 1.9 2.1 2.3 2.5 2.7 2.9 3.1 3.3 3.5 3.7 3.9

Time [s]

AC Voltage [pu]

Vac rms

Vref

b) 18kV AC voltage.

-14

-12

-10

-8

-6

-4

-2

0

2

4

6

8

10

12

14

16

18

0.7 0.9 1.1 1.3 1.5 1.7 1.9 2.1 2.3 2.5 2.7 2.9 3.1 3.3 3.5 3.7 3.9

Time [s]

Control angle [deg]

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1

1.1

Control magnitude

control angle

control magnitude

c) Converter control variables (control magnitude Mm, control angle Mϕ).

25

30

35

40

45

50

55

60

65

0.7 0.9 1.1 1.3 1.5 1.7 1.9 2.1 2.3 2.5 2.7 2.9 3.1 3.3 3.5 3.7 3.9

Time [s]

DC Voltage [kV]

0.8

0.9

1

1.1

1.2

1.3

1.4

1.5

1.6

current [kA]

CCPC AC current

CCPC DC voltage

Vdc ref

d) CCPC DC voltage and AC current. Figure 14. PSCAD simulation responses for a single PS cycle assuming

compensation with CCPC and 580 capacitors.

8

-45-40-35-30-25-20-15-10-50510152025303540455055

1.2 1.6 2 2.4 2.8 3.2 3.6 4 4.4

Time [s]

Power [MW]

P -PSCAD

Pls

Plsref

Figure 15. PSCAD simulation responses for two PS cycles assuming

compensation with CCPC and 430 capacitors.

Figure 14c) is the simulation of CCPC control variables

Mm and Mϕ. It is important to observe that Mm stays close to, but it does not exceed the saturation limit of Mm=1,

confirming that an appropriate design approach is used.

The compensator DC voltage profile is shown in Figure

14d). At the beginning of the cycle the capacitors are fully

charged at 60kV. The DC voltage is decreasing when CCPC

delivers energy to the network and increasing when absorbing

energy. At the end of the positive half-cycle the DC voltage is

close to the minimum limit Vdcmin=36kV indicating optimum

component sizing. The transitions between active power and

DC voltage control are without noticeable overshootings.

CERN has also indicated potential benefits in using a

compensator with reduced storage capacity. In this case the

DC capacitor rating is determined solely by the amount of

energy returned to the network during the negative half-cycle.

Using (1-2) it is found that 430 capacitor units would be

required to enable storage of approximately 11MJ during the

negative half-cycle. Figure 15 shows that there is an

incomplete compensation during the positive half-cycle but it

is possible to fully recover the energy returned during the

negative half-cycle of the PS pulse. This figure demonstrates

that a compensator with reduced energy storage capacity

would also be a feasible technical option.

Note that the CCPC compensator element could be used

with renewable sources and a range of other industrial loads

like: traction drives, large induction motors or arc furnaces.

VIII. CONCLUSIONS

CERN’s Proton Synchrotron particle accelerator demands

very short and steep pulses of active and reactive power that

have a negative impact on the network power quality. A

compensating device, Converter Controlled Pulse

Compensator, consisting of a parallel connected VSC

converter with large DC capacitors, is found to be a suitable

compensation option.

The controllability analysis shows that active power control

can be achieved using q-axis of the PWM converter input, and

AC voltage control is obtained through d-axis converter input.

By analyzing the eigenvalue location it is shown that the

dynamic interactions between the two control channels are

small and large gains can be employed with simple PI

controllers. Further studies with wider range of parameters

show that the above results are primarily applicable to systems

with a large DC capacitance and low converter losses. The

PSCAD/EMTDC simulations of the PS accelerator cycles

show that reactive power compensation with CCPC achieves

excellent AC voltage control and the energy storage capability

enables that the active power exchange with the network

remains constant throughout the entire load cycle.

IX. REFERENCES

[1] K. Kahle. J. Pedersen, T. Larsson, M. de Oliveira, “The new 150 Mvar, 18 kV Static Var Compensator at CERN: Background, Design and

Commissioning,” CIRED 2003 [2] N. G. Hingorani, L. Gyugyi: “Understanding FACTS: Concepts and

Technology of Flexible AC Transmission Systems,” IEEE Press, 2000

[3] J. N. Baker, A. Collinson, “Electrical Energy storage at the turn of the millennium,” Power Engineering Journal Volume: 13, Issue: 3, pp: 107

– 112, June 1999.

[4] A. B. Arsoy, Y. L.iu, P.F.Ribeiro, F.Wang,: “StatCom – SMES,” Industry Applications Magazine, IEEE ,Volume: 9 ,Issue: 2, pp:21 – 28,

March-April 2003.

[5] C. Shen, Z. Yang; M. L. Crow, S. Atcitty: “Control of STATCOM with energy storage device,” Power Engineering Society Winter Meeting,

2000. IEEE , Volume: 4, pp:2722 – 2728, 23-27 Jan. 2000

[6] Manitoba HVDC Research Center “PSCAD/EMTDC users manual,” Winnipeg 2003.

[7] N. Mohan, T. M. Undeland, W. P. Robbins “Power Electronics Converters, Applications and Design,” John Wiley & Sons, 1995

[8] Papic, I.; Zunko, P.; Povh, D.; Weinhold, M.; “Basic control of unified power flow controller,” IEEE Transactions on Power Systems,

Vol. 12, no 4, Nov. 1997 Pp: 1734 - 1739. [9] P. Kundur: “Power System Stability and Control,” McGraw Hill, Inc.

1994.

[10] D. Jovcic N. Pahalawaththa, M. Zavahir “Analytical Modelling of HVDC Systems.” IEEE Trans. on PD, Vol. 14, No 2, pp. 506-511, April

1999

[11] D. Jovcic L. A. Lamont, L. Xu: “VSC Transmission model for analytical studies," Power Engineering Society General Meeting, 2003, IEEE,

Volume: 3, pp:1737 – 1742, 13-17 July 2003.

X. APPENDIX

CCPC CONTROLLER GAINS

Comment Gain value

Kp2 1.4 [1/kV] DC voltage control

Ti2 0.02 [skV]

Kp3 1 [1/kV] Power control

Ti3 0.04 [skV]

Kp1 45 [1/kV] AC voltage control

Ti1 0.00014 [skV]

KPf 0.1 [1/MW]

KQf 1.3 [1/MVA]

Feedforward control

KQdiff 0.01 [s/MVA]

XI. BIOGRAPHIES

Dragan Jovcic (S’97, M’00) obtained a B.Sc. in Control Engineering from

the University of Belgrade, Yugoslavia in 1993 and a Ph.D. degree in Electrical Engineering from the University of Auckland, New Zealand in

1999. He is currently a lecturer with the University of Aberdeen, Scotland where

he has been since 2004. He also worked as a lecturer with University of

Ulster, in period 2000-2004 and as a design Engineer in the New Zealand power industry in period 1999-2000. His research interests lie in the areas of

FACTS, HVDC and control systems.

Karsten Kahle studied Electrical Power Engineering at the FHTW Berlin, Germany, and received his M.Sc. degree from the University of Manchester,

UK, in 1995. He is currently studying part-time towards his Ph.D. degree at

the University of Vienna, Austria. K. Kahle jointed the Electric Power Systems Group of CERN in 1997,

where he is now responsible for project management, system design and

power system analysis. His main interest is on the application of FACTS for particle accelerators.