TABLE 1 Theoretical and Measured Results of a DRA with r, = 14.2, d = 3.0 mm, and a = 9.5 mm

Parameter Cylindrical DRA

at HE21 s

Resonant frequency

Antenna gain

VSWR I 2.5 Bandwidth

Q factor

Radiation efficiency

Theory Measured

Theory Measured

Measured

Me as u r e d

Theory Me as u r e d

7.785 GHz 8.025 GHz

4.8 4.4

6.3%

15.1

100% 90%

for both planes and the discrepancy observed at small eleva- tion angles, particularly for EB, is attributed to the finite size of the ground plane. The difference between the copolarized and cross-polarized levels was greater than 25 dB.

The antenna gain was measured with the use of the same receiving antenna as for the radiation patterns, for several distances away from the DRA, and the value obtained was 4.4, against the predicted value of 4.8. The bandwidth of the DRA for a VSWR I 2.5 was measured to be 6.3%, indicating a Q factor of 15.1. The radiation efficiency of the DRA, measured with the use of the Wheeler cap method [15, 161 with a metallic box of 25 (height) X 50 (width) X 50 (length) mm3, was 90%. The theoretical and experimental results obtained are tabulated in Table 1.

4. CONCLUSIONS

A theoretical formulation together with experimental investi- gation of an unmagnetized ferrite DRA operating at the HEZls mode have been presented. Expressions for the elec- tric field components in the far field have been derived. Radiation patterns and the dependence of directivity on the dimensions of the DRA have been illustrated. The results indicate that the radiation pattern is more directional for small aspect ratios and that the directivity decreases as the height decreases and as the radius of the cylinder increases.

Experimental results showed that the magnetic-wall model employed gives good prediction in resonant frequency, radia- tion pattern, and directivity. More accurate prediction in these parameters would require the use of numerical meth- ods. The gain of the DRA is 30% greater than the gain of an ideal monopole and the bandwidth is about four times greater than the bandwidth of a patch antenna. The radiation perfor- mance of a cylindrical DRA operating at the HEzls mode is superior to the performance of a monopole or patch antenna and hence, makes it a better choice for many antenna re- quirements.

REFERENCES S. A. Long, M. W. McAllister, and L. C. Shen, “The Resonant Cylindrical Dielectric Cavity Antenna,” IEEE Trans. Antennas Propagat., Vol. AP-31, May 1983, pp. 406-412. M. W. McAllister, S. A. Long, and G. L. Conway, “Rectangular Dielectric Resonator Antenna,” Electron. Lett., Vol. 19, March

M. W. McAllister and S. A. Long, “Resonant Hemispherical Dielectric Antenna,” Electron. Lett., Vol. 20, Aug. 1984, pp.

1983, pp. 218-219.

657-659.

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16.

R. K. Mongia, A. Ittipiboon, P. Bhartia, and M. Cuhaci, “Electric Monopole Antenna Using a Dielectric Ring Resonator,” Elec- tron. Lett., Vol. 29, Aug. 1993, pp. 1530-1531. R. K. Mongia and P. Bhartia, “Dielectric Resonator Antennas-A Review and General Design Relations for Reso- nant Frequency and Bandwidth,” Int. J. Microwave Millimetre- WaLle Cornput.-Aided Eng., Vol. 4, 1994, pp. 230-247. J. T. H. St. Martin, Y. M. M. Antar, A. A. Kishk, A. Ittipiboon. and M. Cuhaci, “Dielectric Resonator Antenna Using Aperture Coupling,” Electron. Lett., Vol. 26, Nov. 1990, pp. 2015-2016. R. A. Kranenburg and S. A. Long, “Microstrip Transmission Line Excitation of Dielectric Resonator Antennas,” Electron. Lett., Vol. 24, Sept. 1988, pp. 1156-1157. R. A. Kranenburg, S. A. Long, and J. T. Williams, “Coplanar Waveguide Excitation of Dielectric Resonator Antennas,” IEEE Trans. Antennas Propagat., Vol. AP-39, Jan. 1991, pp. 119-122. Y. Kobayashi and S. Tanaka, “Resonant Modes of a Dielectric Rod Resonator Short-circuited at Both Ends by Parallel Con- ducting Plates,” IEEE Trans. Microwacje Theory Tech., Vol. MTT-

D. Kajfez, A. W. Glisson, and J. James, “Computed Modal Field Distribution for Isolated Dielectric Resonators,” IEEE Trans. Microwaoe Theoly Tech., Vol. M’IT-32, Dec. 1Y84, pp. 1609-1616. R. De Smedt, “Correction due to a Finite Permittivity for a Ring Resonator in Free Space,” IEEE Trans. MicrowaLie Theory Tech.,

R. N. Simons and R. Q. Lee, “Effect of Parasitic Dielectric Resonators on CPW/Aperture-Coupled Dielectric Resonator Antennas,” IEE Proc. Pt. H , Vol. 140, Oct. 1993, pp. 336-338. C. A. Balanis, Advanced Engineering Electromagnetics, Wiley, New York, 1989. J. D. Kraus, Antennas (2nd ed.), McGraw-Hill. New York, 1988. D. M. Pozar and B. Kaufman, “Comparison of Three Methods for the Measurement of Printed Antenna Efficiency,” IEEE Trans. Antennas Propagat., Vol. AP-36, Jan. 1988, pp. 36-39. H. A. Wheeler, “The Radian Sphere around a Small Antenna,” I R E Proc., Aug. 1959, pp. 1325-1331.

28, Oct. 1980, pp. 1077-1085.

Vol. MTT-32, Oct. 1984, pp. 1288-1293.

Receioed 6-26-96

Microwave and Optical Technology Letters, 13/5, 248-251 0 1996 John Wiley & Sons, Inc. CCC 0895-2477/96

COMPACT CIRCULAR WAVEGUIDE MODE CONVERTERS Tanveer UI Haq COMSEARCH 2002 Edrnund Halley Drive Reston, VA 201 91

Kevin J. Webb School of Electrical and Computer Engineering Purdue University West Lafayette, IN 47907

Neal C. Gallagher Department of Electrical Engineering University of Delaware Newark, DE 19716

KEY TERMS Circular waueguides, scattering optimization, mode con Lietters

MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 13, No. 5. December 5 1996 251

ABSTRACT The scattering opiimization meihod is utilized to design compact circular waveguide mode concerters, which are compared with some earlier designs. This approach is used to design a TE,, to TM,, mode conL'erter at 9.94 GHz, which was subsequently fabricated with easily ai,ailable components. At this frequenq, the purity ofthe TM,, mode at the output of the mode coni'efler was measured to be 98.1%, which is cery close to the theoreiical i,alue of 99.5%. 0 1996 John Wilq & Sons. Inc.

1. INTRODUCTION

A scattering optimization method (SOM) for the design of waveguide mode converters was introduced in [ l , 21 for the conceptually simple case of parallel plate waveguides. There are, however, many important applications of mode convert- ers in overmoded circular waveguides [3, 4).

This article presents a procedure and examples for the design of compact mode converters for circular waveguides. Results are provided for a T E , , to T M , , mode converter, which was designed with the SOM, fabricated. and tested for performance.

II. THE SCAlTERING OPTIMIZATION METHOD FOR CIRCULAR WAVEGUIDES

The SOM used for the design of circular waveguides is essentially the same as in [l], except that the variables of optimization are the step radii rather than step heights, as in the case of parallel plate waveguides, and that step protru- sions are used. The design parameters that must be selected for designing a circular waveguide mode converter are AL,,,. the minimum step length, which is decided on the basis of fabrication accuracy: 1. the initial step length (a multiple of AL,,,): L , the initial length of the scatterer (a multiple of I ) ; A r , the optimization radius grid size; rmm, the maximum allowed radius of protrusion outside the waveguide wall. Some discussion on the criteria for the selection of these design parameters is provided in [S].

In the forward solution of the SOM, mode matching is used to solve for the circular waveguide step discontinuities [ 11. Normalized mode functions for circular waveguide, as given in [6], are utilized in this formalism. To confirm the validity and accuracy of the mode matching code, the exam- ples given by English [7], Carin [8], and Nagelberg and Shefer [9] were simulated.

6

4

E 2 C . -

-2 v 0 2 Mode converter length 6 in cm 8 10 12

Figure 1 GHz

Profile of a TEO2 to TE,, mode converter designed for 28

3

2

5 1 .- v1

L

.- .* 2-1 d

-2

-3 I I ' ' I I

Mode converter length in cm

Figure 2 Profile of a TEob to TEo, mode converter designed for 140 GHz

With the use of the SOM, mode converters were designed for practical applications for which earlier designs exist. One of the earlier designs, reported by Otsuka, Shimizu, and Nishi [lo], is a 28-GHz TE,, to TE,, converter with a length of 83.4 cm and an efficiency of 97%. The SOM mode converter for this application, as shown in Figure 1, has rmax = 6 cm, AL, , , = 2.5 mm, A r = 0.1 mm, a length of 10 cm, and an efficiency of 98.71%. Another mode converter reported by KumriC, Thumm, and Wilhelm [3], for TE,, to TE,, conver- sion at 140 GHz, had a length of 179.5 cm and an efficiency of 98.3%. The corresponding SOM mode converter, as shown in Figure 2, has ALmln = 2 mm, rmax = 3.5 cm, A r = 0.1 mm, a length of 37.2 cm, and an efficiency of 96.8%.

It is evident that the SOM can produce shorter mode converters with similar efficiencies. The SOM solution can also be selected based upon bandwidth constraints, which is essentially a measure of the shape of the local optimization surface.

111. TE,, TO TM,, MODE CONVERTER FOR CIRCULAR WAVEGUIDE

To demonstrate the practical feasibility of the SOM, a TE,, to TM,, mode converter was designed, fabricated, and tested; the dominant circular waveguide mode is to be converted to TM,,, the next propagating mode with the same azimuthal symmetry. A mode converter for conversion between these two modes is practical [ l l ] and relatively straightforward to implement and characterize. A design frequency of 9.94 GHz was chosen for convenience.

An extended center-pin SMA launcher mounted on the sidewall of the input circular waveguide was used to excite a TE,, mode in the input waveguide. By selecting a waveguide radius that allows only the TE,, mode to propagate, and a length such that the evanescent TM,, mode has a negligible amplitude at the mode converter input plane, a pure TE,, excitation was achieved. The output waveguide, however, is selected to be larger, allowing both the TE,, and the TM,, modes to propagate.

To keep the cost of production low a standard copper tubing of 0.875-in. outer diameter (O.D.), with a wall thick- ness of 0.032 in., was used for the input waveguide, and a brass tubing of 1.75-in. O.D. and a wall thickness of 0.063 in.

252 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 13, No. 5, December 5 1996

was used for the output waveguide. This commercially avail- able tubing is prone to slight fabrication errors. The average inner radius of the input waveguide is 1.012 cm, which gives a cutoff frequency of 8.76 GHz for the TE,, mode and a cutoff of 18.0 GHz for the TM,, mode. The output waveguide average inner radius is 2.0625 cm, which sets a cutoff of 8.87 GHz for the TM,, mode.

To develop the staircase structure of the mode converter, it was decided to fabricate 0.2S-in.-thick aluminum discs with holes of appropriate radii. These discs, when stacked to- gether, would form the mode converter structure. The com- mercially available sheet had variations in its thickness. After grinding, the average thickness came close to 0.2505 in. In our design procedure, therefore, A L m i n was selected to be 0.2505 in. (0.6363 cm). A piece of 4.5-in. O.D. aluminum tubing with a wall thickness of 0.032 in. was selected for housing the component discs of the mode converter to facili- tate concentric alignment. The inner radius for this tube is 5.64 cm. Keeping a safe margin, rmax for the mode converter design was selected to be 4 cm. This is a case where both ALmin and rmax have been selected purely due to physical and material constraints. The other design parameters se- lected for the mode converter are L = 1.002 in. and 1 =

0.2505 in. The increment used for step radii during optimization, Ar,

is critical in establishing the extent of errors that can be tolerated in the fabrication process. To have a sufficient margin of error, it was decided to accommodate fabrication errors of up to f0.004 in. Various mode converters were designed, starting with different initial guesses and different values of Ar. A fabrication error analysis was carried out for each one of them. Two of these designs which use the same value for A r are shown in Figure 3. Table 1 shows a compari- son of the two designs. The first of these designs was selected for fabrication. It has 10 steps in the staircase model with a total length of approximately 2.5 in. and has a conversion efficiency of 99.5% at the design frequency of 9.94 GHz. Although 30 evanescent modes were used in each waveguide section for the forward solution, somewhat fewer give a satisfactory solution.

The mode converter was fabricated and assembled, as shown in Figure 4. Figure 5 shows a photograph of the complete unit.

IV. CHARACTERIZATION Various methods have been reported in the literature for characterizing mode converters for overmoded waveguides. Examples are k-spectrometer near-field measurements [ 121, slotted line measurements [9], and far-field radiation mea- surements [13]. The first two methods require a dual-mode matched load at the output of the mode converter and require holes or slots to be machined in the waveguide wall. The k-spectrometer method also requires special fixtures for precise angular measurements. Limitations of the slotted line measurement method are that when there are more than two propagating modes, it does not give a unique solution and is difficult to apply. For the nonstandard waveguide used in our design, a dual-mode matched load was not commercially available. Also, with the electrically small waveguides used, holes or slots in the waveguide wall could alter the distribu- tion of modes. The k-spectrometer and slotted line were therefore not used for the mode content measurement. A modified version of the slotted line method was, however, used to obtain approximate values for the mode percentages

Design 1 2

1.5- I

1.5

'-'I -0.5 0 0.5 1:5 2.5 315 Converter length in inches

Design 2 2

Converter length in inches

Figure 3 Two designs for the TE,, to TM,, mode converter. The top design was fabricated

TABLE 1 Comparison of the Two Mode Converter Designs of Figure 3

Parameter Design 1 Design 2

Length 6.363 cm (2.5 in.) 7.64 cm (3.0 in.)

A r 0.3 mm 0.3 mm Fabrication tolerance in step heights for +0.15 mm kO.1 mm 1 % efficiency change 1% Bandwidth 35 MHz 20 MHz

Efficiency 99.5% 99.37%

in the output waveguide by placing a shorting plane at the end of the output waveguide instead of a matched load.

The method used to accurately characterize the mode converter output was the far-field radiation pattern measure- ment. In this method it is assumed that the modes in the waveguide are perfectly coupled into free space and that there is negligible reflection and cross coupling of modes at the open end of the waveguide. To ensure this condition, a dual-mode horn antenna was fabricated [ 111. The antenna was constructed by folding a 0.02-in.-thick copper sheet into a cone. The cone angle a! of the horn was 6.2.5", and the exit radius of the horn was 2.72 in. A transition section to reduce the discontinuity between the uniform waveguide and the horn was machined. The relative mode content in the output

MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 13, No. 5, December 5 1996 253

I 1-10 0 2 5 a ’ m I

Figure 4 Sectional view of the TE, to T M I , mode converter

Figure 5 Photograph of the fabricated mode converter

waveguide was then established by measuring the far-field pattern of thc horn antenna for various frequencies in an indoor antenna range. The measured percentage power of the T M , , mode in the output waveguide is plotted against frequency in Figure 6. The solid line indicates the calculated conversion efficiency. At 9.94 GHz, a measured T M , , content of Y8.1% was obtained.

For the reflection cocfticient measurement at the input of the mode converter, a HP8510 vector network analyzer was used. This instrument allows reflection coefficient measure- ments in single-moded waveguides [ 141. The input waveguide of thc mode converter has a nonstandard dimension for which no calibration standards are available. A calibration kit was therefore fabricated by machining three offset shorts to enable a one-port calibration of the HP8510. Other stan- dards, like an open and a matched load, however, were not available to verify the calibration. The magnitudc of the reflection coefficient at the input plane of the mode con- verter was measured to be 0.35 at 9.94 GHz, which is high compared to the theoretical value of 0.058. The numerical design procedure accounts for all reflected and transmitted power in various modes. Given the success of the radiation measurement, which was verified for the case of a single waveguide step with our simulations and those of English [7], we suspect mechanical problems for this disparity between the theoretical and measured values. Possible reasons for this

/- x Measured - ’ IhWret lCd / , , , , , , ,

5 9 a8 9.9 9.92 9.94 9.96 99s 10 D2

Frequency in GHz

Figure 6 Percentage power in the TM, , mode at the output versus frequency

could be errors in the calibration standards, misalignment of the input waveguide, or reflections from the exit horn.

V. CONCLUSION The scattering optimization method has been successfully implemented to design compact mode converters for circular waveguide applications. For a practical demonstration of the method, a TE,, to TM, , mode converter was designed and built with commercially available components. A far-field radiation pattern measurement indicates a 98.1% pure TM,, mode in the output waveguide for a center frequency of 9.94 GHz. The input and output waveguides in this example have different radii. This example, therefore, not only demon- strates a mode converter but also a waveguide transition that has the flexibility of launching a specific mode in the output waveguide.

ACKNOWLEDGMENTS We thank C. D. Harrington, who is in charge of the Mechani- cal Workshop of the School of Electrical and Computer Engineering, Purdue University, for his innovative ideas and for so precisely building the mode converter. We are also thankful to ALCOA of Lafayette, IN, who donated pieces of aluminum tubing used in the construction of the mode converter.

REFERENCES 1. T. U. Haq, K. J. Webb, and N. C. Gallagher, “Scattering Opti-

mization Method for the Design of Compact Mode Converters for Waveguides,” IEEE Trans. Microwaoe Theory Tech., Vol. MTI-43, March 1995, pp. 559-565.

2. T. U. Haq, K. J. Webb, and N. C . Gallagher, “Scattering Opti- mization Synthesis of Compact Mode Converters for Wave- guides,” IEEE Antennas Propag. Int. Symp. Digest, June 1994.

3. K. KumriC, M. Thumm, and R. Wilhelm, “Optimization of Mode Converters for Generating the Fundamental TEo, Mode from TE, Gyrotron Output at 140 GHz,” Int. J. E/ec/ron., Vol. 64. Jan. 1988, pp. 77-94.

4. M. J. Buckley, D. A. Stein, and R. J. Vernon, “A Single Period TEoI-TE,, Mode Converter in a Highly Overmoded Circular Waveguide.” IEEE Trans. Microwaoe Theory Tech., Vol. 39, pp. 1301-1306, Aug. 1991.

254 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol 13, No 5, December 5 1996

5.

6.

7.

8.

9.

10.

11.

12.

13.

14.

T. U. Haq, K. J. Webb, and N. C. Gallagher, “Aperiodic Grating Structure for TE,,-TM,, Mode Conversion,” Proc. SPIE, Vol. 2622, 1995, pp. 207-214. N. Marcuvitz (Ed.), Waceguide Handbook, Boston Technical Pub- lishers, Boston, 1964, pp. 66-71. W. J. English, “The Circular Waveguide Step-Discontinuity Mode Transducer,” IEEE Trans. Microwaiie Theory Tech., Oct. 1973, pp.

L. Carin, “Computational Analysis of Cascaded Coaxial and Circular Waveguide Discontinuities,” Master’s thesis, University of Maryland, College Park, MD, 1986. E. R. Nagelberg and J. Shefer, “Mode Conversion in Circular Waveguides,” Bell Syst. Tech. J . , Sept. 1965, pp. 1321-1338. M. Otsuka, M. Shimizu, and M. Nishi, “Development of Mode Converters for 28-GHz Electron Cyclotron Heating System,” Int. J. Electron., Vol. 70, 1991, pp. 989-1004. P. D. Potter, “A New Horn Antenna with Suppressed Sidelobes and Equal Beamwidths,” Microwace J . , Vol. 6, June 1963, pp.

W. Kasparek and G. A. Muller, “The Wavenumber Spectro- meter-An Alternative to Directional Coupler for Multimode Analysis in Oversized Waveguides,” Int. J. Electron., Vol. 64, May

Z . X. Zhang, G. Janzen, G. Miiller, P. G. Schuller, M. Thumm, R. Wilhelm, and V. Erckmann, “Mode Analysis of Gyrotron Radiation by Far Field Measurements,” in Con5 Digest, 8th Int. Con5 on Infrared and Millimeter Waiies, Miami Beach, FL, 1983. Product Note 8510-5, Specifying Calibration Standards for the HP 8510 Network Analyzer, Hewlett Packard, March 1986.

633-636.

71-78.

1988, pp. 5-20.

Received 6-26-96

Microwave and Optical Technology Letters, 13/5, 2.5 1-255 0 1996 John Wiley & Sons, Inc. CCC 0895-2477/96

EXPERIMENTAL RESULTS ON A

SUPERCONDUCTING MICROSTRIP ANTENNA

38-GHZ HIGH-TEMPERATU RE

X. Castel Laboratoire de Chimie du Solide et lnorganique Mol6culaire URNA CNRS 1495 Universite de Rennes 1 35042 Rennes Cedex, France

J. M. Floch Laboratoire “Structures Rayonnantes” URNA CNRS 835 Universite de Rennes 1 35204 Rennes Cedex, France

M. Guilloux-Wry Laboratoire de Chimie du Solide et lnorganique Moleculaire URA CNRS 1495 Universite de Rennes 1 35042 Rennes Cedex, France

K. Mahdjoubi and C. Terret Laboratoire “Structures Rayonnantes” URA CNRS 834 Universite de Rennes 1 35042 Rennes Cedex, France

A. Perrin Laboratoire de Chimie du Solide et lnorganique Moleculaire URNA CNRS 1495 Universit6 de Rennes 1 35042 Rennes Cedex. France

KEY TERMS YBa,Cu,O,. MgO, superconducting microstrip antenna, Ka band

ABSTRACT We have designed and fabricated a superconducting microstrip antenna operating at 38 GHz after we patterned a YBa,Cu,O, thin film epitaxi- ally grown on (100) MgO substrate by laser ablation. The experimental performance of this superconducting antenna was measured in a home- made vacuum setup and compared to a similar silver antenna. Measure- ments show eficiency improvement of the superconducting antenna oc’er the silver one of 1.2 dB at 80 K. 0 1996 John Wilq & Sons, Inc.

1. INTRODUCTION

High-T, superconductors (HTS) are very promising for appli- cations to passive devices. Experimental investigations such as delay lines, resonators, and filters at X-band frequencies have shown that HTS films provide a substantial loss reduc- tion over identical circuits fabricated from normal metals (silver, gold, or copper) [ 11. In fact, superconductors have three properties that make them especially attractive for microwave circuits:

1. Their low surface resistance results in very low propa- gation loss in transmission structures and very high quality factors (Qs) in resonant ones.

2. The frequency-independent magnetic penetration depth of superconductors can be exploited to make disper- sionless transmission structures [2].

3. It is possible to control the resistive or reactive proper- ties of thin films by a superimposed current or with laser radiation [ 3 ] .

Much research has been carried out in the field of mi- crowave applications of superconducting electronics. Super- conducting materials such as TlBaCaCuO [4] and especially YBa,Cu,O, films grown on different substrates have been used [5]. It was expected that for frequencies below 100 GHz, passive microwave components with much higher electrical performances and with considerably reduced geometrical size could be realized [6]. HTS materials can improve antenna elements by reducing the losses in both the feed network and the radiating elements themselves. The prospects for the use of high-T, superconductors to miniaturize antenna systems have been theoretically evaluated [7, 81.

We have investigated experimentally and compared the properties of HTS and normal metal in Ka band with a single-element microstrip patch antenna on a (100) MgO substrate. We report here on our homemade measurement setup at 80 K and 38 GHz. Measurements from the antennas, including input impedance, gain, and efficiencies are pre- sented.

2. HIGH-TEMPERATURE SUPERCONDUCTING FILM

2.1. Deposition Procedure. The YBa,Cu,O, thin film was in situ deposited by pulsed laser ablation from a homemade stoichiometric target (density > 90%), as previously reported [9]. The output from an excimer laser (SOPRA SEL 520) operating with XeCl ( A = 308 nm) was focused onto a rotat- ing YBa,Cu,O, target, achieving an energy density of = 3 J/cm2 with a repetition rate of 2 Hz. The distance between the substrate holder and the target was kept at 43 mm. The 340-nm-thick YBa,Cu,O, thin film used in this work was deposited on a 10 x 10 X 0.5-mm3 (100) MgO single-crystal substrate (supplied by Sumitomo Inc.) at 740 “C at an oxygen

MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 13, No. 5, December 5 1996 255