AN2321 - ww1.microchip.com

AN2321Solar MPPT Battery Charger for the Rural Electrification System

INTRODUCTION

Solar chargers are increasingly gaining momentumwith government agencies pushing towards a greenersolution through the use of energy derived fromrenewable sources. A solar charger mainly functionson the principle of harnessing the energy from the sunand utilizing it to supply electrical energy to devices orfor charging batteries.

Although the solar charger industry has been plaguedby many companies manufacturing solar chargers,most of these are based on the concept of traditionalgrid infrastructure with permanently installed units.Very few have ventured into portable solar units. Suchportable solar units become very handy when it comesto distributed energy solutions, especially in developingcountries. Portable solar power provides an opportunityfor rural areas in developing countries to skip thetraditional grid infrastructure and move directly todistributed solutions. For off-peak usage during thenight, battery banks can be used to store energy. Inaddition, during the day these solar chargers cansupplement the main power supply, thereby yieldingbetter energy savings.

This application note is aimed at approaching such arural market by providing a solution in the form of aportable solar charging system. The details of thehardware design, manipulation of the power stages,implementation of Maximum Power Point Tracking(MPPT) and battery management will be the keyhighlights covered in this application note. It alsocovers explanation of the design of compensators forthe various power converters using a PIC®

microcontroller with its superior class of CoreIndependent Peripherals (CIPs).

SYSTEM SPECIFICATIONS

Power Budget

Endurance

A fully charged system can operate for two days atmaximum load without charging.

The system can operate lights for three weeks on asingle 10-hour charge, not including a 12V chargingload.

Charge Distribution within the System

1. During sunlight, the system will efficientlytransfer the maximum power from the solarpanel to the load, with any extra charge routedto the battery for charging.

2. If the battery is fully charged, then excesscharge is left in the solar panel.

3. In marginal sunlight, power from the solar panelcan be augmented by power from the battery.

4. Without sunlight, the load will be efficientlypowered from the battery.

The system will also include a fail-safe system toprevent damage in the event of incorrect connection tothe solar panel, battery or loads. The system will sendstatus and alarm conditions via a low-power Bluetooth®

interface compatible with laptop and cell phones.

Authors: Namrata DalviSwathi SridharAshutosh TiwariMicrochip Technology Inc. TABLE 1: SYSTEM POWER BUDGET

System Power Use

Solar panel

130W in full sun

Provide system with 1.3 kWh charge in 10 hours

Battery Two 12V@55AHr

Storage capacity for 1.3 kWh of charge

Lighting 2x5W@6hrs 60 Wh (assumes 6 hours of light)

12V@2A 24W 576 Wh (assumes 24-hour usage)

2016 Microchip Technology Inc. DS00002321A-page 1

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BASIC PRINCIPLE

The voltage current and power produced by a solarpanel are highly variable in response to ambientconditions and dramatically dependent on the electricalimpedance of the imposed load (V vs. I). Under anycombination of ambient conditions, it is characterizedby exactly one ideal load impedance, which will resultin operation at VMPPT and maximum power transfer.

Maximum Power Point Tracking (MPPT) is anelectronic system that operates the solar photovoltaic(PV) modules in a manner that allows them to producethe maximum power they are capable of. MPPT isdifferent from a mechanical tracking system (paneltracking) that physically “moves” the modules to makethem point directly at the sun. Maximum Power PointTracking is electronic tracking, usually digital. TheMPPT controller checks the variable load requirementsand panel output at any instant. It then simulates theload conditions to get maximum power from the panelusing digital techniques. MPPT changes the current orvoltage output to load to simulate ideal load conditionsfor maximum power from the panel. Most modernMPPTs are around 93-97% efficient in the conversion.They deliver a 20-45% power gain in winter and 10-15% in summer. Actual gain can vary widely dependingon weather, temperature, battery state of charge, andother factors. The MPPT controller will harvest morepower from the solar array. The performanceadvantage is substantial (10-40%) when the solar celltemperature is low (below 45°C), or very high (above75°C), or when irradiance is very low.

A typical solar panel power graph (Figure 1) shows theopen circuit voltage to the right of the maximum powerpoint. The open circuit voltage (VOC) is the maximumvoltage that the panel outputs, because no power isbeing drawn from the circuit. The short circuit current ofthe panel (ISC) is another important parameter,because it is the absolute maximum current that can bereceived from the panel.

The maximum amount of power that can be extractedfrom a panel depends on three important factors:irradiance, temperature and load.

Figure 1 shows the effect of different irradiance levelson the panel voltage, current and power. Irradiancemainly changes the panel operating current.Temperature changes the panel voltage operatingpoint. To match the ideal panel impedance to loadimpedance, a DC-DC converter is used. For example,a 5V/2A (i.e., 10W) load is supplied from a 20W PVpanel with MPP at 17.5V/1.15A. The panel short circuitcurrent is 1.25A. If the load is connected directly to thepanel, then it will provide 5V*1.15A = 5.75W max toload and not 10W. Thus, the panel will not operate atMPP. A DC-DC converter can be used to variablychange the load conditions with varying duty cycle tomodify load voltage or current seen from the panel totrack MPP.

Since solar MPPT controllers are installed on remotefields, and sometimes, on remote locations, specialconsiderations should be taken when designing thesystem. One suggestion would be to design the systemas a single module potted in plastic. Doing this wouldprovide protection against dust, water and help withmoisture resistance. Another suggestion would be toadd heat sinks for passive air cooling. This would helpincrease the longevity of the parts and boards. Thesystem should be designed as such that even if theconnections are wired incorrectly, the unit should notget damaged.

FIGURE 1: SOLAR PANEL CHARACTERISTICS

DS00002321A-page 2 2016 Microchip Technology Inc.

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There are several MPPT algorithms that can be easilyimplemented using an 8-bit PIC® microcontroller:

1. Perturb and Observe (P&O): As the namesuggests, the current or voltage output of MPPTDC-DC converter is increased or decreased inlinear steps to alter its normal or regular state. Ifthere is an increase in power output from thepanel, increase perturbation further to see ifoutput power increases again. If output power isless, reduce the perturbed output by linearsteps. This process of perturbation andobservation slowly reaches to MPP of the panel.The algorithm is very simple in terms ofimplementation. However, there are a fewdisadvantages to this type of system, such asthe inherent steady-state oscillations at theMaximum Power Point (MPP), if the linear stepsize is too large. Also, if the step size is too smallor the system takes too long to respond toperturbation, it will take a longer time to reachMPP.

2. Incremental Conductance: In this algorithm, thecurrent or voltage output from MPPT DC-DCconverter is incremented or decremented inlinear or nonlinear steps. But instead of the

output power being measured, the ratio of powerdifference to voltage difference (i.e., dP/dV), isused to determine the MPP. As shown inFigure 1, dP/dV is a typical MPP curve providedby a panel manufacturer. If the ratio is negative,the MPP is on the left, and so panel current I, orpanel voltage V, should be reduced. If the ratiois positive, the MPP is on the right and the panelcurrent I, or panel voltage V, should beincreased. If the ratio is zero, the controller hasreached MPP. The algorithm is a bit complex forimplementation compared to P&O. If the stepsize is linear, the algorithm exhibits steady-stateoscillations at MPP. But this can be solved withnonlinear step size. The value of step size canbe changed with respect to dP/dV output. Theratio output can also be used in a PID controlleras error value. Then the oscillations can befiltered out, and the system also respondsquickly to change in MPP with respect toatmospheric changes.

For details about the algorithms, refer to the applicationnote AN1521, “Practical Guide to Implementing SolarPanel MPPT Algorithms” (DS00001521).

FIGURE 2: SOLAR MPPT BATTERY CHARGER BLOCK DIAGRAM

DC-DC Boost Converter

PIC16F177X MCU

12V 55AHr

+ -

ADC VBAT

16 bit PWM

130W, 12V Solar Panel

PWM

DC-DC converter Control

IPANEL

MPPT Algorithm

Battery Charge

Algorithm

Panel Current sense

COG PWMPWMPWMPWMP

Micrel MOSFET Driver MIC4104

12V 55AHr

12V 55 AHr batteries connected in series + -

24VBAT

Panel

Current Mirror

12 VDC 2A power for charging of mobile electronics

PWM

EUSART

Bluetooth Module RN4020

UART Interface

Radio

MODEM

Status information via a low power Bluetooth link

OPA

COG CMP

PRG

RC

OPADAC PRG

DAC

CMP

DADAC

OPA

DC-DC Buck

Converter

12V 5W LED Bulbs Dimmable

Dimmer1

RC

Dimmer2

12V, 420 mA LDO

Linear Regulator

VDD 5V

VPANEL VBAT2

VCC 11V for MOSFET Driver

VBOOST

VBOOST

CMP

DCDC DCDCDAC

IPANEL

LED Dimming Engine

12V, 420 mA DC-DC Buck Converter

1DC-DC Buck Converter

16 bit PWM

OPA

tDSM

CMP

PRG

RC

WM

OPADAC

LED Dimming Engine Using internal peripherals OP-AMP, DAC, PRG, Comparator, PWM, COG and DSM

12V, 2A

VBAT2 For Charge Balancing

VBAT2

IPANEL

LDO 3.3V For MCU and RN4020

LED Dimming

Engine

COG 16 bitt

MP

G

PWM

VPANEL

RS FS

Dimming CTRL2

Dimming CTRL1


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SOLAR PANELS

130W/12V solar panels used in the system are readilyavailable on the market. A list of solar panelmanufacturers is provided in Appendix D: “SolarPanel Manufacturers”.

MPPT DC-DC BOOST CONVERTER

The panel voltage will be in the range of 15V-22V. Thiswill be stepped up to 26.8V using a boost converter tocharge the 24V battery. The PIC microcontrollergenerates the necessary control signals to drive theDC-DC boost converter that converts the solar panelpower to charge the battery and to supply the load.Peak current-mode controlled (PCMC) DC-DC boostconverter is used. The peripherals used are 10-bitPWM, COG, op amp, comparators, DAC, PRG, andFVR to generate the PWM signals for the DC-DC boostconverter. The MPPT is implemented using acomparator with DAC reference. The value of DAC iscontrolled using the MPPT algorithm explained above.For details about the implementation of PCMC DC-DCboost converter, refer to Section “PCMC DC-DCBoost Converter with MPPT”.

Battery

The design uses two 12V, 55 AHr deep-cycle sealedlead acid (SLA) marine batteries connected in series tomake a 24V battery. These batteries are typically usedfor boats and marine applications to handle deepdischarge. Most car batteries can do high startingcurrent, but are only designed for 10-20% discharge,and their lifetime suffers if it is discharged deeper thanthat. The other advantage marine batteries have is thatthey are typically sealed, therefore the user does nothave to worry about acid leakage. The big advantageof a deep-cycle lead acid battery is that it can becharged from a constant voltage source with whatevercurrent is received from the solar panel. This meansthere is no need for separate switchers for the MPPTand battery charging, as only one with a constantoutput voltage and variable current can be used. Ifmultiple batteries are connected in series, a chargebalancing is required as explained in Section“Battery Charging and Charge Balancing” below.While deep-cycle batteries are preferable, normal carand tractor batteries can be used with the system. Thiswill be able to detect and prevent damage if thebatteries are connected incorrectly or in reversedirection.

System Load

Three DC-DC buck converters are used for generatingthe 12V output going to the two LED bulbs and mobileelectronic charger load. The DC-DC buck convertersare controlled using on-chip op amp, DSM, DAC, PRG,comparator, 10- and 16-bit PWM and COG peripheralsof the PIC microcontroller.

LED Bulbs

There is a provision to connect two 12V/5W LEDs thatare powered by one of two different types of drivers.LED driver outputs may be constant voltage (usually12 V or 24 V) or constant current (e.g., 350 mA,700 mA or 1050 mA). This system has an output ofconstant voltage 12V.

LED DIMMING

Constant voltage LED drivers can be dimmed via aPWM method.

The dimming engine for the two LED bulbs isimplemented using on-chip op amp, comparator, PRG,16-bit PWM, CCP, DAC, DSM and COG peripherals ofthe PIC microcontroller.

The dimming of the LED bulbs is implemented usingtwo switch buttons. Depending upon the dimming levelthe duty cycle of the PWM going to the LED can beadjusted. For details about the LED dimming engineimplementation, refer to Section “LED DimmingEngine”.

Current consumed by the LED bulbs is sensed by acomparator. In case of short circuit or currentexceeding 500 mA, the comparator output will go highand, since it is connected to the auto-shutdown pin ofthe COG, this will cut the 12V power going to the LED.

12 VDC POWER FOR CHARGING OF MOBILE ELECTRONICS

The system will provide a 12V, 2A output for driving astandard automobile cigarette lighter socket.

Current consumed by the load will be sensed by acomparator. In case of short circuit or currentexceeding 2A, the comparator output will be triggered.The comparator output is connected to the auto-shutdown pin of the COG. This will cut the 12V powergoing to the load.

TOTAL SYSTEM LOAD

For six hours of light per day, two 5W LEDs willconsume 2x5W @ 6hrs = 60 WH.

The 12V output at 2A will require 24Wx24 = 576 WH,assuming a 24-hour usage.

The fully charged system can operate for two days atmaximum load without charging.


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The system can operate lights for three weeks on asingle 10-hour charge if 12V charging output is notused.

Load Regulation

The output of DC-DC buck converter is connected todifferent loads, which is controlled or regulated by themicrocontroller. There are different types of load used,LED lamps, small chargers.

If the battery is deep discharged and needs recharged,the loads can be switched off and all current goes to thebattery, making it charge faster. If the battery is fullycharged, most of the power goes to different loads andthe battery charging is off. This load regulation is donein the MCU using PWM control.

HARDWARE DESIGN CONSIDERATIONS

PIC16F1779 Advantages and Features

PIC16F1779 is an 8-bit PIC microcontroller thatcombines the processing power of CIPs and intelligentanalog peripherals with a functionality of amicrocontroller on a single chip to create a cost-effective solution. PIC16F1779 includes manyperipherals that are especially useful for switch modepower supply, power management, and medicalmonitoring applications.

PIC16F1779 provides the following peripherals:

• Fixed Voltage Reference (FVR)• 10-bit Analog-to-Digital Converter (ADC) • 5-bit Digital-to-Analog Converter (DAC)• 10-bit Digital-to-Analog Converter (DAC)• Op amp• Programmable Ramp Generator (PRG)• High-Speed Comparator: High-Speed Comparator

modules which can use 5-bit/10-bit DACreferences for comparing two analog inputvoltages. The comparator is designed to operateacross the full range of the supply voltage (rail-to-rail operation).

• 10-bit Pulse-Width Modulation module (PWM)• 16-bit Pulse-Width Modulation module (PWM)• Complementary Output Generator (COG)• Zero-Cross Detect (ZCD)• Configurable Logic Cell (CLC)• Data Signal Modulator (DSM)• Master Synchronous Serial Port (MSSP)• Enhanced Universal Synchronous Asynchronous

Receiver (EUSART) • 8-bit Timers• 16-bit Timers

The implementation of a solar MPPT charger with LEDloads system makes use of the following peripherals ofPIC16F1779 to achieve optimum performance:

• Analog-to-Digital Converter (ADC)• Digital-to-Analog Converter (DAC)• Op amp• Programmable Ramp Generator (PRG)• High-Speed Comparator module • 10-bit Pulse-Width Modulation module (PWM)• 16-bit Pulse-Width Modulation module• Complementary Output Generator (COG)• Data Signal Modulator (DSM)• 8-bit Timer

When using a DC-DC converter it is essential to havegood voltage regulation and transient responses over awide load current range. Voltage-mode control andcurrent-mode control are the major control strategies.Current-mode control has good dynamic performanceand inherent properties, such as short circuitprotection. These advantages make current-modecontrol more suitable for mission-critical applications.

PCMC DC-DC Boost Converter withMPPT

The DC-DC boost converter with PCMC converts thesolar panel power to charge the battery and to supplythe load. It also includes MPPT.

DESIGN REQUIREMENTS

The design requirements are shown in Table 2 below:

For the component selection in boost converter powersection, refer to Section Appendix F: “HardwareDesign for PCMC Boost Converter”. Coilcraftprovides design tools for selection of a power inductorto be used in a DC-DC converter. Select Design Tools Power Tools DC-DC Inductor finder. Accordingto calculations the inductor value selected is 15 μH withpeak current of 18A and RMS current of 20A. Coilcraftpart no. SER2918H-153.

TABLE 2: DESIGN REQUIREMENTS

Parameter Specs Unit

Nominal Input Voltage VIN(nom) 17 VDC

Maximum Input Voltage VIN(max) 21.25 VDC

Minimum Input Voltage VIN(min) 12 VDC

Output Voltage VOUT 26.8 VDC

Maximum Output Current IIN(max)

4.5 A

Minimum Output Current IIN(min) 200 mA

Inductor Ripple Current Ratio r 30 %

Maximum Output Voltage Ripple ΔVOUT

300 mVp-p

Switching Frequency Fs 250 kHz

Estimated Efficiency > 90 %


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CONTROL SYSTEM DESIGN

Control loop design is a significant part in the design ofpower converters. It is implemented to ensure stabilityof the controller. For a high-performance powerconverter design, the control must be designed for highgain and bandwidth to ensure good regulation and fastresponse. The output variations due to line or loadchanges are brought to a minimum using feedbackcontrol loop techniques. There are various controltechniques that control the duty cycle of the converter.The most popular control technique is Peak Current-Mode Control (PCMC).

The control signals for the boost converter aregenerated using PCMC. The current-mode control hastwo feedback loops: the inner current loop and theouter voltage loop. In the voltage loop, the outputvoltage is compared with a reference voltage. The erroris processed by the compensator network to generatethe reference signal for the inner current loop. Thereference voltage is generated using the DAC

peripheral. The compensator is implemented usinginternal op amp. In the current loop, the referencesignal is compared with the measured inductor current.When the inductor current reaches the peak value, thecomparator generates a signal and the switch is turnedoff. The PWM signals for the synchronous DC-DCconverter are generated using PWM and thecomplementary output generator (COG) peripheral.The rising edge and the frequency of the PWM signalis determined using a 10-bit PWM peripheralconnected to the COG. The falling edge of the PWM isdetermined by comparator 1. This generates therequired duty cycle to maintain the output voltagewithin the specified limit. Figure 3 illustrates the PeakCurrent-Mode Control in a boost converter. Theperipherals used in the design can be configuredquickly for the required settings using MPLAB CodeConfigurator (MCC), as shown in Appendix C:“Peripheral Configuration Using MCC”.

FIGURE 3: PCMC DC-DC BOOST CONVERTER WITH MPPT


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The plant transfer function in current-mode control isgiven by Equation 1 below:

EQUATION 1: PLANT TRANSFER FUNCTION IN PCMC

PCMC suffers from sub-harmonic instability for dutycycles greater than 50%. In this case, the user needsto add a ramp to inductor current for stabilizingoscillations. This is called slope compensation. Theramp can be added to the inductor current voltageimage or subtracted from the reference current level.An artificial ramp is subtracted from the referencesignal of the comparator and therefore meets thefeedback signal at the desired point. The comparatoralways trips the PWM.

Slope Compensation using Internal Programmable Ramp Generator (PRG)

The PRG peripheral of the PIC16F1779 device can beused for the slope compensation to remove the sub-harmonic oscillations.

PRG Module Configuration:

1. PRG discharges the internal capacitor quickly atthe beginning of the PWM period

2. It charges the capacitor at the programmed ratewhich determines the slope

3. The capacitor voltage is subtracted from thevoltage source to produce ramp decay

4. Voltage source is selected as op amp output(i.e., output of the voltage compensator)

5. The ramp is started by capacitor charging whenset rising edge input goes true

6. The ramp is stopped and the capacitor isdischarged when set falling edge input goes true

7. For rising and falling edge timing input, PWMcan be selected. The ramp can be started atPWM rising edge and stopped at PWM fallingedge

8. Set the polarity of rising timing input as active-high. Set the polarity of falling timing input asactive-low

9. Slope calculations are described in the followingsection

To account for sub-cycle oscillations the user needs toadd a high-frequency term to the existing powerconverter transfer function, as shown in Equation 2.

EQUATION 2: PLANT TRANSFER FUNCTION WITH HF TERM

The high-frequency transfer function is given byEquation 3.

EQUATION 3: HIGH-FREQUENCY TRANSFER FUNCTION

The double pole frequency is at half the switchingfrequency Ts and is given by Equation 4 below.

EQUATION 4: DOUBLE POLE FREQUENCY

The damping factor QP is given by Equation 5.

EQUATION 5: DAMPING FACTOR QP

The compensation ramp factor is given by Equation 6.

EQUATION 6: COMPENSATION RAMP FACTOR

fp (s) K

1s

wz------+

1s

wRHP----------------–

1s

wp-------+

----------------------------------------------------------=

Where

wp2

RC--------,=

wRHPR 1 D–

2L

-------------------------------=

and

wz1

RC

C------------=

fp s fp s fh s =

fh s 1

1s

wn Qp-------------------

s2

wn2

-------+ +

--------------------------------------------=

wnTs-----=

QP1

mc D 0.5–

--------------------------------------------=

mc 1SeSn------+=


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The compensation ramp se is given by Equation 7.

EQUATION 7: COMPENSATION RAMP

The slope of the current waveform is given byEquation 8.

EQUATION 8: INDUCTOR CURRENT UPSLOPE

The amount of ramp to be added to the system can becalculated by first setting QP to 1 and solving the aboveequations. But the user should add a ramp that is morethan half of downslope of the inductor current.

For a boost converter, the downslope of the inductorcurrent is given by Equation 9 below.

EQUATION 9: INDUCTOR CURRENT DOWNSLOPE

The criteria for slope compensation is shown inEquation 10.

EQUATION 10: CRITERIA FOR SLOPE COMPENSATION

The ramp compensation is done as shown inEquation 11.

EQUATION 11: COMPENSATION RAMP CALCULATION

The Block Diagram of the Overall Control System

The small signal block diagram of the power converterfor peak current-mode control is given by Figure 4.

FIGURE 4: BLOCK DIAGRAM FOR POWER CONVERTER FOR PCMC

Note: ISET<4:0> = 0x1 for compensating aramp slope of 0.25.

Se

Vp p–Ts

---------------=

Sn

Von Ri

L--------------------=

m2

Vo Vin– Ri

L------------------------------------=

Se

m22

-------

Se26.8 12– 0.25

2 15--------------------------------------------V/S

Se 0.1233 V/S

Power Stage

Fm

(s)

Ri Kr Kf

Where,

VIN = the perturbation of the input voltage

VO = the perturbation of the output voltage

iL = the perturbation of the inductor current

GVD(s) = the control to output transfer function

GID(s) = inductor current transfer function

fH(s) = sampling gain term

RI = sense resistor

FM = modulator gain

KF and KR = feed forward and feedback gains


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Compensator Design for Peak Current-Mode Control (PCMC)

It is relatively easy to design the current loop in aPCMC power converter, because there is nocompensator involved. When the current loop isclosed, the control to output transfer function GVC(S) isgiven by Equation 12 below.

EQUATION 12: CONTROL TO OUTPUT TRANSFER FUNCTION

GVC(S) can be simplified to a second order system, asshown in Equation 13. A current mirror was used in thecurrent design loop. With a current mirror the value ofthe sense resistor Ri is given by Equation 13.

EQUATION 13: SIMPLIFIED GVC(S)

The GVC(S) of the system for the given specifications isshown in Equation 14.

EQUATION 14: GVC(S) OF THE SYSTEM

Once the control-to-output transfer function iscalculated, the compensator is designed in such a waythat a good bandwidth with a phase margin greaterthan 45° is obtained. The poles and zeros of thecompensator should be placed by analyzing thecontrol-to-output transfer function of the converter. Theopen-loop bode plot obtained using Scilab is shown inFigure 5 below.

FIGURE 5: OPEN-LOOP BODE PLOT

Current-mode control boost converter can becompensated with a type II compensator. A type IIcompensator has two poles: one at origin and one atzero. The transfer function of a type II compensator isgiven by Equation 15.

EQUATION 15: TYPE II COMPENSATOR TRANSFER FUNCTION

The poles and zeros of the above transfer function aregiven by Equation 16.

EQUATION 16: POLES AND ZEROS OF TYPE II COMPENSATOR

Gvc s voˆ

vc-----

Fm Gvd s

1 Fm Ri He s Gid s Fm Kr Gvd s –+--------------------------------------------------------------------------------------------------------------------------= =

Where,

Kr

D'2

Ts Ri

2L-----------------------------=

Gid(s) and Gvd(s) for boost converter are given as:

Gid s

2Vo

D'2

R---------------- 1

RC2

-------- s+

1L

D'2

R------------ s LC

D'2

-------- s2

+ + -------------------------------------------------------------=

Gvd s Vo

D'------

1L

RD'2

------------s–

1L

D'2

R------------s

LC

D'2

-------- s2

+ + ---------------------------------------------------------=

Ri R2

R4R1------ 0.02

22018

--------- 0.25= = =

fRHPR 1 D–

22Lmin

-------------------------------6 1 0.55–

2

2 12 106–

------------------------------------------- 15836.3 Hz= = =

wRHP

fRHP2

-------------- 99500 rad/s= =

wP2

RC--------

2

6 151 106–

------------------------------------- 2207.5 rad/s= = =

Gvc s

R 1 D– 1s

wRHP----------------–

2Ri 1s

wP-------+

--------------------------------------------------------------

Gvc s

R 1 D– 1s

wRHP----------------–

2Ri 1s

wP-------+

--------------------------------------------------------------

5.4 5.427 105–

s–

1 4.53 104–

s+ ------------------------------------------------------

-200

-150

-100

-50

0

50

-200

-150

-100

-50

0

50

1 10 100 1000 10000 100000 1000000

Pha

se (°

)

Gai

n (d

B)

Frequency (Hz)

Bode Plot of GDC(1-s/RHP)/(1+s/p)

Hc s 1 R2 C2s+

R1 C2s 1 R2 C3s+ --------------------------------------------------------------------=

fP01

2 R1 C2 ----------------------------- fP1 1

2 R2 C3 ----------------------------- fZ1 1

2 R2 C2 -----------------------------= = =


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The above transfer function uses the approximation C2>> C3. The pole at zero forms at the integrator section,and it is given fP0. fP0 needs to be set to get the desiredcrossover frequency. fP1 is usually set at the EquivalentSeries Resistance (ESR) zero of the plant. ESR zero isat a higher frequency than the desired crossoverfrequency. Hence, it is set between fSW/2 and fSW. FZ1 isset at the output pole of the plant.

To design the compensator for the converter, followthese steps:

1. Choose the desired crossover frequency fC.Here the target crossover frequency of thecompensator is considered as fC = 5000 Hz.

2. fP0 is obtained from fP0 = fC /(A/B). A/B is the DCgain of the plant. For the simplified transferfunction for PCMC, the DC gain is obtained asshown in Equation 17.

EQUATION 17: PLANT DC GAIN

3. After setting R1 = 33 kΩ and fP0 = 1000 Hz, C2will be calculated as shown in Equation 18.

EQUATION 18: C2 CALCULATION

The standard value C2 = 4.7 nF will be used.

4. By setting fZ1 at fP(fP = 400 Hz), R2 will be ascalculated in Equation 19.

EQUATION 19: R2 CALCULATION

The standard value R2 = 100 kΩ was used.

5. C3 is calculated by setting fP1 at fSW/2(fP1 = 125kHz), as indicated in Equation 20.


The standard value C3 = 12 pF was used.

The compensator transfer function is computed asshown in Equation 21.

EQUATION 21: COMPENSATOR TRANSFER FUNCTION

Once the gain and phase plots of the open-looptransfer function are finalized, the system elementsmay need to be changed in order to get the bestpossible bandwidth and phase margin. Usually, thelocation of the poles and zeros are adjusted to get theoptimum gain margin and phase margin.

MPPT Operation

The MPPT tracking code is added to the basic outputregulation code and battery-charging algorithm.

The battery charge system operates in three modes:

• Constant Current• Constant Voltage• Charge Termination or Float mode

The MPPT tracking becomes relevant only whenbattery charge system is in Constant Current orConstant Voltage mode.

During battery charging, the output voltage of the boostconverter will be set equal to the desired fully-chargedbattery voltage. The battery will draw as much currentas possible, and the PWM duty will increase. Anothercomparator is used to limit the PWM duty, and in turnthe current delivered by the boost converter. Thereference point for the comparator will be set usingDAC which determines the current drawn from thesolar panel. While tracking the MPP panel, a number ofinput voltage and current samples are summedtogether for noise reduction, and then fed to theselected MPPT algorithm. When the required batteryvoltage is reached and the battery charging current hasfallen below C/10, then the Float mode is activated. Inthis mode the MPP tracking is not required, as thebattery is almost fully charged. Whenever the battery isfully charged, the PWM duty can be set to the minimumvalue just to float charge the battery and excess powerleft in the solar panel. If in float charge state the batteryvoltage falls below a certain threshold, then the Chargemode is activated again. If the panel voltage dropsbelow a certain threshold and/or during night time,when solar power is not available, the PWM controlsignals going to the boost converter are turned off.

AB---

R 1 D– 2Ri

-------------------------- 6 1 0.55– 2 0.25

---------------------------------- 5.4==

As a result,

fP0

fcA B-----------

50005.4

------------ 925 1000 Hz= = =

C21

2 R1 fP0-----------------------------------

1

2 33 103 1000

---------------------------------------------------- 4.82 nF= = =

R21

2 C2 fZ1

-----------------------------------1

2 4.7 109–

400------------------------------------------------------- 84.65 k= = =

C31

2 R2 fP1-----------------------------------

1

2 100 103

125000------------------------------------------------------------- 12.73 pF= = =

Hc s 1 R2 C2s+

R1 C2s 1 R2 C3s+ --------------------------------------------------------------------

1 4.7 104–

s+

1.551 104–

1.8612 1010–

s2

+ -----------------------------------------------------------------------------------------= =


AN2321

Battery Charge Balancing

Every battery has different discharge/recharge ratewhen connected in series, even if they have similarproperties. This is due to temperature, pressureaffecting battery chemistry. If one of the batteriescharges faster to full state, it will provide higherimpedance to source, thus reducing the current and therate at which other batteries charge. If theseincompletely charged batteries are used, their lifespanmay decrease. To solve this problem, battery chargebalancing needs to be implemented in circuits wherebatteries are connected in series.

Because it runs from solar, the user needs to makesure that the battery charging is the most efficient. Sothere is a need of a charge balancer that does notwaste energy like a resistive balancer would. Figure 6shows the charge balancing circuit.

As part of the charging algorithm, the microcontrollerwill periodically measure the voltages at VB12 andVB2.

VB1 is calculated by subtracting VB2 from VB12:

1. If VB1 and VB2 are within 50-100 mV of eachother, then the system goes back to charging

2. If VB1 is greater than VB2 by 100 mV,

- CBP1 is driven high - CBP1 is driven low when CBIS shows an

inductor current in L1 of approximately 1A- CBP1 is held off until CBIS shows an inductor

current in L1 of approximately 100 mA- every 5-10 cycles the voltages need to be

checked to make sure they are within50-100 mV; if they are within range, then goback to charging

3. If VB2 is greater than VB1 by 100 mV,

- CBP2 is driven high - CBP2 is driven low when CBIS shows an

inductor current in L1 of approximately 1A- CBP2 is held off until CBIS shows an inductor

current in L1 of approximately 100mA- every 5-10 cycles check if the voltages are

within 50-100 mV, if they are within range, then go back to charging

FIGURE 6: CHARGE BALANCING


AN2321

PCMC DC-DC Buck Converter for 12V/2ALoad

Synchronous DC-DC buck converter with PCMC isused for converting the battery power or the boostconverter output to 12V for supplying up to 2A currentfor mobile charging electronics.

DESIGN REQUIREMENTS

The specifications of the buck converter design areshown in Table 3 below.

For the component selection in boost converter powersection, refer to Appendix G: “Hardware DesignPCMC Buck Converter 12V/2A”. Coilcraft providesthe design tool for selecting a power inductor to beused in the DC-DC converter. Refer to Design Tools Power Tools DC-DC Inductor finder.

According to calculations, the inductor value selected is47 μH with a peak current of 4.6A and RMS current of4A. Coilcraft part no. MSS1210-473.

TABLE 3: DESIGN REQUIREMENTS

Parameter Specs Unit

Nominal Input Voltage Vin(nom) 24 VDC

Maximum Input Voltage Vin(max) 29 VDC

Minimum Input Voltage Vin(min) 20 VDC

Output Voltage Vout 12 VDC

Maximum Output Current Iin(max) 2 A

Minimum Output Current Iin(min) 200 mA

Inductor Ripple Current Ratio r 40 %

Maximum Output Voltage Ripple ΔVout

200 mVp-p

Switching Frequency Fs 250 kHz

Estimated Efficiency > 93 %


AN2321

Control System Design

Using the feedback control loops the duty cycle of thepower switch is controlled so that there will not be anyvariations in the output voltage. There are variouscontrol techniques that control the duty cycle of theconverter. The most popular control technique is Peakcurrent-mode control. PCMC DC-DC buck converter isdesigned using CIPs of PIC16F1779, such as DAC, opamp, PRG, comparator, Timer2, PWM and COG, asshown in Figure 7.

FIGURE 7: PCMC BUCK CONVERTER


AN2321

Compensator Design for Peak Current-Mode Control Buck Converter

In the current-mode control there are two loops:

• Inner Current Loop • Outer Voltage Loop

CURRENT LOOP DESIGN

In the control system, the current loop should bedesigned first. Since there is no compensator in thecurrent loop of a PCMC converter system, the currentloop design is relatively simple.

Two issues should be taken into consideration indesigning the current loop:

1. The method and gain of the inductor/switchcurrent sensing

2. The slope of the ramp signal for slopecompensation

Inductor/switch current sensing is done using a currentmirror. With a current mirror the value of the senseresistor Ri is given by Equation 22.

EQUATION 22: SENSE RESISTOR

SLOPE COMPENSATION CALCULATION

Usually, mc (i.e., the slope of the ramp signal to besubtracted from the error amplifier output forelimination of the subharmonic oscillations) is chosenin the range between ½ of m2 (down-slope of theinductor current) and m2; but the optimum slopecompensation is often found empirically. For a buckconverter, the downslope of the inductor current duringTOFF of the switching cycle is given by Equation 23.

EQUATION 23: SLOPE COMPENSATION CALCULATIONS

VOLTAGE-LOOP DESIGN

With current-mode control, the inductor of the buckconverter becomes a current-controlled source.

In PCMC, the open-loop control to output transferfunction of the power plant, GVC(s), is a combination ofthree terms: DC gain, a power stage small signalmodel, and a high-frequency transfer function given byEquation 24.

EQUATION 24: CONTROL TO OUTPUT TRANSFER FUNCTION OF THE POWER PLANT

In PCMC, the open-loop bode plot of gain and phase isas shown in Figure 8.

FIGURE 8: BODE PLOT OF PCMC PLANT

Ri R2

R4R1------ 0.05

22010

--------- 1.1= = =

m2

Vo Ri

L------------------=

The criteria for slope compensation is:

Se

m22

-------

So compensating ramp will be:

Se12 1.12 47

------------------- V S

Se 0.1404 V S

GVC s GDC

1S

ESR---------------+

1SP--------+

------------------------------- 1

1S

nQp---------------

S2

n2

----------+ +

-------------------------------------------=

Where,

GDC

RLOADRSENSE-----------------------

1

1RLOADTSW

L---------------------------------+

------------------------------------------- 61.1-------

1

16 4 47-------------------+

----------------------------- 4.69= = =

ESR 2FESR 2 12ESRCOUT------------------------------------ 1

ESRCOUT----------------------------= = =

P1

RLOADCOUT--------------------------------------

16 50------------------- 3333 FP

P2-------- 530= = = = =

n

TS------ Fn i.e half the switching frequency=

QP1

mcD' 0.5– -----------------------------------=

-200

-150

-100

-50

0

50

-200

-150

-100

-50

0

50

1 10 100 1000 10000 100000 1000000

Pha

se (°

)

Gai

n (d

B)

Frequency (Hz)

Pole FP

Cross-Over Frequency FC

Phase Margin

GDC

GDC *FP /F


AN2321

The PCMC open-loop plant is a single order system.Thus, there is no need of phase boost at desiredcrossover frequency. Generally, a type II compensator,as shown in Figure 9, will be sufficient for the PCMCsystem.

FIGURE 9: TYPE II VOLTAGE COMPENSATOR

The transfer function of the type II compensator isgiven by Equation 25.

EQUATION 25: TYPE II COMPENSATOR TRANSFER FUNCTION

The above transfer function uses the approximationC2 >> C3. Where the frequencies of poles and zerosare as shown in Equation 26, below:

EQUATION 26: POLES AND ZEROS OF COMPENSATOR

There will be an integrator or the pole at origin (i.e.,FP0). The second pole, FP1 of the compensator, can beplaced at ESR zero frequency or half the switchingfrequency, whichever is lower. The zero, FZ1, can beplaced at 1/5 of the desired crossover frequency orlesser than that.

The bode plots of type II compensator will be as shownin Figure 10.

FIGURE 10: BODE PLOT OF TYPE II COMPENSATOR

The following steps can be used for designing thecompensator:

1. Choose the desired crossover frequency FC.

2. Decide the poles and zeros of the compensator.

• FZ1 = 200<FP

• FP1 = lower of FESR or FSW/2 = 125000

3. The open-loop gain of the plant at desiredcrossover frequency can be calculated asshown in Equation 27.

EQUATION 27: OPEN-LOOP GAIN OF THE PLANT AT DESIRED CROSSOVER FREQUENCY

4. The compensator should provide gain of -GFC

(i.e., +12.7 dB) to have gain of 0 dB at FZ1 at FC.

5. The gain of the compensator at FZ1 is shown inEquation 28.

EQUATION 28: R2 CALCULATION

H S VC

VOUT----------------

1 R2C2S+

RaC2S 1 R2C3S+ --------------------------------------------------= =

FP01

2RaC2--------------------- FP1 1

2R2C3--------------------- FZ1 1

2R2C2---------------------= = =

GFc 20LOG10 GDC

FPF

------- 20LOG10 4.69

53012500---------------

12.7 db–= = =

GFZ1 20LOG10

GC0FZ1------------ 20LOG10

1

2RaC21

2R2C2---------------------

-------------------------------------------- 20LOG10

R2Ra------ 12.7= = = =

Selecting Ra = 20 K, R2 can be calculated using:

R2 Ra 10

GFZ120

--------------- 20K 10

12.720

---------- 83.6K= = =

The standard resistor value is 91 K.


AN2321

6. C2 can be calculated using Equation 29.


7. C3 can be calculated using Equation 30.


The combined bode plots of the closed-loop system forthe peak current-mode control will be as shown inFigure 11.

FIGURE 11: BODE PLOT FOR PCMC CLOSED-LOOP SYSTEM

LED Dimming Engine

If the LED is dimmed by turning its switch mode powersupply (SMPS) on/off with a timer-based PWM, acouple of problems may arise. As shown below, whenthe SMPS is off, the LED discharges the outputcapacitor giving the LED a slow turn off. When theSMPS is turned back on, the output of the erroramplifier is too high because the loop was attemptingto increase the output voltage during the off time. Theresult is that the output voltage is driven too high beforethe loop can correct.

FIGURE 12: TIMING DIAGRAM OF LED DIMMING ENGINE

C21

2R2FZ1-------------------------

1291K 175--------------------------------- 8.74 nF= = =

The standard resistor value is 10 nF.

C31

2R2FP2-------------------------

1291K 125000----------------------------------------- 9.994 pF= = =

The standard resistor value is 10 pF.

-250

-200

-150

-100

-50

0

50

100

-250

-200

-150

-100

-50

0

50

100

1 10 100 1000 10000 100000 1000000 10000000

Pha

se (°

)

Gai

n (d

B)

Frequency (Hz)

100100

Closed loop Plant Compensator

Cross-Over Frequency FC

Phase Margin


AN2321

Therefore, turning on and off the SMPS causes a slowdim out and a bright pulse at the turn on. This appearsas scintillation in the light output.

To correct the problem, three steps must beimplemented in synchronization with the PWM:

1. The LED must be disconnected from the outputcapacitor to prevent a discharge of the capacitorand a slow dim-out during the dimming off timeand then the LED must be reconnected duringthe dimming on time. This is typicallyaccomplished by a MOSFET on the cathodeside of the LED.

2. The time-based trigger of PWM pulses in theCOG must be shut down during the dimming off

time to prevent overcharging of the outputcapacitor. During the dimming on time, the time-based trigger must resume. This is typicallyaccomplished by using the DSM to gate thetime-based pulses into the COG rising input.

3. The output of the op amp must be tri-stated toprevent changes to the loop filter output duringthe dimming off time. During the dimming ontime, it must be reconnected. This isaccomplished by using the output enableoverride function of the op amp. Thisdisconnects the external loop filter componentsfrom the op amp and the PRG.

Figure 13 below shows an example block diagram.

FIGURE 13: LED DIMMING ENGINE USING PIC16F177X

The 16-bit PWM is just below the COG. Its output drivesQ4 which turns off the LED during the dimming off time.It also tri-states the output of the op amp and it gatesthe output of the PWM so that during the off time norising events are sent to the COG.


AN2321

FIRMWARE

Most of the functionality of the control system isimplemented using CIPs of PIC16F1779. This onlyrequires the peripherals’ initialization, as afterwardsthese work independently of the CPU. As the usage ofCIPs does not require much CPU intervention, theremaining CPU bandwidth can be used for addingother enhancement features to the design.

MPPT Algorithm

With varying weather conditions, the maximum powerpoint for the solar panel also varies. MPPT controlleralgorithm will track this maximum power point in anyweather condition. As mentioned above, two different

algorithms are used for MPPT. A timer is usedperiodically to interrupt the MCU for MPPT tracking,usually at 10 Hz or less. It measures the input andoutput power of the system. It changes the currentsupplied to load and battery charging using PWM toreduce or increase the input current from the solarpanel and achieve maximum power point.

A DC-DC boost converter is used to boost input voltagefor battery charging and supplying to load, which is alsopart of an MPPT controller. This DC-DC converter isrunning at a higher speed than the MPPT tracker. TheDC-DC converter acts as load to the MPPT algorithmand simulated ideal load for maximum power pointtracking, hence it runs at a higher speed.

FIGURE 14: MPPT FLOWCHART

Timer Expired?

YES

Read Vpanel, Ipanel,Vboost, Vbat, Vbuck, Vled1 and

Vled2 from ADC

NO

PO or InCond? PO dP>0

dV>0

YES

dV>0NO

D=D-dD

Ppanel=Vpanel*Ipanel;dP=Ppanel-Ppanel_old;dV=Vpanel-Vpanel_old;

dI=Ipanel-Ipanel_old;

YES

NOD=D+dD

YES D=D+dD

Vpanel_old=Vpanel;Ppanel_old=Ppanel;

dV=0 InCond

dI=0

dI>0D=D-dD

D=D+dD

YES

No Change

YES

dP/dV=0

NO

NO

YES

NONo Change YES

dP/dV>0

NO

YES

NO

RETURN A

A

NO

MPPT Correction

Yes

Note 1: 1. The D is the duty cycle of the PWM for boost converter.

2: 2. The dD is the small increment/decrement factor for duty cycle.


AN2321

Battery Charging and Charge Balancing

The PIC microcontroller also monitors the batterycharging process (i.e., battery voltage using ADC), andprovides status information based on battery condition.The battery charging state is also monitored by abattery charge balancing circuit. If there is voltagediscrepancy between battery voltages, the chargebalancing circuit activates and starts balancing voltageor charge levels of the batteries.

LED Dimming

The dimming level for the LED bulb can be set usingtwo switches SW1 and SW2. For dimming control ofthe LED1 bulb, both SW1 and SW2 should be pressedfor a few seconds. Then SW1 should be pressed again.To increase the brightness press SW2 and SW1 fordecreasing it. Similarly, for dimming control of the LED2bulb, press both SW1 and SW2 for a few seconds, thenpress SW2. Again, to increase the brightness pressSW2 and SW1 for decreasing it.

Monitoring

The system monitors the solar panel output anddisables the MPPT boost controller in darkness. Thesystem also monitors the battery voltage while batteryhealth information, such as deep discharge battery, canbe communicated to the user. The system monitors thebattery charge level at regular intervals, and duringbattery discharge the remaining battery life informationcan be communicated to the user using low-powerBluetooth over a cell phone.

Automated Functions for Fault Safety

FAULT DETECTION AND CORRECTION

Short-circuit protection is provided for LED bulbs andthe 12V/2A charger output. The output of the DC-DCbuck converters is monitored using ADC. In case of ashort circuit, the output will go to zero or very lowvoltage. If the ADC value of the output goes below a setthreshold, then the short-circuit condition is detected.Then the DC-DC converter goes in the Auto-shutdownmode and the short-circuit event is communicated tothe user. The DC-DC converter remains in Auto-shutdown mode until the short circuit is removed.

Low-Power Bluetooth Communication forStatus and Alarm Reporting

The on-board RN4020 is used for communicating thestatus information and alarms on the Android™application using the Bluetooth low-energycommunication. The RN4020 module communicatesvia a PIC MCU using UART. The brightness of aparticular LED bulb can be controlled with the slider onthe Android application. The dimming level can be sentto the system using the Android application and BLEcommunication.

The status information such as remaining batterycharge, panel voltage, panel current and alarms (shortcircuit in the 12V outputs) can be sent to the user usingBLE.

Resource Utilization

Table 4 summarizes the resources used in the solar-based rural electric power system.

TABLE 4: RESOURCE UTILIZATION

Parameter Total Available Used

Flash Memory (Words) 16K

Data SRAM (Bytes) 2K

High-Endurance Flash (Bytes) 128

Peripherals ADC, DAC, FVR, op amp, comparator, PRG, CCP, PWM, COG, ZCD, MSSP, EUSART, Timers, DSM

ADC, DAC, FVR, op amp, compara-tor, PRG, PWM, COG, 16-bit PWM, EUSART, Timers, DSM


AN2321

Algorithm

FIGURE 15: FIRMWARE FLOWCHART

PERFORMANCE DIFFERENTIATORSOF THE DESIGN

There is a significant number of differentiators in thisdesign that can be attributed to the use of thePIC16F1779 microcontroller. These differentiators areabove the basic functionalities that the system issupposed to offer. A few of the more important ones arelisted below.

Simplicity of Circuitry

Most of the functionalities of the control system areimplemented using CIPs of the PIC16F1779 device.The primary factor that leads to the simplicity of thisdesign is that it is possible to control up to four differentDC-DC converters with a single PIC MCU, thuseliminating a lot of additional circuitry that aconventional design would need. Using CIPs greatlyimproves the overall performance and implementationof the system because the use of CIPs do not requiremuch CPU intervention. Hence, the remaining CPUbandwidth can be used for adding other enhancementfeatures to the design. Secondly, the elimination ofexternal hardware components, which are now part ofthe microcontroller, greatly improves the reliability ofthe system in terms of avoiding failure of externalcomponents due to improper layout, routing andthermal stress. All these factors are already taken careof inside the microcontroller for the CIPs.

The slope compensation is one such example. It is avery important module in case of a PCMC DC-DCconverter. There are many advantages to using theinternal slope compensation (PRG) of amicrocontroller. It requires less device pins to be usedin the application. When using external slopecompensation circuit, for changing or adjusting theslope of the ramp for particular application, the value ofthe resistor or the capacitor has to be changed inhardware. But, if using internal slope compensation(PRG), the ramp slope can be changed easily just bychanging the register in firmware (i.e., without anyhardware change).

Monitoring

Various fail-safe and power-saving functionalities canbe achieved with this design because of its inherentcapability of including monitoring functions in thePIC16F1779 microcontroller. Some importantmonitoring features are:

• System monitors the solar panel output and itdisables the MPPT boost controller in darkness

• System monitors the battery voltage and thebattery health information, such as deepdischarge battery, and communicates them to theuser

Timer Expired?

YES

Read VPanel, IPanel,VBOOST, VBAT, VBUCK, VLed1 and VLed2

from ADC

NO

MPPT Algorithm

MPPT Correction

Yes

START

Initialize:ADC, COMP, DAC, Op-Amp, PWM,

Timers, COG, PRG, 16 bit PWM, DSM

Start timer at 1kHz or higher for settings ADC

sampling rate

Auto shutdown COG for 250 cycles

Short CKT

Yes

LED Dimming

Dimming Key Press

Yes

NO

NO

RETURNA

A

No


AN2321

• System monitors the battery charge level atregular intervals, and during battery discharge theremaining battery life information can becommunicated to the user using low-powerBluetooth over a cell phone

Control

Controllability of the various sections of this system toyield maximum efficiency, and addition of intelligenceto the system are key differentiators from othersystems.

• MPPT algorithm is implemented to operate thesolar panel at its maximum power

• Battery charge balancing is implemented toincrease the battery life

• Battery charging is terminated once batteries arefully charged by disabling the boost converter

• As seen above, LED dimming by turning on andoff the SMPS using conventional method causesa slow dim out and a bright pulse at the turn on.This appears as scintillation in the light output.This problem can be easily eliminated by buildingthe LED dimming engine using CIPs such asDSM, COG, 16-bit PWM, op amp, PRG andcomparator. This also removes color distortion.

Automated Functions

Automated functions are implemented for fault safetypurpose. Fail-safe operation and safety to operatingpersonnel has been given utmost importance in thisdesign. The system is designed to take care ofunexpected operation and certain scenarios due toexternal factors. A few of them are listed below:

• Fault detection and correction: Short circuitprotection is provided for LED bulbs and the 12V/2A charger output. If the short circuit condition isdetected, the DC-DC converter goes in the Auto-shutdown mode and the short condition iscommunicated to the user. The DC-DC converterremains in Auto-shutdown mode until the shortcircuit is removed.

• There is a check for battery presence, and if thebatteries are connected in proper polarity.

• For maximizing battery life and improvedefficiency, the MPPT is disabled in darkness. The12V output is disabled when not in use. The LEDbulbs are disabled if not in use. The LEDs go inAuto-shutdown mode after three hours with slowdim-out. The auto-shutdown of the LEDs isterminated if any of the dimming control key ispressed.

• If the battery charge is low, then the 12V/2Acharger output is disabled, and the LED bulbs areoperated in low intensity.

Communications

The low-power Bluetooth connection is provided forstatus and alarm reporting on mobile phones.

SCALABILITY

The system is designed for a 130W solar panel. Thesystem power can be scaled up by using higherwattage solar panels and a battery with higher AHrrating. 150W is about the maximum that this systemshould be configured for. For higher wattage the heatdissipation will become problematic. It should be takeninto account that the cooling is designed for passiveoperation.

The system can also be scaled down to have just one,two, or three lighting capability by eliminating 12V/2Aoutput, or by making it identical with output 1 andoutput 2.

CONCLUSION

This application note provides the design details of asolar MPPT charger for rural electrification systems,and the implementation of it using the intelligent analogand core independent peripherals of a PIC16F1779microcontroller. The availability of a variety of intelligentanalog peripherals, such as Analog-to-DigitalConverter (ADC), 5- and 10-bit Digital-to-AnalogConverter (DAC), op amp, Analog comparator andProgrammable Ramp Generator (PRG) along with coreindependent 10- and 16-bit Pulse-Width Modulator(PWM) and Complementary Output Generator (COG)make it suitable for power supply applications. Asdescribed in the application note, the control system forup to four different DC-DC converters and LEDdimming engine can be implemented using a singlePIC16F1779 microcontroller.


AN2321

APPENDIX A: REFERENCES

1. https://etd.auburn.edu/bitstream/handle/10415/328/HE_DAKE_26.pdf;sequence=1

2. Erikson, Robert W. and Maksimovic, Dragan –”Fundamentals of Power Electronics” (SecondEdition), ©2001, Springer Science and BusinessMedia, Inc.

3. L.H.Dixon, “Control Loop Cookbook,” UnitrodePower Supply Design Seminar Handbook, 1990

4. Dr. Ray Ridley-Designer’s Series Part V,“Current-Mode Control Modeling”

5. “Understanding and Applying Current-ModeControl Theory” (snva555), www.ti.com/lit/an/snva555/snva555.pdf

6. AN1521, “Practical Guide to Implementing SolarPanel MPPT Algorithms” (DS00001521)

7. PIC16F1769 Dual Independent Channel PowerSupply Demonstration – www.microchip.com/promo/dual-independent-demo

8. Inductor selector for DC-DC converter from Coil-craft – www.coilcraft.com

9. TB3140, “Programmable Ramp GeneratorTechnical Brief” (DS90003140)


http://www.microchip.com/promo/dual-independent-demo

http://www.microchip.com/promo/dual-independent-demo

http://www.coilcraft.com/apps/selector/selector_1.cfm

2

01

6 M

icroch

ip T

ech

no

log

y Inc.

DS

00

00

23

21

A-p

ag

e 2

3

AN

2321

AP

FIG

MCLR

GND

PGD/DAT

PGC/CLK

MCLR1

VDD2

VSS3

PGD4

PGC5

N.C.6

MCLR

VDD

VSS

PGD

PGC

N.C.

J4

112

J2

-F

D3 1kR15

1k

R16

39kR17

10kR18

0.010uFC10

GND

13

1 D6

GND

VBAT2

0.1uFC26

GND

0.1uFC32

123456

J5

100R

R30

100R

R31

100R

R27

100R

R28

100R

R29

RX

TX

GND

0.1uF

C29

10k

R26

0.1uFC28

GND

Batt1

Batt2

MMSZ5236BS-7-F

D4

112

J3

176

4

235

8

M5 DMP4025SFG

176

4

235

8M6

DMP4025SFG

10A

F1

CMD/MLDPWAKE_SW

39 μF25V

C31S1

CT2

P8208NLT

10A

F2

82kR13

10k

R14VBAT

0.01

0uF

0603

C9

VSS

C76

0.1 μF

C75

PGND

GND

100RR19

3.3V

3.3V

3.3V

3.3V

BATT_2

PENDIX B: SCHEMATICS, PCB LAYOUT AND REFERENCE DESIGN

URE B-1: BOOST CONVERTER

112

J1

0.05R

R1

39kR6

10RR2

100R

R5

10kR7

82RR3 82R

R4

PGND

GND

0.010 uFC5

VDD1

HB 2

HO 3

HS 4

HI5

LI6

VSS7

LO 8

U1

10kR21

10kR20

3RR9

3R1%

R8

GND

0.1 uFC13

GND

4.7 uFC14

MMSZ5236BS-7

1 23

BAT54SLT1

D5

2

BAT54SLT

ICBLNC

CHRG_BLNC2OPA1_IN-

OPA1_OUT

PWM_HO

PWM_LO

IPANEL

VPANEL

PWM_HO

PWM_LO

PWM2

PWM1

BOOST

1

GND

2

VFB 3EN4

SW 6

VIN5BOOST

GND VFBEN

SWVIN

U2MCP16301 SOT-23-6

OUT 1

SNS 2

SHDN 3

GND 4ERR5TAP6FB7IN8

U3

MIC2951-03YM TR

4.7uF

C16

0.1uFC20

10kR23

S3N

D7

1N4148

D8

4.7uF

C17

BATT_2

140kR22

VCC

SP

0.1uF50V

C21

1kR25

100pF50V0603

C25

100kR24

0.1uF50V

C24

VCC5V

GND

OPA3_IN-

RX

TXILED1

ILED2

ILOAD1OPA2_OUT

VBAT2

CHRG_BLNC2

PWM_DIM2

IPANELOPA1_OUT

VPANEL PWM_DIM1OPA1_IN-

PWM1

PWM2

VBAT

PWM3

PWM4

PWM5

PWM7

OPA3_OUTLED1_DIM

WAKE_SW

LED2_DIMOPA4_IN-OPA4_OUT

ICBLNC

0.1uFC27

GND

MCLRPGD/DAT

PGC/CLK

BOOST CONVERTERSolar Panel

+24V

TP1 SP

MOSFET Driver

VCC 10V/12V for MOSFET Driver

B140-FDICT-NDD10

10μF

50V

C3

0.010 uFC15

VCC

32

41

Q2BCM61B

32

41

Q1

BCM62B

DFLS160-7D1

SER2918H-15318 uH

L1

MSS1260-184 180uH

L2

L3

MSS6132-563

DFLS160-7D2

MMSZ5236BS-7-F

D9 CMD/MLDPCHRG_BLNC1

CHRG_BLNC1

176

4

23 5

8

M2

BSC039N06NS

17 6

4

2 358

M1

BSC039N06NS

OPA2_IN-

33kR10

1.2kR11

110k

R12

12pFC11

3.3nFC12

0.0RR84

0.0RR83

10 μF35V

C18

10 μF35V

C19

220 uF50V

C2210 μF16V

C23

873 1

XFRMR

0.010uF

C30

S3N

D13

470pFC34

10RR42

VDD1

HB 2

HO 3

HS 4

HI5

LI6

VSS7

LO 8

U8

10kR87

10kR88

0.1uFC69

GND

4.7uFC70

VCC

0.0R

R81

0.0R

R80

3R

R82 4

1,2,3

5,6,7,8

DMT6016LSS-13

M3

3R

R95 4

1,2,3

5,6,7,8

DMT6016LSS-13

M4

GND

RX

TX

WAKE-SW

CMD/MLDP

VSSVSS

VSS

GND1

AIO22

AIO13

AIO04

UART_TX5

UART_RX6

WAKE_SW7

CMD/MLDP8

GND

9

PIO1/SC

K10

MLD

P_EV

/PIO

2/SS

11

PIO4/MISO

13

WS/PIO3/MOSI

12

CTS

/PIO

514

WAKE_

HW

15

GND

16

SPI/PIO 17

RTS/PIO6 18

PIO7 19

RSVD 20

RSVD 21

RSVD 22VDD

23GND 24

U9

PGND

PGND

GND

0.0R

R96

0.0R

R97

PGND GND VSS

OUT 1

SNS 2

SHDN 3

GND 4ERR5TAP6FB7IN8

U10

MIC2951-03YM TR0.1uF50V

C71

1kR99

100 pFC73

0.1uF50V

C74

3.3V

GND

10 μF16V

C72

3.3V

91KR98

150K

R100

PGND

3.3V

TP5

TP6

OPA3IN0-/AN19/RC71

AN24/RD42

C7IN3-/C8IN3-/AN25/RD53

AN26/RD64

AN27/RD75

VSS6

VDD7

INT/AN12/RB08

OPA2OUT/AN10/RB19

OPA2IN0-/AN8/RB210

CCP2/AN9/RB311

NC12

NC_113

AN11/RB414

T1G/AN13/RB515

ICDCLK/ICSPCLK/RB616

ICDDAT/ICSPDAT/RB717

VPP/MCLR/RE318

CxIN0-/AN0/RA019

OPA1OUT/AN1/RA120

AN2/RA221

VREF+/AN3/RA322RA4/T0CKI 23OPA1IN0-/RA5/AN4 24RE0/AN5 25RE1/AN6 26RE2/AN7 27VDD_1 28VSS_1 29RA7/OSC1/CLKIN 30RA6/OSC2/CLKOUT 31RC0/SOSCO/T1CKI 32NC_2 33NC_3 34RC1/SOSCI/CCP2 35C5IN2-/C6IN2-/RC2/AN14 36CxIN4-/RC3/AN15 37RD0/AN20 38OPA4OUT/RD1/AN21 39OPA4IN0-/RD2/AN22 40RD3/AN23 41RC4/AN16 42RC5/AN17 43OPA3OUT/RC6/AN18 44OPA3IN0-/AN19/RC7

AN24/RD4C7IN3-/C8IN3-/AN25/RD5AN26/RD6AN27/RD7VSSVDDINT/AN12/RB0OPA2OUT/AN10/RB1OPA2IN0-/AN8/RB2CCP2/AN9/RB3NCNC_1AN11/RB4T1G/AN13/RB5ICDCLK/ICSPCLK/RB6ICDDAT/ICSPDAT/RB7VPP/MCLR/RE3CxIN0-/AN0/RA0OPA1OUT/AN1/RA1AN2/RA2VREF+/AN3/RA3 RA4/T0CKI

OPA1IN0-/RA5/AN4RE0/AN5RE1/AN6RE2/AN7VDD_1VSS_1

RA7/OSC1/CLKINRA6/OSC2/CLKOUTRC0/SOSCO/T1CKI

NC_2NC_3

RC1/SOSCI/CCP2C5IN2-/C6IN2-/RC2/AN14

CxIN4-/RC3/AN15RD0/AN20

OPA4OUT/RD1/AN21OPA4IN0-/RD2/AN22

RD3/AN23RC4/AN16RC5/AN17

OPA3OUT/RC6/AN18

U4

PIC16F1519-x_PT

10μF

50V

C81

10μF

50V

C2

10μF

50V

C79

10μF

50V

C80

220uF 50V

C1 220u

F50

V C8

10μF

50V

C7

10μF

50V

C87

10μF

50V

C86

1μF

50V

C6

10μF

50V

C83

10μF

50V

C84

AN

2321

DS

00

00

23

21

A-p

ag

e 2

4

20

16

Micro

chip

Te

chn

olo

gy In

c.

OPA2_OUT

112

J6

OPA3_OUT

13

2BC847AQ11

1

112

J7

OPA4_OUT

13

2847AQ12

GND

LED1_DIM

LED2_DIM

12V, 2A

12V, 500mA LED Bulb

12V, 500mA LED Bulb

17 6

4

2 358

M13

MCP87055 (PDFN)

17 6

4

2 358

M14

MCP87055 (PDFN)

112

TP4

TP3

TP2

0.0RR85

0.0RR86

F

2

1kR45

1kR62

1.1nF

C43

PGND

PGND

5V

5V

1kR46

1kR61

1kR79

1kR78

0.1uF

C77S2

GND

0.1uF

C78S3 23

14

HCPL-181

U12

23

14

HCPL-181

U11

1kR101

1kR102

112

HDR-2.54 Male 1x

2

J9

112

HDR-2.54 Male 1x

2

J103.3V

3.3V

FIGURE B-2: BUCK CONVERTER

GND

GND

OPA2_IN-

ILOAD1

3R

R65

3R

R66

VCC6

PWM-HI2

PWM-LO3

GND4 LODR 5

HIDR 8

PHASE 1

BOOT 7

U7

MCP14700-E/SN

1uF50V0603

C61

PWM3

PWM40.1uF

C59

+24V

+24V

10R

R36

100R

R3710kR3310k

R32

0.1uF

C36

GND

GND

OPA3_IN-

ILED1

3R

R34

VCC6

PWM-HI2

PWM-LO3

GND4 LODR 5

HIDR 8

PHASE 1

BOOT 7

U5

MCP14700-E/SN

1uF

C37

PWM5

0.1uF

C35

+24V

PWM_DIM

+24V

10R

R52

100R

R5310kR48

10kR47

0.1uF

C48

GND

GND

OPA4_IN-

ILED2

3R

R49

VCC6

PWM-HI2

PWM-LO3

GND4 LODR 5

HIDR 8

PHASE 1

BOOT 7

U6

MCP14700-E/SN

1uF

C49

PWM7

0.1uFC47

+24V

BC

PWM_DIM2

+24V

Buck Converter 24V to 12V, 500mA

Buck Converter 24V to 12V, 500mA, for LED2

Buck Converter 24V to 12V, 2A

MOSFET Driver 2

ILOAD1

Loadl C

urrent Sense

0.1RR51

10R

R52

100R

R53ILED2

R52

Load2 Current Sense

0.1RR77

10R

R36

100R

R37ILED1

Loadl C

urrent Sense

4

1,2,3

5,6,7,8

DMT6016LSS-13

M7

4

1,2,3

5,6,7,8

DMT6016LSS-13

M9

4

1,2,3

5,6,7,8

DMT6016LSS-13

M11

4

1,2,3

5,6,7,8

DMT6016LSS-13

M12

DFL

S160

-7 D12

DFL

S160

-7

D11

0.010uFC62

0.010uFC50

0.010uFC38

1.1kR38

1.1kR39

32

41

Q4

BCM61B

32

41

Q3BCM62B

32

41

Q6

BCM61B

32

41

Q5BCM62B

1.1kR55

1.1kR54

32

41

Q8

BCM61B

32

41

Q7BCM62B

680RR70

680RR71

10μF50V

C33

10μF50V

C45

10μF50V C57

50V10μFC52

50V10μFC53

1μF50V

C51

1μF35V

C63

35V10μF

C64

39μF

35V

C65

1μF50V

C39

L4MSS1260-184 180uH

L5MSS1260-184 180uH

L6MSS1210-473 47uH

J8

0.0R

R89

0.0R

R91

0.0R

R93

0.0R

R94

3.3kR40

300RR41

91k

R4415p

C4

10uF50V

C4010uF50V

C41

3.3kR56

300RR57

91k

R6015pF

C55

1.1nF

C56

5.6kR72

510RR73

15pF

C67

1.1nF

C68

91k

R76

10kR63

10kR64

0.05R

R67

100R

R69

10R

R68

0.1uF

C60

470pFC46

470pFC54

470pFC58

10kR35

10RR43

10kR50

10RR58

10RR59

PGND PGND

PGNDPGND

PGND PGND PGND

GND

GND

GND

5V

5V

5V

35V10μF

C85

AN2321

APPENDIX C: PERIPHERALCONFIGURATIONUSING MCC

The MPLAB® Code Configurator (MCC) plug-in forMPLAB X can be used to configure the peripherals inPIC® MCUs. The MCC provides Graphical UserInterface (GUI) tools to easily understand theconfiguration and select the required settings. Thisreduces the time for development. The MCC alsoprovides some built-in functions for working withspecific peripherals.

To install MCC, go to Tools Plugins AvailablePlugins MPLAB X Code Configurator Install.

To start and use the MCC after installation, go toToolsEmbedded MPLAB Code Configurator.

Figure C-1 to Figure C-8 show the configuration of thePIC16F1779 device peripherals such as DAC, op amp,PRG, comparator, Timer2, PWM, COG and DSMrespectively, used in the implementation of the solarMPPT charger system.

FIGURE C-1: DAC CONFIGURATION IN MCC FOR PIC16F1779

FIGURE C-2: OP AMP CONFIGURATION INMCC FOR PIC16F1779

FIGURE C-3: PRG CONFIGURATION IN MCC FOR PIC16F1779


AN2321

FIGURE C-4: COMPARATOR CONFIGURATION IN MCC FOR PIC16F1779

FIGURE C-5: TIMER2 CONFIGURATION IN MCC FOR PIC16F1779


AN2321

FIGURE C-6: PWM CONFIGURATION IN MCC FOR PIC16F1779

FIGURE C-7: COG CONFIGURATION IN MCC FOR PIC16F1779


AN2321

FIGURE C-8: DSM CONFIGURATION IN MCC FOR PIC16F1779


AN2321

APPENDIX D: SOLAR PANEL MANUFACTURERS

1. Tata power solar panel manufacturer in India –www.tatapowersolar.com

2. Moser Baer Solar (Model number 12140P canbe used) – www.moserbaersolar.com

3. Akshay solar from India has a variety of solarpanels – www.akshayasolar.com

4. Chinese manufacturers of 130W solar panelsare listed on the below website – www.made-in-china.com/

APPENDIX E: LOW-POWERBLUETOOTHCOMMUNICATION

The system can provide status, via a low-powerBluetooth link.

The status information can be:

• Remaining Battery Charge• Current Solar Panel Output (voltage/current)• Current Load (current)


AN2321

APPENDIX F: HARDWARE DESIGN FOR PCMC BOOST CONVERTER

Selection of Components in the Power Circuit

This section explains the critical parameters to beconsidered for each of the components in a boostconverter, and provides the calculation of thecomponent values based on the design requirements.The choices of components have a significant impacton the converter performance. The selectedcomponents are listed in the BOM section along withtheir manufacturer part number.

For more details about design equations andcomponent selection, refer to AN1207, “Switch ModePower Supply (SMPS) Topologies (Part II)”.

Inductor Selection

Coilcraft provides design tools for selection of a powerinductor to be used in a DC-DC converter. Refer toDesign Tools Power Tools DC-DC Inductorfinder.

While selecting an inductor for the boost converter, thefollowing parameters need to be considered:

• Minimum and Maximum Input Voltage• Output Voltage • Switching Frequency• Maximum Ripple Current • Duty Cycle

At VIN(MIN) = 12V and VOUT = 26.8V, the maximum dutycycle DMAX will be calculated as per Equation F-1:

EQUATION F-1: MAXIMUM DUTY CYCLE

For further calculations, the maximum duty cycle isused.

The design and selection of an inductor can influencethe choice and the cost for all other components of theconverter.

The inductance (L) is calculated as shown in equationEquation F-2.

EQUATION F-2: INDUCTOR CALCULATION

Higher-operating frequency allows the usage of smallinductor values. Operating at higher frequenciesincreases the switching losses in the power MOSFET,thus decreasing the efficiency of the converter.

Choosing the operating frequency should be doneaccordingly.

For the given specifications, the inductance value iscalculated as shown in Equation F-3 below.

EQUATION F-3: BOOST CONVERTER INDUCTOR CALCULATION

The root mean square (RMS) inductor current is givenby calculations in Equation F-4.

EQUATION F-4: RMS INDUCTOR CURRENT

Dmax

Vout Vin min –

Vout------------------------------------------

26.8-1226.8

------------------ 0.5522= = =

L H Vo D 1 D–

2 106

Io r f------------------------------------------------------------=

Where,

r = ripple current ratio (r = 0.3 as per design require-ments)

L H Vo D 1 D–

2 106

Io r f------------------------------------------------------------

26.8 0.55 1 0.55– 2 10

64.5 0.3 250000

----------------------------------------------------------------------------- 8.844 H= = =

ILrms

Vout Vsynch+ Iout max

Vin min ---------------------------------------------------------------------------- 26.8 0.8+ 4.5

12-------------------------------------------- 10.35A= =


AN2321

The ripple inductor current is given by calculations inEquation F-5.

EQUATION F-5: INDUCTOR RIPPLE CURRENT

The peak inductor current is given by Equation F-6.

EQUATION F-6: PEAK INDUCTOR CURRENT

The inductor chosen is 15 μH with peak current of 18Aand RMS current of 20A. Coilcraft part no. SER2918H-153.

MOSFET Selection

The selection of the MOSFET for the boost convertermust be based on the following parameters:

1. As the MOSFET conducts the inductor currentduring the ON time, it must have low RDSON tominimize conduction losses and to improveefficiency.

2. The drain current rating of MOSFET should behigh, as it carries the peak inductor current.

3. The drain to source voltage rating of theMOSFET should be greater than the outputvoltage and also have margins for some spikes.

4. The gate charge of the MOSFET should be lowin order to decrease switching losses. A highgate charge would mean greater transition timesbetween ON and OFF times of the MOSFET. Athigher frequencies it will result in greaterconduction losses, thus decreasing theefficiency of the converter.

Diode Selection

The selection of the diode for the boost converter isbased on the following parameters:

1. The forward voltage drop should be as low aspossible to keep the conduction losses to aminimum. The power dissipated in the diode canbe calculated as a product of forward voltagedrop and load current.

2. The current handling capability should be atleast two times the maximum output current.

Output Capacitor Selection

The main function of the output capacitor is to providethe load current during the ON time of the switch, andto filter the output voltage ripple. During the ON time, noenergy from the input is transferred to the output.Hence, there will be a voltage drop.

The most important parameter to be considered duringthe selection of the capacitors is the Effective SeriesResistance (ESR) of the capacitor. There will also besome drop caused by the Equivalent Series Resistanceof the output capacitor. Low ESR capacitors will berequired at the output to reduce this ripple. In additionto that, ESR also affects the loop stability requirements.A parallel network of capacitors can be used at theoutput, which not only helps in lowering the ESR, butalso divides the current ripple. A combination ofelectrolytic and ceramic capacitors is used at theoutput.

For the given specifications, the minimum capacitancevalue is calculated as shown in Equation F-7.

EQUATION F-7: MINIMUM OUTPUT CAPACITANCE

Any capacitance above the calculated minimum valuecan be used to suit the specifications of the converter.More capacitors can be used in parallel to get a lowerESR, and ensure that the ripple is at its minimum.

Input Capacitor Selection

Input capacitors are added to reduce the input voltageripple. As the ESR is low for ceramic capacitors, theyare placed at the input to reduce the ripple.

ILripple

ILrms Iripple ratio

100-----------------------------------------------------------

10.35 3100

----------------------- 3.105A= = =

ILpeak ILrms

ILripple2

----------------------+ 10.353.105

2-------------+ 11.9025A= = =

Cout min

Iout D

Fs Vout-----------------------------

4.5 0.55

250000 300 103–

---------------------------------------------------- 33 F= = =


AN2321

Measurement Circuits

The PCMC methodology employs the peak inductorcurrent as the basic parameter for control. Transistor-based current mirror circuit is used for inductor currentmeasurement, as shown in Figure F-1.

FIGURE F-1: CURRENT MEASUREMENT USING CURRENT MIRROR

The voltage corresponding to the current is given bythe equation shown in Figure F-1. The current (I)represents the inductor current.


AN2321

APPENDIX G: HARDWARE DESIGN PCMC BUCK CONVERTER 12V/2A

Selection of Each Component in the Buck Converter Power Circuit

For a buck converter, the external power circuitcomponent selection depends upon parameters likeinput voltage, output voltage, switching frequency,maximum ripple current and maximum output voltageripple.

Inductor Selection

The maximum inductor current ripple and the switchingfrequency determine the size of the inductor. Theinductor size is inversely proportional to the switchingfrequency and the inductor current ripple. Higher-operating frequency allows the use of small inductorvalues. But operating at higher frequencies increasesthe switching losses in the power MOSFET, thusdecreasing the efficiency of the converter. So the trade-off should be made in the inductor size and switchingfrequency. As the inductor current ripple increases, thesize of the filter capacitor also increases, in order tomeet the required output voltage ripple specifications.Generally, the maximum inductor current ripple of20-40% of the full-load current will be considered. Theinductor with less DCR (DC resistance) should beselected to minimize the conduction losses due to theinductor winding resistance.

The minimum inductance to ensure ContinuousConduction mode is given by Equation G-1.

EQUATION G-1: INDUCTANCE CALCULATION BOOST CONVERTER

For the specifications given in Table 1, consideringVIN = VIN(max), the inductor value should be(Equation G-2):

EQUATION G-2: CALCULATED INDUCTANCE

The inductor peak current is given by Equation G-3.

EQUATION G-3: INDUCTOR PEAK CURRENT

The inductor selected in this design is the CoilcraftMSS1210-473 of value 47 µH, having a saturationcurrent rating of 4.6A, a RMS current rating of 4A anda maximum DCR of 0.056Ω.

Control MOSFET Selection

Control MOSFET will be conducted during the TON

period of the switching period. The drain current ratingof the MOSFET should be high, as it carries the peakinductor current. Both control and synchronousMOSFETs will be exposed to the input voltage at somepoint during the switching cycle, so both must have adrain-source breakdown voltage greater than VIN.

There are two types of power losses associated withMOSFETs, switching losses and conduction losses.Switching losses will come up during the turn ON andOFF events of the MOSFET. Conduction loss willappear when the MOSFET is turned on. The total gatecharge parameter of the MOSFET will determine theswitching losses. The higher the total gate charge is,the greater will be the transition times between the ONand OFF states of the MOSFET. Therefore, at higherfrequencies it will result in greater switching losses,thus decreasing the efficiency of the converter. AMOSFET with very low RDSON should be selected tominimize conduction losses and to improve efficiency.

The control MOSFET selected in this design isDMT6016LSS-13, having a drain-source breakdownvoltage of 60V, a continuous drain current rating of9.2A, an RDSON of 18 mΩ, and a total gate charge of17 nC.

Voltage cross inductor is given by:

VL LdILdt

---------=

Voltage cross inductor in TOFF time:

VL TOFF LIL

1 D– TS------------------------------- VOUT= =

Hence:

LVOUT 1 D–

IL FS-----------------------------------------

VOUT 1 VOUT VIN–

IL FS--------------------------------------------------------------------= =

LVOUT 1 VOUT VIN–

IL FS--------------------------------------------------------------------

12 1 12 29– 0.4 2 250000------------------------------------------ 35.17 H= = =

IL Peak IOUT

IL2

---------+ IOUT

40% of IOUT2

----------------------------------+ 20.4 2

2----------------+ 2.4A= = = =


AN2321

Synchronous MOSFET Selection

Synchronous MOSFET will be conducted during theTOFF period of the switching cycle. Peak current ratingand drain-source breakdown voltage of thesynchronous MOSFET will be similar to that of controlMOSFET.

When the control MOSFET switches off, the voltage atthe MOSFET side of the inductor goes negative, thusthe voltage across the synchronous MOSFET is nearlyzero when the synchronous MOSFET switches on.

Therefore, the switching losses of the synchronousMOSFET are negligible, so the total gate chargespecification of the synchronous MOSFET isnegligible. Only the RDSON characteristic of thesynchronous MOSFET is important.

The synchronous MOSFET selected in this design isthe same as the control MOSFET.

Output Capacitor Selection

The output capacitor filters out the ripple content of theinductor current and delivers the stable output voltageto the load. The output capacitor also has to ensure thatload steps at the output can be supported before theregulator/feedback control loop is able to react. LowESR capacitors will be required at the output to reduceripple in the output voltage. In addition to that, the ESRalso affects the feedback control loop stabilityrequirements. There are different types of capacitorsavailable in the market, such as multilayer ceramic,tantalum and aluminum electrolytic. Multilayer ceramiccapacitors have lower ESR, and the aluminumelectrolytic capacitors have higher ESR. But multilayerceramic capacitors are available up to 100 µF only. Acombination of both electrolytic and multilayer ceramiccapacitors can be used at the output to reduce theoverall ESR of the parallel combination, and toincrease the output capacitance. Capacitors have ahigh tolerance, usually below its nominal value.Capacitors should be selected based on the followingparameters:

• Maximum Voltage• Maximum Ripple Current• ESR Ratings at the Temperature• Frequency of the Application• De-ratings for DC Bias and Temperature

The capacitance required at the output can becalculated using Equation G-4.

EQUATION G-4: OUTPUT CAPACITOR BUCK CONVERTER

Any capacitance above the calculated minimum valuecan be used to suit the specifications of the converter.More capacitors can be paralleled-up to get lower ESR,and ensure that the ripple is at its minimum.

The output capacitor selected in this design is a parallelcombination of an C3225X7R1E106M250AC (10 µF),a C3225X7R1H105M160AA (1 µF) TDK MLCCcapacitor, and a PLV1H390MDL1 (39 µH) aluminumelectrolytic capacitor.

COUT min

IOUT D

FS VOUT----------------------------------

2 12 20

250000 200 103–

---------------------------------------------------- 24 F= = =


Note the following details of the code protection feature on Microchip devices:

• Microchip products meet the specification contained in their particular Microchip Data Sheet.

• Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the intended manner and under normal conditions.

• There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data Sheets. Most likely, the person doing so is engaged in theft of intellectual property.

• Microchip is willing to work with the customer who is concerned about the integrity of their code.

• Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not mean that we are guaranteeing the product as “unbreakable.”

Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of ourproducts. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such actsallow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act.

Information contained in this publication regarding deviceapplications and the like is provided only for your convenienceand may be superseded by updates. It is your responsibility toensure that your application meets with your specifications.MICROCHIP MAKES NO REPRESENTATIONS ORWARRANTIES OF ANY KIND WHETHER EXPRESS ORIMPLIED, WRITTEN OR ORAL, STATUTORY OROTHERWISE, RELATED TO THE INFORMATION,INCLUDING BUT NOT LIMITED TO ITS CONDITION,QUALITY, PERFORMANCE, MERCHANTABILITY ORFITNESS FOR PURPOSE. Microchip disclaims all liabilityarising from this information and its use. Use of Microchipdevices in life support and/or safety applications is entirely atthe buyer’s risk, and the buyer agrees to defend, indemnify andhold harmless Microchip from any and all damages, claims,suits, or expenses resulting from such use. No licenses areconveyed, implicitly or otherwise, under any Microchipintellectual property rights unless otherwise stated.

2016 Microchip Technology Inc.

Microchip received ISO/TS-16949:2009 certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona; Gresham, Oregon and design centers in California and India. The Company’s quality system processes and procedures are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping devices, Serial EEPROMs, microperipherals, nonvolatile memory and analog products. In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001:2000 certified.

QUALITY MANAGEMENT SYSTEM CERTIFIED BY DNV

== ISO/TS 16949 ==

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The Microchip name and logo, the Microchip logo, AnyRate, AVR, AVR logo, AVR Freaks, BeaconThings, BitCloud, CryptoMemory, CryptoRF, dsPIC, FlashFlex, flexPWR, Heldo, JukeBlox, KEELOQ, KEELOQ logo, Kleer, LANCheck, LINK MD, maXStylus, maXTouch, MediaLB, megaAVR, MOST, MOST logo, MPLAB, OptoLyzer, PIC, picoPower, PICSTART, PIC32 logo, Prochip Designer, QTouch, RightTouch, SAM-BA, SpyNIC, SST, SST Logo, SuperFlash, tinyAVR, UNI/O, and XMEGA are registered trademarks of Microchip Technology Incorporated in the U.S.A. and other countries.

ClockWorks, The Embedded Control Solutions Company, EtherSynch, Hyper Speed Control, HyperLight Load, IntelliMOS, mTouch, Precision Edge, and Quiet-Wire are registered trademarks of Microchip Technology Incorporated in the U.S.A.

Adjacent Key Suppression, AKS, Analog-for-the-Digital Age, Any Capacitor, AnyIn, AnyOut, BodyCom, chipKIT, chipKIT logo, CodeGuard, CryptoAuthentication, CryptoCompanion, CryptoController, dsPICDEM, dsPICDEM.net, Dynamic Average Matching, DAM, ECAN, EtherGREEN, In-Circuit Serial Programming, ICSP, Inter-Chip Connectivity, JitterBlocker, KleerNet, KleerNet logo, Mindi, MiWi, motorBench, MPASM, MPF, MPLAB Certified logo, MPLIB, MPLINK, MultiTRAK, NetDetach, Omniscient Code Generation, PICDEM, PICDEM.net, PICkit, PICtail, PureSilicon, QMatrix, RightTouch logo, REAL ICE, Ripple Blocker, SAM-ICE, Serial Quad I/O, SMART-I.S., SQI, SuperSwitcher, SuperSwitcher II, Total Endurance, TSHARC, USBCheck, VariSense, ViewSpan, WiperLock, Wireless DNA, and ZENA are trademarks of Microchip Technology Incorporated in the U.S.A. and other countries.

SQTP is a service mark of Microchip Technology Incorporated in the U.S.A.

Silicon Storage Technology is a registered trademark of Microchip Technology Inc. in other countries.

GestIC is a registered trademark of Microchip Technology Germany II GmbH & Co. KG, a subsidiary of Microchip Technology Inc., in other countries.

All other trademarks mentioned herein are property of their respective companies.

© 2016, Microchip Technology Incorporated, All Rights Reserved.

ISBN: 978-1-5224-1217-5

DS00002321A-page 35


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AN2321 - ww1.microchip.com