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An optical ASK and FSK phase diversity transmission system
Citation for published version (APA):Boom, van den, H. P. A., Etten, van, W. C., Krom, de, W. H. C., Bennekom, van, P. K., Huijskens, F., Niessen,L., & Leijer, de, F. (1992). An optical ASK and FSK phase diversity transmission system. (EUT report. E, Fac. ofElectrical Engineering; Vol. 92-E-268). Technische Universiteit Eindhoven.
Document status and date:Published: 01/01/1992
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An Optical ASK and FSK Phase Diversity Transmission System
by H. van den Boom and W. van Etten, W.H.C. de Krom, P. van Bennekom, F. Huijskens, L. Niessen, F. de Leijer.
EUT Report 92-E-268 ISBN 90-6144-268-0
Qr' or a .,
Eindhoven University of Technology Research Reports
Eindhoven University of Technology
ISSN 0167-9708
Faculty of Electrical Engineering
Eindhoven, The Netherlands
Coden: TEUEDE
AN OPTICAL ASK AND FSK PHASE DIVERSITY
TRANSMISSION SYSTEM
by
H. van den Boom and W. van Etten, W.H.C. de Krom,
P. van Bennekom, F. Huijskens, L. Niessen, F. de Leijer.
EUT Report 92-E-268
ISBN 90-6144-268-0
Eindhoven
December 1992
CIP-GEGEVENS KONINKLIJKE BIBLIOTHEEK, DEN HAAG
Optical
An optical ASK and FSK phase diversity transmission system I by H. van den Boom ... let al.]. - Eindhoven: Eindhoven University of Technology, Faculty of Electrical Engineering. - Fig., tab. - (EUT report, lSSN 0167-9708 ; 92-E-268) Met lit. opg. ISBN 90-6144-268-0 NUGI 832 Trefw.: optische datatransmissie-systemen.
ABSTRACT
This report describes the results of the contribution of the Telecommunications Division
of the Faculty of Electrical Engineering, Eindhoven University of Technology to the lOP
Electro Optics, cluster I project: • Phase Diversity System'.
Specifications of subsystems have been derived, together with tolerances and consequen
ces of these tolerances for the final system performance. For the optical network of the
phase diversity receiver a manufacturing set-up for {3x3} fused biconical taper fiber
couplers has been developed. In order to characterize planar optical networks, a set-up
has been constructed to measure the phase relations at 1523 nm. The optical frequency of
the local oscillator laser has to be locked on to the frequency of the received optical
signal. A detailed description of this locking circuit is given. A complete optical {3x3}
phase diversity transmission system has been developed that can be used as a testbed for
subsystems. The sensitivity of the receiver at a BER of 10-' is -47.2 dBm, which is 4.2
dB better than the value of the specificatons.
Boom, H. van den and W. van Etten, W.H.C. de Krom, P. van Bennekom,
F. Huijskens, L. Niessen, F. de Leijer.
AN OPTICAL ASK AND FSK PHASE DIVERSITY TRANSMISSION SYSTEM.
Faculty of Electrical Engineering, Eindhoven University of Technology,
The Netherlands, 1992. EUT Report 92-E-268.
Address of the authors:
Telecommunications Division
Faculty of Electrical Engineering
Eindhoven University of Technology
P.O. Box 513, 5600 MB Eindhoven, The Netherlands
- IV -
CONTENTS
I INTRODUCTION I
2 ANAL YSIS OF THE PHASE DIVERSITY SYSTEM 3
2.1 Coherent system design 3
2.2 Phase diversity reception 4
2.3 Description of the transmission system 6
2.4 Sensitivity penalty due to non-ideal components 8
2.4.1 Impact of the laser phase noise and IF/LP bandwidth 8
2.4.2 Impact of the laser intensity noise 1l
2.4.3 Impact of phase mismatch and gain imbalance 13
2.4.4 Impact of the extinction ratio of the external modulator 15
3 THE TRANSMITTER AND LOCAL OSCILLATOR LASERS 20
3.1 The DFB lasers 20
3.2 Laser fiber coupling 22
3.3 Optical isolation 24
3.4 The temperature controllers 26
3.5 The current control circuit 28
4 THE TRANSMITTER 31
4.1 The optical amplitude modulator 31
4.2 The FSK modulator 33
5 THE RECEIVER 35
5.1 The optical network 35
5.1.1 The 3x3 fiber coupler 35
5.1.2 Phase diversity network measurement set-up 38
5.2 The PINFET OlE converter module 44
5.3 The IF section 46
5.4 The frequency controller 52
5.4.1 Principles of the frequency locked loop 53
5.4.2 The realized frequency locked loop 54
5.4.3 Conclusions 55
5.5 The clock and data recovery circuit 55
6 EXPERIMENTAL RESULTS
6.1 IF spectra
6.2 BER measurements
6.3 The video codec
REFERENCES
List of abbreviations
APPENDIX I Specifications
- v -
APPENDIX 2 Data sheets of the Philips DFB lasers
APPENDIX 3 Data of the GEC optical amplitude modulator
APPENDIX 4 Circuit diagrams
APPENDIX 5 Source code listings of the frequency locked loop
59
59
63
64
67
71
72
73
79
80
92
1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1
I
1
Chapter 1
Introduction
Optical coherent detection offers the benefits of a larger receiver sensitivity (10 a 20 dB)
and of a more dense multiplexing of channels in the frequency domain. Those advantages
are to be exploited for subscriber networks in the Integrated Broadband Communication
Network (lBCN).
A phase diversity receiver is extremely tolerant for laser linewidths compared with an
optical homodyne receiver and requires a smaller bandwidth of the IF electronics
compared with a heterodyne receiver.
The history of the project is as follows.
In September 1987 the steering committee SGOG (Stuurgroep Gemeenschappelijk Onder
zoek Glasveze1communicatie in Nederland) initiated a research project entitled "Integrated
Optics voor het IBCN" and a proposal was submitted to the Ministry of Economic
Affairs, asking for financial support. This ministry, however, placed the proposal in the
broader research field Electro-optics and asked the IOP/IC-technology committee for
advice. This committee in its turn set up a sub-committee lOP Electro-Optics advisory
board. Based on letters of intend of the SGOG partners, the board issued a workplan on
electro-optics and invited, among others, the partners to submit project proposals in
accordance with the subjects defined in the workplan.
A combination of research groups from PTT Research Neher Laboratory, Eindhoven
University of Technology, Delft University of Technology, University of Twente and
Physics and Electronics Laboratory TNO submitted a proposal which was deduced from
the former SGOG proposal and which was entitled "Opto-electronic IC's for the Integrat
ed Broadband Communication Network (lBCN)". This proposal, although slightly
adapted, has been approved by the committee and was named "Phase Diversity System" .
2
The contracts have been undersigned in October 1989, which meant the starting point of
the execution of the project. Both written and oral progress reports have been delivered
on February 15 1990, September 6 1990, February 14 and September 5 1991.
3
Chapter 2
Analysis of the phase diversity system
In this chapter the realized ASK (Amplitude-shift Keying) and FSK (Frequency-shift
Keying) phase diversity transmission system is described and the performance conse
quences of non-ideal components are discussed. Specifications of subsystems and system
components are given. First the design of the system is described.
2.1 Coherent system design
To gain full advantage of the huge transmission capacity of optical fibers the use of
coherent optical reception is the best solution. Optical coherent detection offers two
important advantages with respect to conventional intensity modulation/direct detection:
1) improved receiver sensitivity;
2) greatly enhanced frequency selectivity.
In optical coherent detection the received signal is mixed with the signal of a local
oscillator. In this wayan IF signal (Intermediate Frequency) is obtained. The spectrum of
the IF signal is equal to the convolution of the spectrum of the received and local
oscillator signal. Just as in radio communication reception, if the local oscillator is phase
locked with the received optical signal this is called homodyne detection, if not this is
called heterodyne detection. With the present day trend towards higher bit rates, a
conventional heterodyne receiver is not practical because of the large bandwidth required
for the photodiodes, preamplifiers, filters and demodulators of the receiver (3-5 times the
bit rate) [1]. For homodyne receivers the baseband bandwidth is sufficient, and it has the
additional advantages of lower noise and a smaller frequency band in optical coherent
multicarrier systems. However a homodyne receiver requires phase-locking between the
transmitter and local oscillator waves, which is hard to realize. With a heterodyne phase
diversity receiver it is possible to choose a lower IF compared with a conventional
4
heterodyne receiver so the bandwidth required of the IF electronics is lower. It is even
possible to choose an IF of zero, which is often called a pseudo-homodyne receiver. This
receiver does not use phase-locking and the laser linewidth requirements are relaxed as
compared to those of a synchronous receiver [2] [3].
The developed transmission system uses a heterodyne phase diversity detection scheme.
The modulation scheme is either 140 Mb/s ASK or 140 Mb/s FSK. At the receiving end,
for FSK the single filter demodulation method is used, so the same receiver can be used
for the ASK as well as for the FSK modulation scheme. For ASK modulation an
expensive external optical amplitude modulator has to be used. FSK modulation can be
realized by direct modulation of the transmitter laser bias current which is easier,
however the transmission bandwidth for FSK modulation is much greater than for ASK
modulation.
2.2 Phase diversity reception
With direct detection systems the received optical signal is directly applied to a
photodiode which generates a current proportional to the optical power received. The
coherent receiver adds a locally generated optical wave to the optical received wave. This
combined signal is applied to the photodiode. Let us suppose, that both the signal wave
and the local oscillator wave are polarized linearly in the same direction and let us denote
these two waves by respectively,
(2.1)
(2.2)
The photodiode current is proportional to the power of the sum of the waves which is
(2.3)
5
In this expression the third term is the most interesting one. This term contains all the
information of the received signal, namely the amplitude Po, the frequency Wo and the
phase ,I>o(r). The magnitude of this third term can be increased by boosting the local
oscillator power PI' The first term Po is the power of the signal received which provides
the photocurrent for direct detection and is weak compared to the third term. The second
term PI from the power of the local oscillator is a DC term.
In case of ASK modulation the received optical power Po is modulated and in case of
FSK modulation the frequency of the received optical signal Wo is modulated. If for
instance the optical frequencies of both lasers are the same so WO-WI =0, the third term
will become zero if cPo(t)-,p'(r) = 7r12, rendering the system useless. This problem can be
solved by using a phase diversity detection scheme. In Figure 2.1 a general diagram of an
optical coherent phase diversity receiver with low pass filters (LP) and square law or
envelope detectors (D) is shown. In this diagram a {3x3} optical network, for instance a
fused fiber coupler, is used. Alternatively, a planer optical network can be used, and then
a {4x4} optical network seems to be more suitable.
~
P, Optical
P, hybrid
"
Fig. 2.1: General diagram of an optical coherent phase diversity receiver with low pass filters
(LP) and square law or envelope detectors (D).
When each output signal is detected by a photodiode, de signal components of the N
currents (N = 3,4) have 27rIN phase difference between them, provided that each output
has the same output amplitude. When the currents are squared, a baseband term arises, as
well as a double frequency term. Adding the squared current terms cancels out the double
6
frequency terms, whereas the baseband terms increase the signal in a single branch N ti
mes. In this way fading due to the phase noise of the lasers is avoided.
2.3 Description of the transmission system
In Pig. 2.2. a block diagram of the system is shown. The upper part of the figure is the
transmitting-end and the lower part is the receiving-end of the system. The transmitter
and the receiver are connected by a standard single mode fiber (SMP).
Because the optical frequency of the laser is very susceptible to variations in temperature
a temperature controller is used for stabilization. The laser diode is mounted on a peltier
element and the temperature is detected by an NTC-resistor. Because the laser diodes are
very sensitive to optical reflections a 90 dB optical isolator is inserted right behind the
taper. In case of FSK modulation the output of the isolator is directly connected to the
transmission fiber (SMF). Por ASK modulation the isolator is connected to the transmis
sion fiber via the optical modulator.
For FSK modulation the digital input signal is connected to the current control circuit
which modulates the current of the transmitter laser diode. This current control circuit
also provides a stable DC current for the laser diode. By modulating the current, the
output power of the laser is modulated as well, but this amplitude modulation is negligi
ble. For ASK modulation an external optical amplitude modulator is used. In this case it
is not possible to modulate the current, because by modulating the current the optical
frequency of the laser is modulated as well.
At the receiving-end the optical signal received, is mixed with the optical output signal of
the local oscillator laser diode by the {3x3} optical network. Delft University of Technol
ogy, Department of Electrical Engineering, Laboratory of Telecommunication and
Remote Sensing Technology (TUD/TTT) has developed a planar {4x4} optical network.
A phase measurement set-up for these planar networks has been developed. The local
oscillator laser diode is optically isolated as well and both the temperature and the current
are stabilized in the same way as the transmitter laser diode. The polarization of the
output of the local oscillator laser is controlled by a hand polarization controller. The
optical signals are detected by three PIN-PET modules. These PIN-FET modules consist
7
of a photodiode and a front-end amplifier. In order to obtain a stable difference frequency
or intermediate frequency (IF), the optical frequency of the local oscillator laser is
controlled by the frequency controller, which controls the laser current. The output
signals of the PIN-FET modules are filtered by the IF lowpass filters (LPF) and demodu
lated by square law detectors. The signals of the three branches are added and filtered by
a post detection filter (LPF). The data clock and data are recovered by a phase-locked
loop and the sampler, respectively.
current control circuit
FSK-IN j
temperature controller
transmitter laser d·ode I
.~ peltier element
isolator optical ::;;;. modulator
ASK-IN J TRANSMITTER
PHASE DIVERSITY RECEIVER
Fig. 2.2: Block diagram of the phase diversity system.
8
2.4 Sensitivity penalty due to non· ideal components
2.4.1 bnpact of laser phase nolore and the IF/LP bandwidth
The spontaneous emission causes phase fluctuations in the laser output. This phase noise
decreases the coherence time of the laser light. Limited coherence time manifests itself as
a broadening of the laser spectrum. The laser spectrum, which under ideal conditions
consists of one sharp line, can be shown to have a Lorentzian line shape due to the
presence of phase noise.
LASER SPECfRUM
-5
~
~ -10 '0
.5 ~
M
~ -15 a
"-
-20
-25_5':---_':-4 --_73 ---"::"2---":-1--~0- 1 2 3 4 5
Frequency (in Hz) X108
Fig. 2.3: Example of an equivalent baseband laser spectrum due to laser phase noise.
The laser spectrum is given by the following equation [4-5]
21r[(.1v/2)' + (f-vJ'] (2.4)
where Vo is the resonant frequency of the laser cavity and .1v is the laser linewidth, which
9
is defined as the 3.0 dB linewidth, also called the Full-Width-Half-Maximum (FWHM)
laser linewidth. It is obvious that laser phase noise is a nonnegligible impairment in phase
modulated communication systems. However, even coherent systems employing
amplitude modulation suffer from sensitivity degradation due to the laser phase noise.
This has to do with the unavoidable filtering of the IF signal received. Filtering of phase
noise can lead to a conversion of laser phase noise into amplitude noise, which for
amplitude modulation is an important impairment [6].
Laser phase noise also causes a broadening of the IF spectrum, and therefore, the
bandwidth of the IF filter should be increased in order to accommodate the necessary
signal power. The latter implies that the noise bandwidth is extended and the impact of
the shot and thermal noise is increased at the expense of receiver sensitivity [7].
If the intersymbol interference may be neglected, the power spectrum of the signal at the
input of the IF filter is given by the following convolution
(2.5)
where SM(f) is the modulation spectrum. In case of ASK modulation SM(f) is given by [8]
sin2( 7rTof/-/'» TTl/-/)'
+ sin2( 7rTl/+/'» 1 TToif+/'Y J'
(2.6)
where A is a constant, and!, represents the carrier frequency. Equation (2.6) implies that
when Av increases, the power spectrum 0(1) of the signal at the input of the IF filter
broadens. Consequently the bandwidth of the IF filter should be increased in order to
accommodate the same amount of signal power as in case without laser phase noise.
However, this also increases the (shot) noise bandwidth, and therefore, results in a
sensitivity penalty. The loss in receiver sensitivity (SNR) due to the increase of the noise
bandwidth, is depicted in Fig. 2.4 versus the normalized laser linewidth AvTo. This for a
in the frequency domain rectangular IF filter, which passes 95 % of the incoming signal
power.
10
7 ---_ ...... / . ./--
6
5 ill , , TI ./
/ c 4 /
co /
>. / ~
Iii 3 c / IlJ
CL
2
OL-~~--~~--~~--~~--~~~~~~--~~
0.00 0.30 0.60 0.90 1.20 1.50
Normalized laser linewidth
Fig. 2.4: Loss of SNR versus the normalized laser line width AvTo.
When large IF filter bandwidths are used in ASK receivers to overcome the spectrum
broadening due to laser phase noise, a post-detection filter, with a bandwidth much
smaller than the IF filter bandwidth, is usually applied to improve the receiver sensitivity
[2]. By matching of the IF filter and the post -detection filter, an optimal IF filter
bandwidth can be found for which the sensitivity degradation is minimal. Further
increasing of the bandwidth results in an excess sensitivity degradation, since too much
(shot) noise is passed. On the other hand, if the IF filter bandwidth is too small, a BER
floor seriously degrades the receiver performance, due to the conversion of laser phase
noise into intensity noise and vice versa [9]. For ASK systems the BER floor depends on
the relation between the normalized IF filter bandwidth (BifTo) and the normalized laser
linewidth (Ll.vTo). The larger BifTo, the lower the BER floor.
For optimized IF and post-detection filtering, phase diversity ASK receivers have the
same phase noise performance as heterodyne ASK receivers, and they can tolerate the
same amount of laser phase noise [4,10,11]. For ASK receivers, the sensitivity penalty,
due to laser phase noise versus the normalized laser linewidth, is given in Fig. 2.5 for a
11
BER=IO-9, and optimized IF and post-detection filtering [10,11].
3.00 ~--- ------1
2.50
~ 2.00
g >, 1.50
== III C OJ
Q 1.00
0_50
I , 0.00 ~ __ L-~ __ ~ __ ~ __ ~ ___ -L __ ~ __ ~ __ ~' ~
0 0.5 1 1-5 2 25 Normalized laser' linewidth
Fig. 2.5: Sensitivity penalty versus tJ.VTo for ASK receivers (SER = 10-9).
2.4.2 Impact of the laser intensity noise
Beside coherent light due to the stimulated recombination of Electron Hole Pairs (EHP's)
in the laser cavity, a semiconductor laser also emits socalled incoherent light generated by
the spontaneous recombination of EHP's in the laser cavity. For semiconductor lasers, the
spontaneous emission of photons in the laser cavity gives onset to noise problems. This
spontaneous emission is the main cause for intensity noise. Due to the small device size,
the optical energy storage in the cavity is small, and therefore, the sensitivity to the
perturbations of spontaneous emission in semiconductor lasers is usually significant. The
influence of the spontaneous emission could be reduced by increasing the size of the laser
cavity. Intensity noise manifests itself as a random fluctuation of the optical power, and is
especially pronounced near the threshold and diminishes as the laser bias current is
increased. Statistically, intensity noise behaves like shot noise, after detection by a
photodiode which implies that it is also Gaussian distributed and has a flat power
spectrum (white noise) over a wide frequency range. Intensity noise is usually character-
12
ized by the so-called Relative Intensity Noise (RIN). The RIN spectral density is defined
as the variance of the optical power fluctuations, divided by the mean optical
power squared in 1Hz bandwidth (dBlHz).
The sensitivity degradation for the {2x2} and {3x3} phase diversity ASK receiver has
been analyzed in [9]. The sensitivity penalty has been computed with respect to the
performance of an ideal heterodyne ASK receiver. The bit rate is taken to be ISO Mbitls
and the influence of the phase noise has been neglected. The results are depicted in
Fig. 2.6 for three practical RIN levels -ISO, -135, and -125 dB/Hz, respectively.
30 -------_. __ ._---------
ill 25 -125
D
g 20
>. " ~
" <Il C Qi 15 0.
-- U5
>. ~
;; 10 .,
iii c Q!
(j) -150 ...... ----
5 -----I
~.-~--=---:::.- - --I 0 "---__ '-_---' __ ---' __ ----.l. ___ --"-___ J -30 -25 -20 -15 -10 -5 o
Local oscillator power (in dBm)
Fig. 2.6: Sensitivity penalty versus P, for several RIN values for a {2x2} ASK receiver
{ __ I and a {3x3} ASK receiver '- __ I with both an optimal threshold level.
For both the {2x2} and the {3x3} phase diversity ASK receiver an optimum value for the
local oscillator power (PJ is found, for which the sensitivity penalty with respect to the
ideal heterodyne ASK receiver is minimal. This value depends on the position of the
threshold level and the local oscillator laser RIN. For each combination of PL and a RIN
level, the threshold level ought to be optimized in order to minimize the BER, implying a
minimum sensitivity penalty. For an optimum threshold value and a BER= 10-9, the {3x3}
13
receiver outperforms the {2x2} receiver for a local oscillator power larger than the
optimum value of P" The reverse is true for values of PL smaller than this optimum
value.
2.4.3 Impact of phase mismatch and gain imbalance
The key component of all phase diversity receivers is an optical hybrid that provides the
means of recovering the modulated optical signal. All phase diversity receivers are
sensitive to gain imbalance and an aberration in the phase relations between the different
receiver branches. The aberration in the phase relations can originate from the optical
hybrid, and the gain imbalance in the receiver branches can be caused, firstly, by an
unequal amplification at the different IF stages, or secondly, by an unequal power
distribution over the outputs of the optical hybrid. In [9] the BER's have been calculated
for a {2x2} and a {3x3} phase diversity ASK receiver, using non-Gaussian Probability
Density Functions (PDF's) for the signals at the threshold comparator, and various values
for the percentage of gain imbalance and the phase mismatch.
The gain imbalance has been modeled by placing an additional amplifier in each receiver
branch. One of the receiver branches is taken as a reference with an amplification factor
equal to one. The amplification factor of the other branch in the {2x2} phase diversity
receiver is given by (1 +0), introducing a gain imbalance of 100*0% (-1 <0< 1). For the
{3x3} phase diversity ASK receiver, the gain imbalance is introduced in the same way
except for one difference. The percentage of gain imbalance is assumed to be symmetri
cal.
The phase mismatch for the {2x2} receiver is modeled by introducing an extra phase shift
term in one of the two relations for the photocurrents at the IF stage, taking the phase of
one receiver branch as a reference. For the {3x3} receiver, the phase mismatch is
modeled by introducing an extra phase shift term, in two out of the three relations for the
photocurrents at the IF stage, taking one receiver branch as a reference.
In [9] upper-bounds for the gain imbalance and the phase mismatch in both receiver
structures have been derived. The criteria was given a certain percentage of gain
imbalance the maximum allowable phase mismatch, which increases the signal power P
required to achieve a BER = 10·' by maximal 0.5 dB. In the Tables 2.1 and 2.2, maxi
mum values for the phase mismatch are presented, for a gain imbalance of +/-5% and
14
+1-10%, respectively.
Table 2.1
Maximum allowable phase mismatch for a {2x2} and a {3x3} phase diversity ASK receiver as a function of the gain imbalance.
BER = 10-9, non-optimal threshold, penalty margin = 0.5 dB
Gain imbalance 0% 5% 10% -5% -10%
{2x2} ASK receiver ±6.0° ±7.0° ±S,SO ±4.0° -----
{3x3} ASK receiver non-symmetrical ±9.0° ± 11.8° ---- ±4,SO -----configuration
{3x3} ASK receiver 5's" 6.2° ---- 4.0° -----
symmetrical -5.0° _7.0° _1.5° configuration ---- -----
---- = not possible within a penalty margin of 0.5dB
± = for positive and negative values of the phase mismatch
Tabel2.2
Maximum allowable phase mismatch for a {2x2} and a {3x3} phase diversity ASK receiver as a function of the gain imbalance. BER = 10-9
, optimal threshold, penalty margin = 0.5 dB
Gain imbalance 0% 5% 10% -5% -10%
{2x2} ASK receiver ±8.5° ±8.0° ±6's" ±8.0° ±6.2°
{3x3} ASK receiver non-symmetrical ±13.0° ±13.0° ± 12.0° ±12.0° ±10.0° configuration
{3x3} ASK receiver 8.0° 7.0° 5's" 9.1 ° 9.0° symmetrical
-7.8° _7.8° -7.8° -5.8° _4.0° configuration
± = for positive and negative values of the phase mismatch
15
In [9] has been concluded, that in comparison with the ideal situation, without any of
these imperfections, the signal value at the input of the threshold comparator becomes a
function of the radial offset frequency Wif. This results in a time varying character of the
BER, which leads to a sensitivity degradation. The sensitivity degradation has been
calculated with respect to the ideal {2x2} or (3x3} phase diversity ASK receiver for a
BER= 10~9. Since ASK demodulation highly depends on the threshold value, the BER's
have been calculated for a threshold value optimized for the case without any imperfec
tions, and for a threshold value optimized for a certain combination of gain imbalance and
phase mismatch. By optimizing the threshold level, the sensitivity penalty can be
significantly reduced.
2.4.4 Impact of the extinction ratio of the external modulator
The ASK modulation of the optical signal wave transmitted is usually obtained by means
of an external amplitude modulator [12]. The modulator switches the amplitude of the
optical wave transmitted between two levels, corresponding to the information signal. In
case of ideal operation, the amplitude of the wave transmitted should be switched between
zero and a certain maximum level. However, for practical available amplitude modula
tors, the lowest level of zero (no power transmission) is difficult to obtain at high bit
rates. The ratio of the amplitude of the signal part of the IF photocurrent in case a space
is sent and in case a mark is sent is an important parameter. This ratio is defined as the
Extinction Ratio (ER) of the external amplitude modulator. In [9] the sensitivity degrada
tion, due to the use of an amplitude modulator with nonzero ER, has been analyzed for a
{2x2} and {3x3} phase diversity ASK receiver. The sensitivity degradation has been
computed with respect to the ideal (2x2} and {3x3} phase diversity ASK receiver,
respectively, and special attention is given to the optimal threshold level in dependency of
the ER, and the impact of nonzero ER's on the shape of the IF modulation spectrum.
In Figure 2.7, the results for a {2x2} phase diverisity ASK receiver are presented for a
(non-)optimized threshold value, and I) varying from 0.0 to 0.25.
From this figure it can be concluded that the degradation due to nonzero values of the
ER, can be significantly reduced by optimization of the threshold level. For the {3x3}
phase diversity ASK receiver, the results are depicted in Figure 2.8.
m u
.~
>-~
III c QJ [L
16
5 ------------------.-.--------- --- /j non-optimal/I threshold
4
3 optimal threshold _______
2
o~~~~l-----L----L----L---~
0.0 0.05 0.1 0.15 0.2 0.25
The Extinction Ration
Fig. 2.7: Sensitivity penalty versus extinction ratio for a {2x2} phase diversity ASK receiver,
and a BER ~ 10".
5
non-optimal
threshold 4
m u
3 g >- optimal ....
threshold iii c 2 QJ
Q.
o~~~~-~-~--~--~ 0.0 0.05 0.1 0.15 0.2 0.25
The Extinction Ratio
Fig. 2.8: Sensitivity penalty versus extinction ratio for a {3x3} phase diversity ASK receiver,
and a BER ~ 1 0".
17
From this figure, it can be concluded that similar as for the {2x2} receiver, the degrada
tion due to a nonzero ER can be reduced by optimization of the threshold level.
The IF ASK spectrum for nonzero extinction ratios.
After detection of the composite wave and appropriate IF low-pass filtering, the signal
values in both receiver structures are given by [9]
sp) = ~ RVPLPS b(t) cosrt;(t)],
where
N = 2,3
jj(t) = ... j/2 + c/>(t)
jj(t) = 2?rj/3 + c/>(t)
(the number of receiver branches),
for the {2x2} phase diversity ASK receiver,
for the {3x3} phase diversity ASK receiver,
(2.7)
and b(t) is the signaling waveform. It is a known feature of two lasers with a Lorentzian
power spectrum, that after combining and mixing, the resulting power spectrum is also
Lorentzian. It has a FWHM linewidth which equals the sum of both laser linewidths
separately [9]. This implies that for the computation of the ASK modulated laser
spectrum, it is sufficient to perform the following convolution
(2.8)
where Sk,,,,(j) is the equivalent baseband laser power spectrum, which has a Lorentzian
line shape. Moreover, since the transmitting and local oscillator laser have equal
characteristics, the FWHM bandwidth is twice the laser linewidth of a single laser. M(j)
represents the modulation spectrum of b(t). The modulation spectrum M(j) can be
calculated to be [9]
18
(2.9)
Substitution of M(f) in Equation (2.8) finally results in the (equivalent baseband) ASK
modulated power spectrum Sm(f) for an ER of a.
(2.10)
A computed baseband ASK modulation power spectrum is given in Figure 2.9.
~
CQ -0 c v ~
~ .e
0
·1
-2
-3
-4
-5
-6
-7
-8
-9
-10 -1 -0.8 -0.6 -0.4 -0.2 0 0.2
Frequency (in Hz)
0.4 0.6 0.8
1 1
xl(}'l
Fig. 2.9: The computed equivalent baseband ASK modulated power spectrum for a PRBS of
140 Mbitls, an extinction ratio of 118, and FWHM laser line widths of 30MHz each.
19
For an increasing value of the ER, the FWHM bandwidth decreases. This can easily be
explained, since the power spectrum Sm(/) is the convolution of the modulation spectrum
and the laser spectrum, the spectrum Sm(/) is broader than both power spectra separately.
However, for values of the ER approaching towards one, M (/) becomes a delta function.
The convolution of Sm(/) and M(/) then results in the un modulated laser spectrum, with a
FWHM bandwidth equal to the sum of the FWHM laser Iinewidths.
In conclusion, the sensitivity degradation for the {2x2} and {3x3} phase diversity ASK
receiver, due to the use of an external amplitude modulator with a nonzero ER, highly
depends on the position of the threshold level. By optimization of the threshold level the
sensitivity penalty can be significantly reduced. For an optimized threshold level and a
BER = 10··, the {3x3} receiver is less sensitive to a nonzero value of the ER than the
{2x2} receiver. For the ER of 118 and a BER = 10··, the sensitivity penalty for the {2x2}
and the {3x3} phase diversity ASK receiver has been calculated to be 2.25 dB, and 2.13
dB, respectively (see Figure 2.7 and 2.8).
For ASK modulation, the FWHM bandwidth of the IF power spectrum is broadened in
comparison with the FWHM bandwidth of the un modulated laser spectrum. Increasing the
value of the ER leads to a smaller modulation index and therefore to a smaller value of
the FWHM bandwidth.
20
Chapter 3
The transmitter and local oscillator lasers
In the transmitter as well as the receiver the same kind of laser, constructions for the
laser fiber coupling, optical isolation and control circuits for temperature and DC current
where applied. Therefore these subjects are described in a separate chapter.
3.1 The DFB lasers
For coherent detection single mode lasers have to be used. The same type of laser diode
is used for both the transmitter laser and the local oscillator laser at the receiving end.
The laser diodes are GaInAsP DFB (Distributed Feed Back) lasers from Philips. A part
number or other designation is unknown.
The lasers had to be provided with a temperature sensor, a thermo-electric heatpump and
a laser fiber coupling mechanism. The laser itself is approximately 0.3 mm in square and
is soldered on a small heatsink. This small heatsink is mounted on a second gold
metalized copper heatsink of about 2 x 6 mm. Since the optical frequency of the lasers is
susceptible to variations in temperature (10 - 20 GHz/K), it is stabilized by a temperature
control circuit. The temperature sensor (Siemens K19 thermistor) is mounted as close to
the laser as possible. The sensor is provided with thermo conducting paste and glued
between the heatsink, where the laser is soldered on, and the ridge of the copper heatsink.
The copper heatsink with the laser diode is mounted on a Peltier thermo-electric heat
pump.
At 25 degrees C the wavelength of the transmitter laser is 1539.2 nm and 1538.2 nm for
the local oscillator laser. The wavelengths or optical frequencies of the lasers are matched
by varying the temperature of the lasers. In Appendix 2 the data and characteristics of the
laser diodes are given. The maximum output power of the lasers is 5 mW. The linewidths
of the lasers are measured with a self homodyning interferometer. With self homodyning
21
the optical output signal and a delayed optical output signal of a laser are combined. This
signal is detected by a photodiode. The spectrum of the photocurrent is measured by
means of a spectrum analyzer. With this method the bandwidth of the measured spectrum
is twice the bandwidth of the spectrum of the laser because the laser output signal is
combined with a delayed signal which has the same frequency and spectrum. In Fig. 3.1
the spectrum of the transmitter laser is shown at 4 m W output power. The horizontal
frequency scale must be divided by two so the 3 dB bandwidth of the spectrum is equal to
the FWHM (Full Width Half Maximum) linewidth of the laser which is approximately
20 MHz.
~ L -21. 00 dBm - -- --- --~~ - - -TTE 10 [B .00 dB/DI V - -- -~- ---
-~ ----~ -----
I~ ~
)-~ \ ~ .. ~ _____ 1 _____
-----I
f----- - i I 1"'~~\l1
l ----
,~
STRRT 0 Hz 'RB 1.00 MHz -VB 3.00 kHz
-----~ -~~~ MIC1~ ---- -----~- -----~-
-
i -----+- -~- --~- ------- ~
I
~
-I--~-- -~I---I __ J_ . ---
1
I
i STOP 50.00 MHz
ST 50.00 msec
Fig. 3.1: Spectrum of the transmitter laser at 4 mW output power.
The Relative Intensity Noise (RIN) is measured as well, using the HP 71400 Lightwave
Signal Analyzer. RIN is defined as the variance of the optical power fluctuations, divided
by the mean optical power squared. This noise of the lasers highly depends on any
reflected light returning into the laser cavity, among other things. In Fig. 3.2 the relative
22
intensity noise and the noise floor of the analyzer are shown. The optical fiber connectors
were provided with index matching liquid, to minimize the displayed intensity noise. Both
spectra are rather flat and the traces hardly differ so the RIN is smaller than -128 dBlHz.
RL -9 81 dBm RTTE 0 B 5.00 dB/D V RVG WR - .8 dEm MARK R R. . N. 1. 00 GHz -62. 1 dB 1
START 0 Hz RS 3.00 MHz
(1 ~ z )
'VB 10.0 kHz
MKR #lA~KQ 1 000 GHz
..L
-6 .61 8e (1 Hz) L GHTW VE Of TICRL
SRMF LE
~
STOP 2.900 GHz ST 290.0 msee
Fig. 3.2: Relative Intensity Noise and the noise floor of the analyzer.
3.2 Laser fiber coupling
Since the DFB lasers were not packaged and pigtailed, the coupling of the laser light into
a fiber had to be realized in our laboratory.
Because of the rather large dimensions of the optical isolators and to avoid backreflection
into the laser cavity, the isolators could not be placed in the optical path directly.
To avoid a change in the linear state of polarization of the laser light, the .coupling
between the laser and the isolator was made with a single-mode polarization maintaining
23
fiber. The fiber had a length of approximately 30 cm.
For the laser fiber coupling, the hemispherical taper-end method was used [32]. The
fabrication process consists of the following steps :
1 - Elongation of a fiber by heating and pulling it in an electric arc discharge. The
biconical shape formed this way, has a region of minimal diameter, the waist. The
process is stopped when the desired diameter of the waist is reached.
2 - Cutting the fiber near the waist.
3 - Heating the top of the taper on which a hemispherical surface is formed by the surface
tension of the glass.
Low coupling efficiency would be obtained if one would apply above method directly to a
Panda fiber because of the presence of the SAP's (Stress Applying Parts ).
To solve this problem a Panda fiber was fused to a standard single-mode fiber, see
Fig. 3.3.
panda standa,d
1 2
L __ -L-I ==== 3 4
Fig. 3.3. Fabrication of hemispherical tip.
The waist was drawn in the standard fiber close to the fused splice at a distance of 2 to 3
cm. In the short section of the standard fiber the state of polarization of the light coupled
from the laser into the standard fiber will not change significantly. The manufacturing
was done with the setup that we normally use for making fused biconical taper couplers.
For easier handling and aligning, the uncoated part of the fused fibers was fixed in a
silica capillary, with the tapered-end leaving at one side.
The curvature of the radius of the hemisphere tips is about 20 to 30 /lm. The maximum
coupling is obtained at distances from fibertip to laser varying 20 to 40 /Lm. The coupling
24
efficiency slightly depends on the divergence of the laserlight.
We measured efficiencies of about 21 to 28 %. Improvement can be obtained by making
the radius smaller. For smaller radii of the fibertip, it should be positioned closer to the
laser to get efficient coupling. However, it will increase the risk of damaging the laser
facet during experiments.
The capillary with the protruding hemispherical fibertip is poSitioned before the DFB
laser by a precision translation stage for the z-axis and two adjustment constructions, both
for x and y direction. Each adjustment construction possesses a way to move in the x and
y direction, see Fig. 3.4. A translation in the y-axes is done by tilting the upper arm with
an adjustment screw, so that it pivots about a point (q). The arm acts as a lever, so that
about one fourth of the screw translation is effected on the taper. Tilting the lower arm
results in a x-translation caused by pivoting about point p. In fact the two movements are
not in orthogonal planes, but are displacements on the outline of two rather large circles.
The adjustment construction, closest to the laser, serves for the coarse alignment. The one
clamping the rear side of the capillary is meant for fine adjustment, see Fig. 3.4.
Likewise the block on which the laser is mounted, the construction is also made of alumi
num to minimize the effects of thermal expansion on the coupling behaviour.
,-taper )
r-;~Hq
Fig. 3.4: x-v manipulator and Laser-fiber manipulator.
3.3 Optical isolation
If laser energy, travelling in a fiber optic system, is reflected back into the cavity of the
transmitting laser several unwanted effects can arise. For instance, amplitude instability.
To reduce and eliminate effects of optical feedback, it is necessary to optically isolate the
laser.
25
In this system it is done the following,
1 -Preventing reflection from the laser pigtail, see chapter 3.2.
2 -Preventing reflections from the downstream system by using optical isolators.
Ad .1 The end of the Panda fiber is polished to get a flat endface with an angle of 10
degrees to the normal on the optical path, see Fig. 3.5. In this way returned power from
Fresnel backreflections at the end of the pigtail will be launched in the fiber cladding and
fade out. A collimated beam is necessary to pass through the optical isolator. Therefore a
small spherical lens is positioned right behind the pigtail. The spherical lens is partly
flattened and this flattened plane is fitted at an angle of 6 degrees to avoid backreflection
into the pigtail. The lens has been provided with an antireflection V-coating for 1550 nm.
The collimated beam diameter is about 500 I'm. A theoretical reflection loss of 90 dB can
be reached in this construction, also called lensferrule. These lensferrules were fabricated
at Philips Research Laboratories in Eindhoven.
.---------~ r---------, I I I
I
I
I I _________ J L _________ ---.J
-90 dB lensferr'"ule Isolator 1 isolator 2 lensferrule
Fig. 3.5: The optical isolation.
Ad.2 The optical isolator is placed in the collimated beam behind the 90 dB lensferrule. It
consists of two cascaded Faraday rotators and polarizers in the same housing, and has a
measured isolation of 64 dB at 1550 nm. They were manufactured by OFR (USA). The
collimated beam is coupled into another lensferrule, which is connected to another short
piece of polarization maintaining fiber (Panda). This second lensferrule has no angle nor
is the lens flattened.
The measured optical spectrum of the DFB-Iaser was still unstable and so it was decided
26
to place a second isolator in series with the first. The second has an isolation of 42 dB
and has a single Faraday rotator. Much of the unwanted effects are eliminated this way.
Application of two optical isolators makes it also possible to position each one at a small
angle with respect to the optical path to avoid backreflections. The second isolator
compensates the parallel displacement of the collimated beam caused by the first one.
The polarizer of the first isolator is placed parallel to the plane of polarization of the
beam to maximize the transmission. As the double Faraday rotator in this isolator rotates
the polarization 90 degrees, the second isolator has to be adjusted to the new orientation.
The second isolator changes the plane of polarization 45 degrees and the orientation of the
receiving lensferrule is fitted to launch the polarized beam in the Panda fiber parallel to
the SAP's.
The lensferrules are clamped in V -grooves of a massive block and cannot be tilted. The
transmission loss caused by the lensferrules is 1.4 dB. The loss of the lensferrules and
both isolators is 2.6 dB, transmission 55%.
The isolation of the composite isolating system was beyond our measurement ability and
was found to be better then 75 dB.
3.4 The temperature controllers
The temperature controllers regulate the temperature of the transmitting and the local
oscillator lasers, see Fig. 3.6.
The lasers are selected for a matched optical frequency but, in order to get the frequency
difference within the locking range of the frequency controller, the lasers should be
tunable.
As the optical frequency of both lasers is dependant of the temperature of the laserchip, it
is possible to tune the frequency so that the frequency difference between the local
oscillator laser and the transmitting laser equals the receiver intermediate frequency.
This implies that the temperature must be adjustable over a range large enough to tune the
lasers within the frequency controller locking range and that temperature stability should
be adequate to keep them there.
27
TRA: ISM I TTER RECEIVER
IN
Fig. 3.6: The temperature controllers in the system.
DATA ~ OUT
CLOCK .- ... OUT
The temperature Gontroller consists of a temperature sensor, a control circuit and a Peltier
thermo-electric heatpump which can cool or heat the laserchip, see Fig. 3.7.
The sensor is a Siemens K19 thermistor that was chosen for its large sensitiyity and small
dimensions. This sensor is mounted on the laser heatsink as close to the chip as possible.
The temperature information from the sensor is used by the control circuit to drive the
thermo-electric heatpump.
SENSOR ~.
(L.O ) LASER •
T.E .HEATPUMP .. TEMPERATURE CONTROLLER ,- ------- -------- ,
I
~ 0 ~. I
I
-K I
I I I I
~, I" I
~ I
I I I I I BRIDGE AMPLIF. SETPOINT COMPAR. DRIVER I I I L ____________________________________ ~
Fig. 3.7: Schematic diagram of the temperature controller loop.
28
Resistance variation of the thermistor is converted to a voltage variation by the bridge
circuit. This voltage (approximately 47 mY IK) is amplified by an amplifier designed
around the Burr Brown INA 10 I low drift operational amplifier, bringing the signal level
to I V/K.
The output of the setpoint circuit, which is adjustable on a front panel dial, is compared
with the I V IK output of the amplifier.
The comparator can be used in a proportional mode, or in an integrating mode. The
proportional mode is used during the startup of the system in order to avoid large
temperature overshoots which could damage the lasers.
After an initial temperature stabilization the integrating mode is switched on by means of
a front panel selector switch.
The comparator controls the driver circuit. In order to be able to provide a positive
current (=cooling) or a negative current (=heating) to the Peltier device, a bridge design
of the driver circuit has been chosen. The driver also incorporates a current limiting
circuit to avoid overloading of the Peltier device.
3.5. The current control circuit
A block diagram of the current control circuit is shown in Fig. 3.8.
DISPLAY
~r.
REFERENCE ~
CURRENT ... I
SOURCE SOURCE p + ~
MODULATOR • ...
LASER DIODE
Fig. 3.8: Block diagram of the current control circuit.
29
The schematic diagram is given in Fig. 3.9.
LD
Rv
MOD. REF Rb 1-41----1+
Re
Fig. 3.9: Schematic diagram of the current control circuit.
The DC current through the laser Diode (LD) is detected by means of resistor Re and fed
to the input of the OpAmp. This voltage is compared with the voltage of a reference
source (LM385Z2.5). This source has a low temperature coefficient. The current can be
controlled with the variable resistor Rv. The resistor Re defines the maximum laser
current. The voltage across this resistor Re is the input signal for the laser current circuit
and is also coupled with a digital display. The LD can be damaged because of peak cur
rents when for instance the current control circuit is switched on or off. Therefore the
circuit will increase the DC current through the laser slowly (20 Sec) when it is switched
on and vice versa when it is switched off.
The modulator is made up with a differential amplifier and consists of two HF transistors.
The current of one branch of the amplifier is added by the LD current. A third transistor
is used as a current source for the differential amplifier. This current can be adjusted with
a variable resistor and so controls the modulation depth.
The input signal for the modulator is an ECL-signal.
30
The temperature adjustment is displayed with a second digital display, which is connected
with the temperature control circuit. Two variable resistors are used, one for the zero
point adjustment and one for the gain of the detected voltage respectively.
31
Chapter 4
The transmitter
Both lasers and control circuits are equal for transmitter and receiver. These parts are de
scribed in chapter 3. The additional parts of the transmitter, the external ASK modulator
and the FSK modulator, are described in this chapter.
4.1 The optical amplitude modulator
In case of ASK modulation direct modulation of the laser bias current it is not possible in
coherent detection systems because of laser chirping. In this case an external amplitude
modulator must be used. Because of the relatively low driving voltage the integrated
electro optical 1550 nm amplitude modulator from GEe Y-35-5370-01 is chosen which
has a bandwidth of 2.5 GHz. In Appendix 3 you find the data sheets of the modulator.
The modulator operates according to the Mach Zehnder travelling wave principle and is
made in lithium niobate. In Fig. 4.1 a schematic diagram of the modulator is shown.
RF,owce
Optical input Optical output
Fig. 4.1: Schematic diagram of the optical amplitude modulator.
32
Polarized light is launched into the optical waveguide, where it is split into two equal
parts. When a voltage is applied to the central electrode, opposing electric fields cause the
electro optic effect to advance one wave and retard the other. This is optimized by using
a travelling wave electrode structure so that the microwave and optical wave propagate
together. The light is recombined in the Y junction. In phase waves are transmitted
through the output guide and out of phase waves are configured into a higher order mode
which is lost into the substrate. In Fig. 4.2 the normalized optical transmission of the
modulator is shown as a function of the DC input voltage.
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
o~~~~~~~~~~~~~~~~~rr~~~~~~ -2.50 -2.00 -1.50 -1.00 -0.50 0.00 0.50 1.00 1.50 2.00 2.50 3.00
DC input voltage [V)
Fig. 4.2: Normalized optical transmission of the optical amplitude modulator
as a function of the DC input voltage.
The insertion loss of the modulator when the transmission is maximum (bias voltage is -
1. 7 V) is 7.1 dB and the extinction ratio or the maximum onloff ratio (bias voltage is 2.5
V) is 26 dB. To obtain maximum onloff ratio of the optical output in case of an AC
coupled modulation, a bias voltage of 0.4 V is necessary and the amplitude of the input
signal must be 4.2 Vtt.
33
In order to avoid reflections, the electrical input cable of the modulator must be matched
loaded. According to typical specifications the input impedance should be 50 Ohms.
However, by Time Domain Reflectometry measurement, this turned out to be about 27
Ohms. A series resistor of approximately 23 Ohms is installed to accommodate. A bias
tee is used for AC/DC decoupling.
In Fig. 4.3 the frequency characteristic of the electrical to optical responsivity is shown,
measured with an HP 8702A lightwave component analyzer. The 3 dB bandwidth is
approximately 2.4 GHz.
CH1 B/R
W/A
Cor E/O
TU
t
~ L-
log MAG
E-EC
/ III V vv
1 dB/ REF -55 dB
pJv '\ /\.
\ A/
\" / \ ~ ~
tv...
I
yw V
START .300 000 MHz STOP 3 000.000 000 MHz
Fig. 4.3: Frequency characteristic of the electrical to optical responsivity.
4.2 The FSK modulator
FSK modulation can be obtained by direct modulation of the transmitter laser bias
34
current. The frequency sensitivity of the transmitter laser used is 0.25 GHz/mA. For a
modulation index of 7, the variation in current has to be 4 rnA. This results in a ampli
tude modulation index of less than 10 %. First a simple resistor network with a capacitor
for AC coupling was used. Since it was difficult with this solution to control the modula
tion index, a electronic driver circuit was developed. The circuit diagram of the FSK
modulator can be found in Appendix 4.
35
Chapter 5
The receiver
In this chapter the parts of the receiver will be discussed except for the local oscillator
laser, which is dealt with in chapter 3. These parts are; the optical network, the OlE
converters, the IF section, the frequency controlling and the clock and data recovery
circuit.
5.1 The optical network
The optical network in the receiver is used to mix the received optical signal from the
transmitter with the light of the local oscillator (DFB-Iaser). In this phase diversity
receiver a {3x3}-opticaI fused fiber coupler has been used.
5.1.1 The 3x3 fiber coupler
Such a coupler can be made of optical fiber with the fused biconical taper technique.
The fabrication process is as follows:
In our coupler manufacturing setup, three matched cladding single mode fibers were
aligned after removal of the primary coating. The three fibers are stacked like a triangle
with two fibers parallel and one on top in the groove, formed by the former two. The
fibers are fused together to forme a circular area by heating them with an electric arc
flame placed below them. Then the fibers are pulled and elongated to obtain optical
coupling. During the fabrication, laserlight of 1540 nm is coupled into the upper fiber.
The optical power at the three outputs is monitored and the tapering process is halted
when the desired coupling ratio is achieved, i.e. the optical power is equally divided over
the three output fibers. Good results will only be achieved if the fibers in the fused
section have equal distances to each other.
Of even more importance are the phases between the three outputs which should be 120
36
degrees for the designed phase diversity system. For the output power of the three
branches an extra gain correction can be made in the amplifiers following the PINFET
modules. Such a correction is not possible for the phase relations. Therefore it should be
more effective if during the fabrication the phase at the output ports is monitored instead
of the optical power. In a lossless coupler, the phases between the outputs will automati
cally be equal if the optical power is equally distributed. After packaging the coupling
ratio at the fiber output ports is checked for each branch acting as an optical input and
vice versa. In our {3x3}-coupler (No. 176) one of the outputs found to suffer from severe
optical attenuation when launching optical power in the same fiber at the input side,
resulting in an unequal light distribution at the outputs. This is shown in Fig. 5.1.
~
~ "-..
a; 3 0 "-~
:J "-~ :J 0
OPTICAL POWER DISTRIBUTION 3 x 3 COUPLER No. 176
50
40
30
20
10
0 ,A 2A
Input fibers
3A
_ OUTPUT
18
IIIiI!IIIIIIIIl OUTPUT 28
~ OUTPUT 38
Fig. 5.1: Power distribution of 3x3 coupler No. 176.
However, for the other two input fibers, the power division was equally. Since a phase
diversity receiver only uses two input fibers the latter should not be a problem. Unfortu
nately, the measured phase differences between the three output ports were found not to
be exactly 120 degrees. The optical power launched at the input of a coupler from Sifam
(UK) was fairly equally divided over the three output fibers. This applied for all of the
three inputs. Measurements showed 33 % + 5, as is shown in Fig 5.2. The excess loss
37
was 0.1 dB.
~
*' ij; 3 0 "-~
~
"-~ 8
OPTICAL POWER DISTRIBUTION 3 x 3 COUPLER SIFAM
50,-----------------------------,
40 -30 mIIIIlIIiIl
20 ~
10
0 1A 2A 3A
Input fibers
OUTPUT 1B
OUTPUT 2B
OUTPUT 3B
Fig. 5.2: Power distribution of the 3x3 coupler of Sifam.
The phase differences also showed good results namely 120 + 9 degrees, see Fig. 5.3.
PHASE MEASUREMENT COUPLER SIFAM
360,------------------------------,
1 270
m 180
~ :ll &. 90
o 1A-2A 2A-JA 1A-JA 18-28 28-38 18-38
Irput fibers
_ phase
1B->2B
IIIIIIIIII!III! phase 2B->3B
~ phase 3B-> 18
I2i88:3 phase 1A->2A
f22Zl phase 2A->3A
~ phase 3A->1A
Fig. 5.3: Phase differences of the Sifam 3x3 coupler.
38
For phase measurements the setup from Fig. 5.4. was used.
-3 dB coupler Q:>tlcal Network: Photo-Olodes
Fig. 5.4: Phase measurement setup.
The lightsource is a HeNe-laser for 1520 nm followed by an optical isolator of 32 dB.
The AOM shifts the optical frequency by 80 MHz. The output currents of the photodetec
tors contains the difference frequency of the two input wavelengths. The phase differences
between the output ports were measured by a Vector Volt Meter (VVM), each time using
two input ports.
5.1.2 Phase diversity network measurement set-up
The phase diversity network is an important component in a fiber optic phase diversity
coherent communication system.
Combining both the received optical signal and the output of a local oscillator laser results
in multiple output signals with a certain fixed phase relation, for example 7r12 between the
outputs in the case of a four branch network, while the power distribution should be
equal.
Fig. 5.5: The Phase Diversity Network in (a part of) a coherent receiver.
Imperfections in the network could lead to both phase and splitting-ratio errors. Ratio
39
errors however can be corrected for in the electronic processing part of the system, but
phase errors wi1llead to receiver performance degradation.
For this reason the measurements on the phase diversity networks focussed on the phase
relations of the output signals.
Phase diversity networks can be manufactured by several methods, for example by means
of a fused biconical taper technique. These networks feature a low excess-loss and are
easy to use in a fiber optic system, as the optical signals stay inside a guided(fiber)
system.
Measuring the phase relation of the outputs is fairly straight forward. A simplified layout
is shown in Fig 5.6.
LASER
sa~
Sb
Fig. 5.6: A phase-relation measurement method.
S1
S2
S3
S4
The output of a laser is fed both direct to an input branch of the phase diversity net
work(Sa), as well as via an acoustic optic frequency shifter, that shifts the optical
frequency by 80 MHz with respect to the other input branch(Sb). The outputs branches
(SI..S4) are connected to photodetectors, where the optical signals produce a difference
frequency. The resulting photodetector currents are amplified and measured by means of
an oscilloscope or a vector voltmeter.
The Phase Diversity Networks where realized by T.U.D. by means of a planar technolo
gy. Although the measurement is straight forward, the practical implementation with
regard to the planar network is somewhat more complicated:
40
1) This planar technology doesn't allow to couple fibers direct to the network
branches, therefore we used a prism coupling method at the input side, and
a lens coupling at the output side of the network.
2) Although earlier experiments have been done with (visible) 632 nm ReNe
laserlight, the current planar circuit are designed for (invisible) 1523 nm
Infrared light, that makes optical alignment a lot more difficult.
3) Due to high loss in the coupling area, we encountered problems with
regard to the signal-to-noise ratio of the photodetector signals.
Besides we found that optical reflections degraded the stability of the
measurements.
The most suitable method to couple light into the planar circuit is the prism coupling
technique. Coupling light out of the planar circuit to the photodetectors can be done by
means of a lens coupling, see Fig. 5.7.
Microscope objective
Planar circuit
P, ism COUp I I ng Lens coup ling
Fig. 5.7: Planar circuit coupling methods.
If we want that adjusting the output coupling does not influence the input coupling we
have to fix the planar circuit itself and move the input beam (both angle and location on
the planar circuit) and the output lens coupler. This implies that the total optical process
ing should be mobile, or that a flexible interface, between the processing setup and the
prism coupling optics, should be used, see Fig. 5.8.
41
Fig. 5.8: Input coupling set-ups.
As the mechanical alignment of the optical processing part is rather critical, the last
solution has been chosen to avoid alignment stability problems.
Optical processing.
The light source, a Melles Griot type 05-LlP-171 infrared HeNe laser, delivers a 1523
nm. beam via an optical isolator to a lens-ferrule equipped single mode fiber.
The lens-ferrule can be aligned to the laser beam by means of a custom made alignment
facility, in order to obtain a maximal efficiency.
The fiber is connected to an input branch of a 3dB coupler that splits the power in two
equal parts. The other input branch is connected to a 632 nm SpectraPhysics HeNe laser
to couple some visible light in the optical path for alignment purposes. One output of the
coupler is connected to an acoustic optical frequency modulator, that shifts the optical
frequency by 80 MHz.
Both branches pass an optical attenuator and a polarization controller.
The attenuators reduce measurement instabilities due to optical reflections, as reflections
have to pass the attenuator more than once before entering the photodetectors.
The fibers leaving the polarization controllers are led to the beam delivery system which,
is mounted on the vertical rotation stage. This beam delivery system consists of two
microscope objectives that collimate the beams, a 45° mirror, a beam-combiner and a
cylindrical lens (f=50 mm).
Initialy a non-polarizing beam splitter was used as beam-combiner but as this splitter
introduced a 3dB excess power loss, we replaced it by a beam-combiner that consists of
selective gold coated prisms that are cemented together.
42
By proper aligning, both beams will focus through the cylindrical lens and the prism on
the coupling area of the planar circuit, see Fig. 5.9.
.. . j;..
Fig. 5.9: Input and output coupling area's.
output area
The prism that couples the light in the planar circuit is positioned by hand and not fixed.
In order to obtain coupling an index matching liquid has to be added between the prism
and the planar circuit surface.
The light that leaves the planar circuit at a cleaved edge is collected by a microscope
objective and, enlarged, projected.
A movable mirror that is mounted behind the objective makes it possible to project either
on the photodetectors, or on a infrared sensitive CCTV camera (Hamamatsu) for
alignment purposes, see Fig. 5.10.
As the distance between the projected output spots is small, in spite of magnification, the
spots are spatial separated by means of an image-splitter. (a 45° mirror, one spot on the
mirror, the other three just passing it).
So one spot is always focussed on photodetector no. 1 while the other three can be
scanned by moving photodetector no.2. This arrangement allows phase measurements
43
between output 1 and output 2,3 and 4, see Fig. 5.11.
I mage-sp I j t ter
Fig. 5.10: I.R. camera and photodetectors.
o L Photodetector 1 ~ 1111111111'
(3 , ~< : : -11 111111111 [I]
Photodetector 2
Planar circuit Image-splitter
Fig. 5.11: Image splitting.
Electronic layout.
The lasers have their own powersupplies and did not need further modifications. The
optical frequency modulator is connected to an Isomet driver. The frequency modulator
and the driver had to be adjusted for maximal optical output power.
44
The photodetectors consist of a Philips CPF30 PIN photodiode and a Signetics NE 5211
Transimpedance amplifier.
These two devices are build together inside a casing that provide protection against
interference from outside.
The transimpedance amplifiers are followed by Trontech type WIGE 1O-PS-22 amplifiers
that provide ample signal to the following Hewlett-Packard HP 8508A vector voltmeter.
Due to the high optical losses in the setup and in the planar circuit the signal-to-noise
ratio turned out to be to small to lock the vector voltmeter reliable. Inserting bandpass
filters (BW=20 kHz) between the Trontech amplifiers and the HP vector voltmeter
improved the "locking-range" of the vector voltmeter by more than 30 dB, see Fig. 5.12.
Attenuators Pol. Control
Coup I er
000
I~ Iver !
Photodetcctcr 2
In'Ia~e-SPlltter
.",'d,.c·1 V II V I >, ".cs I Jl I [J[J
Vector Voltmeter
Osc; I loscape
Fig. 5.12: The measurement set-up.
S.2 The PINFET OlE convertor module
This module is situated between the optical network (three optical outputs), and the
I.F.section (three electrical inputs). The module also provide a signal to the frequency
controller. The task of the module is to convert the optical signals to electric signals
45
which can be processed by the following modules, see Fig. 5.13.
TRANSMITTER RECEIVER
~=-l ~
~ E:J
DATA IN
Fig. 5.13: The PINFET DIE converter module in the system.
DATA '~OUT
CLOCK ~OUT
Three optical FC/PC connectors, which are mounted on the front panel, facilitate the
transfer of the light from the patch cords to the pigtails of the B. T &D. PINFETS type
PDC 2200 where the Opto/Electric conversion takes place, see Fig. 5.14.
PINFET OlE CONVERTER ----------------------------1
I
[D> I [I i SPL I I I Ell 1 MSI\ 0 18~,
I
I
-~i~1 II ~ IE]' ~ ___ 2=i _____________________ J
Fig. 5.14: Schematic diagram of the PINFET module.
..... ~
.. ~
.... ~
,.~
More information about the PDC 2200 module can be found in the appendices.
The PDC 2200 module is followed by amplifiers based on an Avantek MMIC type MSA
0785.
46
These amplifiers increase the system gain by 12 dB, and are connected to resistive power
splitters. Their outputs are located on the front panel.
So each of the three branches have two outputs, from which one is used to drive the
I F section. The remaining outputs are used for the frequency controller and for monitor
purposes. Fig. 5.15 shows the overall frequency response of the channels.
Branch I
..
.- '" I_I.,,, . , .... ,. '''1
,
i I
',.",'
, ..
"UI , """ """ u'"'- "" ..
Branch 2
Fig. 5.15: Frequency responses.
o;H, ~'" loO ~.r,
;J"I'! I 1
• I I I
Branch 3
In order to compensate for unbalance between the optical branches, for example due to
the optical network, it is possible to adjust the gain of each channel independently by
means of an internal gain control.
5.3 The J F section
The IF section receives the three signals from the PINFET OlE converter module, see
Fig. 5.16.
After filtering, amplification, detection, and addition of the three branches there IS a
baseband signal available to the clock and data recovery module, see Fig. 5.17.
The first stages of the I F section are the lowpass filters. These filters, having a cutoff
frequency of about 750 MHz, reduce the noise contribution of the preceding stages, but
don't degrade the signals. All three filters are adjusted for an equal response.
Fig. 5.18 shows the filter frequency responses.
After filtering, the signals are amplified by Avantek IVA 05118 variable gain amplifiers,
47
I ~M·I::,Iv111 T l~ RECEIVER
6---.-reMP
1 l~~lr_' _
DATA IN
~~~~v I
Fig. 5.16: The I F section in the system.
I.F SECTION
DATA .OUT
CLOCK · ... OUT
r---------------------------------------~
IL P FILTER AMPL I F I ER SQUARER ADDER P. D. FILTER :
I I I I I I I I I I I : .. ~
i C ';:Jt .... ® ~1~c~iT P DAMPLIFIERi I I 1 ______ ----------------------------------"
Fig. 5.17: Schematic diagram of the I F section.
providing a gain between 0 and 26 dB to the signals. The gain of the amplifiers is
controlled by an automatic gain control circuit which will be described later on. The
outputs of the amplifiers, both inverted and non-inverted, are connected to the R.F. and
L.O. inputs of Avantek lAM 81018 active double balanced mixers, which are used here
as square-law devices. Fig 5.19 shows the input signals of the I F section (simulated 140
Mb/s ASK 0-1 bit sequences) and the output signal of one squarer. (lower trace)ln order to
restore the original ASK signal, the output of an three squarers have to be added. This is
simply done by means of a resistor network. The network is designed for a maximum
isolation between inputs because, the mixcr outputs turned out to be load sensitive. A post
48
i--i--r-t-- .. - . +-+---f-f-----i
:;,-.\--\ f- --+\ .. --- -'-r- lI' : ; I' • t-- --+--- '
__ LL.l, __ LL 1'1.1 STAAT .300 000 I<H1 SlOP t 000.000 DO" _,
Fig. 5.18: I F filter response
A"5,000 no -l~ - -,--------- --- 5"4.000 nS , ~44 "'
_ "on non, '''' AA I £V' ,,"""", '"V""," """" \MAl ,AMnMI nAnnAn ""
,'" ,,,'Annl rUllVv,VVI ~., onl\l'J\J\ l~MM ~AMM fiA~~'v .'." ' I U~JLlllIV'
~IM V\fiIWV IIiMnoi """: - "" " "ru
J V vvv, rVVUUU' uUVUUV' '" '
---
- ,,00 u 0.00 o , o "'
1-- - n~~~\ ~\tt1 - --- - ~\~~I NM~)I~ . __ .
'", 1~1 "' nv ' ~~ l"IU-~ J!iIW ~ ~vw 1\t\i1A~ . . .AlliVVI - ",\1 IVV I J ,I I rvv----,
.- - - _.
- -- .... . - --~- --_._-_. - - .-~
Fig. 5.19: Simulated I F input signals.
detection filter follows the addition, providing an optimal signal-to-noise ratio to the
signal.
Restricted space on the J F section printed circuit board allowed only a rather simple
filter configuration, for which the cut-off frequency has been chosen at about 150 MHz.
If necessary more elaborate filtering can be done between the J F section and the clock
and data recovery module.
Fig. 5.20 shows the post detection filter frequency response.
49
PDF REF -10 0 dBm 10dB!
ATT 10 dB A norm.>a B blank
AVG A 50
~ MKR 155.0
RBW I MHz
VBW I MHz
SWP 50 ms
"0 '\
Hz \
1\ \
\
CENTER 250.0 MHz
- -
MK I 5.0 MHz - .97 d
\. 1\ \ ./
SPAN 500 MHz
Fig. 5.20: Post detection filter respons.
f'
The output of the post detection filter is connected to the post detection amplifier.
This amplifier is necessary to bring the signal to a level which is compatible with the
clock and data recovery module. (0.4-\.5 Vtt).
The design is simple, an Avantek MSA 0785 MMIC is followed by an Avantek 0385
MMIC. The output data is available at a front panel SMA connector.
The D. C. components from the mixer outputs are added and used by the automatic gain
control circuit. This a.g.c. circuit has two front panel controls, a switch to select either
a.g.c. mode or manual mode, and a dial to adjust the data output level.
In the a.g.c. mode the added d.c. components from the mixer outputs are compared with
the setpoint from the level-dial. The result is a constant output level that is, within the
gain range of the variable gain amplifiers, independent of fluctuations of the receiver
input signal and fluctuations of the local oscillator laser powerlevel.
The gain of the amplifiers is proportional to the level-dial setting when the manual mode
is selected.
To check the performance of the I.F. section a few measurements have been carried out
by means of a simulated 140 Mb!s ASK modulated P.R. B.S. data signal.
The plot in Fig. 5.21 shows the eye pattern of the signal at the input channels of the I F
section. These signals simulate the situation where the optical frequency of the transmit-
50
ting laser differs 548 kHz from the optical frequency of the L. O. laser. The pseudo
random generator is synchronized with the difference frequency (548 kHz) in order to
get a stable eye pattern.
;' ',(Jooo wI () 00000 ;' >,0000 [J'.
I., ..... '· ........ ··'!'"
Fig. 5.21: I F input 'EYE' pattern.
The next plot (Fig. 5.22) zooms in on one of these inputs, and gives a more detailed look
at the eye pattern, the lower trace shows a magnified part of the upper trace.
Fig. 5.22: Detailed look at the 'EYE' pattern.
The output of the I F section, before post detection filtering, is shown in Fig. 5.23. This
plot demonstrate that the reconstruction of the baseband signal has been done correctly.
The eye is 'open' and there is no noticeable channel unbalance. Fig. 5.24 shows the eye
51
10.000 I1S 0.00000 s 10.000 n5
Fig. 5.23: I F output before post detection filtering.
pattern after post detection filtering. The signal-to-noise ratio is increased, the eye is
'open', some intersymbol interference can be noticed. The last plot (Fig. 5.25) shows the
-10.000 n13 0.00000 s 10.000 flS
I ..• . . ... :. ;.:.:;<;~;.';';::" .---+-: .-: .. ::.-.. ::-+ ;:\;«-<-. -+-----+----1
Fig. 5.24: The 'EYE' pattern after post detection filtering.
spectrum of the output signal after post detection filtering.
PRBS 2'7-1 NA PDF REF -19.9 dBm !OdB/ I
i
\ I iii 1 kHz I
SWP L' i 10 5 .. 1.
, ! :
J __ J CENTER 250.0 MH7
52
SPAN 100 MH7
Fig. 5.25: The spectrum of the output signal.
5.4 The frequency controller
The frequency of the optical signal of the lasers which are used, is very susceptible to
variations in temperature (10 - 20 GHz/K) and to variations in current (0.5 GHz/mA).
Therefore it is not sufficient to stabilize the temperature of the lasers and the current
through the lasers to obtain a stable difference frequency or Intermediate Frequency (IF).
In the receiver a Frequency Locked Loop (FLL) has to be used in order to synchronize
the frequency of the local oscillator laser with the frequency of the received optical
signal.
The FLL was originally meant to operate at an IF of zero which gives certain specific
problems. Later on the IF is changed to 300 MHz. The design is however very flexible
because it is software controlled, so only the program had to be altered. First the FLL for
zero IF will be described [13].
53
5.4.1 Principles of the frequency locked loop
In this case a frequency discriminator for controlling the local oscillator laser is not
suitable. With zero IF the following problem occurs. If the frequency of the local
oscillator laser is too high, the IF is too high. When the frequency of the local oscillator
laser is too low, the IF is also too high. A frequency discriminator cannot distinguish
between these two situations, so in this way it is not possible to obtain an unambiguous
control signal for the local oscillator laser.
This problem is solved in the following way.
IF signal
... -~o:;:;-ptical"l---t network trans~tter laser
-L-__ --'-photodiode
frequency controller
~I DAC. micro- k---I
controller H '---_---' L __ -'
counter ,--------, I
presca.ler
,---------,II
r------------
local oscillator laser
Fig. 5.26: Block diagram of the frequency locked loop.
To detect the IF, a prescaler and a digital counter is used, see Fig. 5.26. A prescaler is a
high speed frequency divider with an analog input and a digital output. The output signal
of the prescaler is applied to the clock input of a digital counter. The counter is controlled
by a microcontroller (single chip processor). By counting the clock transients in a certain
time interval a value for the IF is measured. The microcontroller can read the value of
the counter and the microcontroller controls the frequency of the local oscillator laser via
a Digital to Analog Convertor (DAC).
The program of the microcontroller consists of two procedures. A capture procedure and
a tracking procedure. By means of the capture procedure a control voltage for the local
oscillator laser is obtained for approximately zero IF. After this procedure the tracking
loop will minimize the IF by means of a trial and error algorithm. In Fig. 5.27. a flow
diagram of the program is shown.
54
capture procedure
Ian niue counter
t.dd .tep to
.. lue lor DAC
rnd counler
K----"YE:::.'-<: haa vallie of counter becOme .maller?
NO
iuverillep
Fig. 5.27: Flow diagram of the program for zero IF.
After the capture procedure the value of the counter is saved. Then the frequency of the
local oscillator laser is changed by a small step by changing the digital input signal of the
DAC. Again the value of the counter is read. If the value of the counter has become
smaller and thus the IF has become smaller, then the frequency of the local oscillator
laser is changed by the same step. If not, the step is inverted and thus the direction is
changed. This procedure is repeated in an endless loop.
5.4.2 The realized frequency locked loop
The described principle is realized with a divide-by-64 prescaler, an 8-bit counter, an
8-bit microcontroller and an 8-bit DAC, so only 4 integrated circuits in total. With this
FLL, the two DFB lasers are synchronized, see Fig. 5.13. Because of the speed of the
controller, the frequency of the local oscillator laser is controlled, via the current through
55
the laser, which in its turn is controlled by the DAC. The tracking range of the FLL is
0.5 GHz. This results in a variation of the optical power of the local oscillator laser of
0.1 dB, which is not dramatic. The frequency steps are about 2 MHz. These steps are
small compared to the difference spectrum of the lasers which is about 40 MHz
(FWHM). The tracking procedure of the program of the microcontroller consists of 20
lines of assembly language. The average time of a loop of the tracking procedure is about
35 (.Is, so maximum about 28,000 iterations per second can be made.
5.4.3 Conclusions
It has been shown that it is possible to make a Frequency Locked Loop with a
microcontroller for a optical receiver with zero Intermediate Frequency. The use of a
microcontroller makes the FLL fast, simple and flexible. It is possible that after long
periods of time the difference of the frequencies of the lasers becomes too large, so that
the FLL gets out of range. It is easily possible to modify the FLL so that the
microcontroller also changes the temperature of the local oscillator laser for long term
stability. Because a trial and error algorithm is used which searches for a minimum, the
FLL is independent of the modulation of the received optical signal or linewidths of the
laser diodes and there are no adjustments necessary in the hardware of the FLL.
By changing software it is also possible to choose an IF unequal to zero. In Appendix 5
the listings of the 8048 source code for zero IF and an IF of 300 MHz are given.
5.5 The clock and data recovery circuit
The baseband signal at the input of the clock and data recovery circuit is received from
the IF section, while at the output regenerated data and clock signals are available, see
Fig. 5.28.
The concept of the module is to use a phase-frequency comparator to compare a reference
frequency to a V.C.O. frequency and generate a signal proportional to the phase
difference between the to sources. After lowpass filtering, amplification, and integration
this output will provide an error signal to correct the V.C.O ..
The baseband signal from the IF section provide the reference, and the V.C.O. will be
TRANSMITTER
DATA IN
56
EL L;-r
TEMP CONTR,
RECEIVER
Fig. 5.28: The clock and data recovery circuit in the system.
DATA ..• OUl
CLOCK ... OUT
locked to it. The V.C.O. output then becomes the clock for the following data processing
equipment.
A difficulty arise due to the fact that the data receiyed has no continues zero to one
transitions, but series of zeros and ones are possible.
A conventional phase comparator will interpret the absence of data transitions as a
V.C.O. leading the reference in phase and will attempt to decrease the V.C.O. frequency
in order to obtain phase-lock. This will result in severe phase-jitter in the case of a long
run length.
In order to avoid this we used a clocked scheme (Fig. 5.29).
The incoming data is applied to a pair of differential comparators followed by a pair of
flip-flops followed by the phase-frequency comparator.
The input signal to one of the comparators is delayed a quarter bit with respect to the
other comparator input. The undelayed signal is applied to the 'D' input of a flip-flop.
This flip-flop is clocked by the V.C.O. and regenerates at its output the data input
delayed in time by the V. C. O ..
The delayed comparator output is applied to the 'set' input of the other flip-flop, which
generates the reference input of the phase-frequency comparator. The output of both flip
flops are applied to the Motorola MC 12040 phase-frequency comparator. When the
inputs are in phase it will result in equal amplitude and duration pulses on the comparator
outputs, which, when subtracted and integrated by the loopfiiter, will yield a zero error
signal for the V.C.O ..
57
B Oi fferen
.~ tial
~ 5 F lip
compar Flop q
~-~ : c
OATA IN Phase-fr-eq, .
Differ-en : V_:~~~ar ,
.~ tial .. ~ d
Flip Flop q ..
compar. e
~
d
0 nEeov, ~\~~ c
v. c.o: ," Loop-
DATA ~ .. q ampl if. ~. OUT In +f i Iter
CLOCK OUT ~ ..
Fig. 5.29: Schematic diagram of the clock and data recovery circuit.
In this situation the clock will sample the data midpoint in its stable state. If the data
remains at either the low or the high state both outputs of the phase-frequency comparator
remain at the non error, low, state, providing no correction signal to the V.C.O .. In this
situation the V.C.O. frequency is maintained.
Data regeneration is done by clocking the received data in a D-type flip-flop. The output
of this flip-flop is available at a frontpanel connector. The positive clock transition is
timed at the middle of the regenerated data bits, to provide a maximum timing margin.
The V.C.O. circuit consists of a Motorola MC 1648 ECI voltage controlled oscillator,
and an external parallel tank circuit consisting of an inductor, a capacitor and a varactor
diode. The inductor and the capacitor were selected for a V.C.O. tuning range of
approximately 135 to 145 MHz. The output of the V.C.O. is available at a connector at
the front of the module.
The loopfilter was designed according to the design guides given in the Motorola data
sheets. After lowpass filtering the Up and Down signals are summed by a LF 356 wide
band amplifier and integrated by an other LF 356 amplifier that controls the V.C.O.
tuning voltage.
During system tests it turned out that the MC 12040 phase-frequency comparator did not
58
function well on the given bitrate. The frequency comparator gave ambiguous frequency
error information to the V.C.O. when the V.C.O. was near phaselock.
The MC 12040 had to be temporary replaced by a phase comparator consisting of an
exclusive-or gate. Due to this modification the V.C.O. has to be manual adjusted to
obtain phaselock; for this a frontpanel frequency control is provided.
The phase comparator can be exchanged as soon as we obtain a phase-frequency compara
tor which is functional at 140 Mb/s.
59
Chapter 6
Experimental results
In this chapter the measured IF spectra for ASK and FSK modulation, the Bit Error Rate
measurements and the TV codec are described successively.
6.1 IF spectra
The linewidths of the DFB lasers have been separately measured by means of a self
homodyning set-up [33]. These results are discussed in section 3.1. At 4 mW output
power the linewidth was approximately 20 MHz FWHM, for both lasers the same.
Difference spectra of both lasers have been measured also at 4 mW output power of each
laser with zero IF and an IF unequal to zero.
The results are in Fig. 6.1.
The output signals of the lasers were mixed with a {2x2} fiber coupler and detected by a
PINFET DIE converter of BT&D. The spectrum of the output signal of the PINFET was
measured by a HP 71400 spectrum analyzer. In Fig. 6.la the optical frequency of both
lasers differs approximately 70 MHz so the IF spectrum lies around this frequency. The
width of the spectrum at 3 dB is 50 MHz. Theoretically this should be 40 MHz but
probably because of optical reflections the spectrum has become widen. In Fig. 6.lb the
beatspectrum is shown with equal optical frequencies. The 3 dB bandwidth is at 30 MHz
while it should be 20 MHz. This can be explained by the fact that the impact of reflec
tions is greater with homodyne detection than with heterodyne detection. When we have
fore instance an isolated laser with a long fiber, refections can cause a spectrum around
zero because of (self) homodyne detection of the reflections with the original optical
signal.
a)
b)
60
-RL 33 ee dB m RTTE lB B
.00 dB/O V MIe OWAVE
I ~\j"
II',
I~ ~J ~ ~
~ START 0 Hz RB 1.BB MHz -VB 3,00 kHz
STOP lBB. B MHz ST lBB.0 osec
RS 1.00 tlHz 3.00 kHz ST 100.0 msee
Fig. 6.1: Difference spectrum of the lasers a) with equal optical frequencies
bl with difference frequency of approximately 70 MHz.
61
In Fig. 6.2 the spectrum of the IF signal is shown for the zero IF case, with and without
FLL. The spectrum analyzer was in the maximum hold mode for 1 hour. In this mode
only maximum values are saved and displayed after the time period of the measurement.
There are hardly differences between the measured IF spectrum with FLL of Fig. 6.2a
and the spectrum of one of the lasers with a self homodyning measurement.
-RL 27 BB dB 0
RTTE lB B HB dB/O V
a) ~ ~ I A Jl II v' y~
r'tu~ , I~
.~
STRRT B Hz RB 3,B8 MHz -VB 18.8 kHz
-RL 27 00 dB m ATTE 10 B 3,00 dB/O V
b)
~ I~,
~
~, ..... ~ i' ~
"
START 0 Hz RB 3.00 MHz -VB 10.0 kHz
f\. '''f,
~
MIC O~RVE
~ ...... ~,
STOP 500.0 MHz ST 50.80 IDsec
MIC O~AVE
low". " ~
STOP 500.0 MHz ST 50,00 osec
Fig. 6.2: IF spectrum for the zero IF case a) without FLL and b) with FLL.
62
In Fig. 6.3 IF spectra are shown in case of 140 Mb/s PRBS (Pseudo Random Binary
Sequence) a) ASK modulation and b) CPFSK modulation with modulation index of 3. The
3 dB bandwidth in case of ASK modulation is 40 MHz.
a)
b)
RL 30 00 dB_ -, -
~TTE 10 B .00 dB/O V Mie OWAVE
~ ~,
r I' ~
~ -
--START B Hz RB 1.00 MHz -VB 3.00 kHz
-RL 26 00 dB _
ATTE 10 B 2.00 dB/Q V
\ \ ""'~
1\ I '-\ J \ ·rr· "'"
START 0 Hz RB 3.00 MHz -VB 10.0 kHz
1 \
~ STOP 100.0 MHz
ST 100.0 _sec
MIe QUAVE
STOP 1.000 GHz S1 100.0 _sec
Fig. 6.3: IF spectra is case of 140 Mb/s PRBS a) ASK and
b) CPFSK mOdulation with modulation index of 3.
63
6.2 BER measurements
In Fig. 6.4 and 6.5 the eye-diagram after post detection filtering and the Bit Error Rate
versus the optical power received For the ASK case are shown, respectively. The
sensitivity at a BER of 10.9 is -47.2 dBm, which is 4.2 dB better than the value in the
specifications.
,----,----,.----,.----,---- ~-------,----- ---- -~-- ~~----
I----+--+----+--+----+----t---+---j--~ -.- ---.-~
TiII,e!) ••• Me1" 2.00 n./dlv
Fig. 6.4: Eye-diagram after post detection filtering.
64
8ER versus OPTICAL POWER RECEIVED
-3 x
x -4
-5 x x
'" It w m '-'
-6 (Jl
0
-7 x
-8 x
-9
-10 '--_..L-_---'-_-'-_---'-_--' __ L----_..L-----l
-55 -54 -53 -52 -51 -50 -49 -48 -47
OPTICAL POWER [d8m 1
Fig. 6.5: Bit error rate versus the optical power received in case of ASK modulation.
6.3 The video codec
For demonstration purposes a 140 Mb/s video codec (coder-decoder) has been developed.
The coder part converts a standard analog video signal into a 140 Mb/s data signal. This
65
signal can be used as an input data signal for the transmitter of the phase diversity
system. The clock and data output signals of the receiver of the phase diversity system
are to be used as input signals for the decoder part of the video codec. The decoder part
converts the received 140 Mbps data signal back to the original analog video signal.
DC Video
Sync_ Scram-Video Multi- Data
res-ADC bIer out In plexer
torer AudlO----';
CLKB ,
CLK 14MHz 14QMHz
Timing
Fig. 6.6: Block diagram of the coder.
In Fig. 6.6. a block diagram of the video codec is shown. The analog input video signal
is amplified and clamped by a DC-restorer circuit and then fed into the video Analog to
Digital Converter (ADC). The sample rate of the ADC is 14 MHz (CLKS). The 8 bit
parallel output of the ADC is given into the multiplexer. Here one bit of a 14 Mbps serial
data stream for audio signals and one synchronization (sync) bit are added. After parallel
to serial conversion the 140 Mbps serial data stream is scrambled by a 5-th order basic
scrambler to avoid baseline wander in the receiver and to give the baseband signal always
enough transitions for the clock recovery circuit. In the decoder part of the codec the
reverse operations take place.
Figure 6.7. shows a block diagram of the video decoder part. The 140 Mbps serial data is
first descrambled by a 5-th order descrambler and then shifted into a shift register. An 11
bit word is loaded into flip flops and a sync detection circuit detects whether the first and
the last bits are sync bits. If so the frame is synchronized and after 10 clock periods a 11
bit word of the shift register is loaded again. If not the frame is not synchronized and
after II clock periods an II bit word is loaded into the flip flops. This shifting of the
frame continues until synchronization occurs. The video bits of the flip flops are shifted
66
into the Digital to Analog Converter (DAC). The output of the converter is buffered.
De- De- h-sync.P.
Data Video Vide o scram- multi- Buffer
out In DAC bIer plexer !-">Audi
eLK I
140MHz
eLK In
Timing
Fig. 6.7: Block diagram of the decoder.
Some rough specifications of the video codec are:
Analog video input/output voltage
Analog video input/output impedance
Digital sampling rate
Coding
Digital word structure
Scramblingl descrambling
Digital transmission rate
Digital input/output voltage
1.0 Vpp
75 ohms
14.0 MHz
8 bit linear PCM
8 bit video, I bit audio, I bit sync
5-th order
140.0 Mbps
ECL-Ievel NRZ
67
References
[1] Siuzdak, J. and W. van Etten
BER evaluation for phase and polarization diversity optical homodyne receivers
using noncoherent ASK and DPSK demodulation,
Journal of Lightwave Technology, Vol. 7 (1989), no. 4, p. 584-599.
[2] Kazovsky, L. and P. Meissner, E. Patzak
ASK multi port optical homodyne receiver,
Journal of Lightwave Technology, Vol. LT-5 (1987), no. 6, p. 770-79l.
[3] Garrett, I. and G. Jacobsen, E. Bodtker, R. Pendersen, J. Kan
Weakly coherent optical systems using lasers with significant phase noise,
Journal of Lightwave Technology, Vol. 6 (1988), no. 10, p. 1520-1526.
[4] Betti, S. et al.
Effect of non-Lorentzian lineshape of a semiconductor laser on a PSK coherent
heterodyne optical receiver,
Electron. Lett., vol. 23 (1987), no. 25, p. 1366-1367.
[5] Franz, J.
Optische iibertragungssysteme mit iiberlagerungsempfang,
Berlin: Springer, 1988.
[6] Foschini, GJ. and G. Vannucci
Characterizing filtered Jightwaves corrupted by phase noise,
Trans. on Information Theory, vol. 34 (1988), no. 6.
[7] Kazovsky. L.G.
Impact of lasel· phase noise on optical heterodyne communication systems",
J. of Optical Communications, vol. 7 (1986), no. 2, p. 66-87.
[8] Shanmugam, K.S.
Digital and Analog Communication Systems,
New York: Wiley, 1985.
[9] Krom, W.H.C. de
Optical Coherent Phase Diversity Systems,
Eindhoven University of Technology, 1992. Doctoral Dissertation.
68
[10] Kazovsky, L.G. and O.K. Tonguz
ASK and FSK coherent lightwave systems: a simplified approximate analysis,
J. Lightwave Technol., vol. 8 (1990), no. 3, p. 338-352.
[II] Foschini, G.J. and L.J. Greenstein, G.V. Vannucci
Noncoherent detection of coherent lightwave signals corrupted by phase noise,
IEEE Trans. on Commun., vol. 36 (1988), no. 3, p. 306-314.
[12] Thylen, L.
Integrated optics in LiNbO recent developments in devices for telecommu
nications,
J. Lightwave Technol., vol. 6 (1988), no. 6, p. 847-861.
[13] Boom, H. van den
A Frequency Locked Loop for a Phase Diversity Optical Homodyne Receiver.
Proc, 16th European Conference on Optical Communication, Amsterdam,
September 16-20, 1990. Volume 1.
Amsterdam: PTT Nederland and Philips Research Laboratories, 1990. P. 377-380.
[14] Siuzdak, J. and W. van Etten
Heterodyne ASK multiport optical receivers using postdetection filtering,
Journal of Lightwave Technology, Vol. 8 (1990), no. 1, p. 71-77.
[15] Krom, W.H.C. de
Impact of local oscillator intensity noise and the threshold level on the perfor
mance of a {2x2} and {3x3} phase-diversity ASK receiver,
Journal of Lightwave Technology, Vol. 9 (1991), no. 5, p. 641-649.
[16] Krom, W.H.C. de
Sensitivity degradation of an optical {2x2} and {3x3} phase-diversity ASK
receiver due to gain imbalance and nonideal phase relations,
Journal of Lightwave Technology, vol. 9 (1991), no. II, p. 1593-1601.
[17] Etten, W. van and 1. van der Plaats
Fundamentals of Optical Fiber Communications,
London: Prentice Hall, 1991.
69
[18] Krom, W.H.C. de
Impact of laser phase noise on the performance of a {3x3} phase and
polarization diversity optical homodyne DPSK receiver,
Journal of Lightwave Technology, Vol. 8 (1990), no. 11, p. 1709-1716.
[19] Etten W. van and H. van den Boom
Fused Biconical Taper Optical Couplers. Procedings symposIUm Sensors and
Actuators, Enschede, The Netherlands, November 15-16, 1990. Ed. by A.
Driessen.
Deventer: Kluwer, 1990. P. 213-216.
[20] Etten, W. van
Coherent Optical Fibre Communication,
Tijdschrift voor het NERG, part 55, Volume 3 (1990), p. 89-97.
[21] Siuzdak, J. and W. van Etten
BER performance for CPFSK phase and polarization diversity coherent optical
receivers,
Journal of Lightwave Technology, vol. 9 (1991), no. 11, p. 1583-1592.
[22] Soldano L.B. and M.K. Smit, A.H. de Vreede, 1.W.M. van Ufelen,
B.H. Verbeek, P. van Bennekom, W.H.C. de Krom, W. van Etten
Newall-passive 4x4 planar optical phase diversity network. Proc. 17th European
Conference on Optical Communication, Paris, September 9-12, 1991. Post
Deadline Papers.
Paris: Societe des Electriciens et des Electroniciens, 1991. P. 96-99.
[23] Henry, C.H.
Phase noise in semiconductor lasers,
1. Lightwave Technol., vol. 23 (1986), p. 298-311.
[24] Kuwahara, H. and M. Sasaki, N. Tokoyo
Efficient coupling from semiconductor lasers into single-mode fibers with
tapered hemispherical ends,
Applied Optics, vol. 19 (1980), no. 15, p. 2578-2583.
70
[25] Groten, M. and W. van Etten
Laser line width measurement in the presence of RIN and using the
recirculating self heterodyne method,
Eindhoven: Faculty of Electrical Engineering, Eindhoven University of
Technology, 1992. EUT Report 92-E-262.
71
List of abbreviations
ADC
ASK
BW
BER
CCTV
CPFSK
DAC
DFB
EHP
ER
FLL
FSK
FWHM
IBeN
IF
IMIDD
lOP
LO
LPF
OlE
PRBS
RIN
SAP
SMF
SNR
TUD
TUE
UT
VVM
Analog to Digital Convertor
Amplitude Shift Keying
BandWidth
Bit Error Rate
Close Circuit TeleVision
Continuous Phase Frequency Shift Keying
Digital to Analog Converter
Distributed FeedBack
Electron Hole Pairs
Extinction Ratio
Frequency Locked Loop
Frequency Shift Keying
Full Width Half Maximum
Integrated Broadband Communication Network
Intermediate Frequency
Intensity Modulation/Direct Detection
Innovatief Onderzoek Programma
Local Oscillator
Low Pass Filter
Optical/Electrical
Probability Density Function
Pseudo Random Binary Sequence
Relative Intensity Noise
Stress Applying Part
Single Mode Fiber
Signal to Noise Ratio
Technische Universiteit Delft
Technische Universiteit Eindhoven
Universiteit Twente
Vector Volt Meter
72
Appendix 1 Specifications
The specifications of the phase diversity are:
optical wavelength : 1535-1540 nm
receiver : {3x3}/{4x4} phase diversity
penalty compared with an
ideal heterodyne receiver : {3x3} : < 0.4 dB / {4x4} : < 0.25 dB [I]
modulation : ASK / CPFSK
bitrate : 140 Mb/s
modulation-index CPFSK :>7
demodulation CPFSK : single filter method
IF : 300 MHz
IF-bandwidth : 650 MHz
LP-bandwidth : 140 MHz
laser linewidth : 30 MHz FWHM penalty < 1.4 dB [14]
RIN : < -135 dB/Hz penalty < 7.0 dB [15]
front end noise : 50.10-24 A2/Hz
responsivity front end : 10 V/mW (R".., = 50 Ohm)
polarization mismatch < 10 degrees penalty < 0.15 dB
extinction ratio ext. mod. > 10 dB penalty' < 1.2 dB [9]
phase mismatch {3x3} hybrid < 5 degrees penalty" < 0.5 dB [16]
gain imbalance {3x3} hybrid < 10 % penalty" < 0.8 dB [16]
phase mismatch {4x4} hybrid < 20 degrees penalty' < 7.3 dB [16]
gain imbalance {4x4} hybrid < 10 % penalty" < 1.2 dB [16]
BER : 10-9
sensitivity {3x3} ASK/CPFSK : -43 dBm / -40 dBm (5 dB margin included)
sensitivity {4x4} ASKlCPFSK : -37 dBm / -34 dBm (5 dB margin included)
'can be optimized when mismatch is known.
2f" I" 'I" 'I T" 'I" I" OJ, " "'1 ~ ] :,;1.5'1" ~
~- ~ ~ Iy --------- 1 > " ]
0.5 ~
o o
1 I !""",!",., •• 't'~~
2a 40 60 ea Current (mA)
~ ,
-d3id'$[] I 6 I
Au HETA.L1Z£O ALUHINA
Gl
(L~ ;\1 1
1 , ~
100
J: E
~ , Q. ~ , o ~
" C1
-'
:I~""'" I~I' "d~,/ ""'~>1 , I ~
3 ! / , ' / /
21:" / ,/ i : I i
/ / -; / ' 1 ' I / '
I / ~ , '
a II I i "3 o " .. ! I",! t t, It, I ,j
(") > :r -= ~ -= " tD ft ::I ::I, Q,. '" _. ::t,
~ " '" N 0 -. -:r tl <>
q po § S '" §, </l - ::r ft tD '"' (\l
or Ul" '"
20 10a 40 60 Current (mA)
ea
~ 0 ...., g. (\l
"0 -J
'" ::r _ . ...... -. -= </l
Temp. (C) : 25 65
tl cil
Ith (rnA) : 18.8 40.4
Es (mW/A) : 108 74 ...... po </l Rs (Ohm) :7.4 7.6 (\l '"1 </l V30 (V) : 1.07 1.06
lop (rnA)
74
Optical spectrum of the transmitter laser
..... \l I
Cap) A'+ sua,+uI
[S) m If)
[S) I\-If) .....
" E
[S)[ If)'-' If) ..... ..c:
3: E
-f1~ mlf) [
[S)(l)CI: E [T) ~.
If)(l)en ..... > .
to I'-If)
:s: u
[S) ..... If)
If) (\j
.....
(f)
:::J 0
If) ..... :::J I z
H
f-Z 0 U
E en c -0
(\j ~
en (\j
(T) '" If) I ~
s:. +' m c Q) c
0 Q)
> (I)
to (I)
3: Q)
s... -0 D-
Q) Q..
+' ;:)
s:. (f)
m Q)
Q) -0 3 0
:L s... Q) Q)
3 -0 0
(L (f)
Linewidth of the transmitter laser
LLJ :::> a:: = = ~=
U 0------<
=
:j
~ ~
E 1 = LJ = :::>
LJ 0------<
F CSJ 0 CSJ CSJ '-
-J .,-i = .,-i LJ ru
[J I LLJ CSJ I-- CSJ
---.J I--
'I ,-,-"
75
~
...::i V ~ ~
~ ~
;;:
~ ~
k#
.. N
i = = CSJ CSJ
CSJ lJJ
D-
= I--(f)
N
= = I--cc a:: I--(f)
U Q.J
(J)
E
CSJ CSJ
CSJ lJJ
I--(f)
N
= ...::,::
CSJ CSJ
= :::>
*
N
= = CSJ CSJ
.,-i
= cc *
2["1" ",i,," "I'''''''-ry~iil'''" 'I~
>' 1.5
• 01 • +'
g
0.5
at .... ..... .. .. , o 20 40 60 se 100
Cu,..,..ent (mR)
~ ,
£1 kd'$[] I 6 I
Au HE TA.UlEO ----ALUMiNA
Q)
~
J: E
+' , Q. +' , o
+' .<: 01
-'
5 r .. 0" .. 0 " 0" ." • '0)' , 0 ' .... 'j
4J / ~ .I /~ I / 3
3 / // i ~ ; / 1
t / / ,
2 / ./ -= L / j
J=(~< .. " .. o .1 o 20 40 60 S0 100
Cu"',..ent (mR)
n ::r ., ~ .. ::J. '" ::to (')
'" 0 ...., ~ ::r " ~ e:-o '" D. = ., 0 '"' 0;-'" ~
-.l 0\
Temp. (C) : 25 65
Ith (rnA) : 23.4 49.3
Es (rnW/A) : 151 103
Rs(Ohm) : 5.0 5.5
V30 (V) : 1.01 0.99
lop (rnA)
" rn 0
t-
~ -,J ; ~
L
~ -
201 I'
[ -30 I
H E II -40
~I!.LI~JjMlMlulkaelllll' mIllllIlIUN!lIMIII· -50 -
1490 1510 1530 1550 1570 Wave 1 ength (nm)
CONTINUOUS 25 C 67.9 rnR 5.0 mW
Power weighted Wavelength 1538.2 nm
Side Mode Suppression -33.2 dB
~ g . ., -~ q c: 3 0 ....,
~ -::r '" g e 0
'" £ ;" 0-....
~ .... -J -J
1590
78
Linewidth of the local oscillator laser
! ~ I ! = I
! = l I = ~ -_. -cc.- .-~----- .... ,-- - - . i u
rT.u:~ I I cs: ~
= I ~= Lfl
~·--f.---I---.J---+----+---+----'~ -----~-. u_ ~ ~ , i , ! ,
r--+--1i- ) ~ i I ~---+--+---------. --- -~-~--.- --- --3~----··- .
~
,
! , I ,
, I I ~ ,
i
I ?~ I
r----t---t----t----t---;.F=~-+----r---j-.--i_--- ---
! ! , I , I , , i , :
N = ~ = = = ::> * ~I I I ##=-
-01= ::> I ~ N N
~1~·~=4f.----+--~~~~~-+---~---.J---+I----+---4---4:: ::
..,-;1 -0 .~ = ~~_.~-~_-:-'~----_._=_.-.. _---r:,~-•...... -.~. -. - ! -]---1= ~ ~
79
Appendix 3 Data of the GEe optical amplitude modulator
FEATURES
• Wide bandwidth
• Low insenion loss
• Low drive voltage
• EMC/EMI-free transmission
APPLICATIONS
• Microwave frequency optical links
• Optical signal processing
• Optical sensor systems
• Coherent transmission
Integrated optical modulator with lid removed
The usc of external modulators with CW laser diodes have considerable advantages when compared with direct modulation; integrated optical modulators eliminate dynamic laser chirping, have gigahertz bandwidths with nailer frequency responses and allow greater signal to noise ratios. This makes them panicularly suitable for usc in communication systems where transmission and manipu1ation of high bandwidth data rates arc required.
TYPICAL OPERATING PARAMETERS AND SPECIFICATIONS
Operating wavelength Bandwidth Drive voltage (Vrr) Electrical impedance Electrical return loss Extinction ratio Optical insenion loss
RF connector Package dimensions
Standard Fibre: Input Output
Environment
Notes
~
(I) (2)
(3)
Y-3'i-5170-01 .L!.ni.ts
1.3 ~m 2.5 GHz 3.0 V 50 Ohms 10 dB 25 dB 6 dB
SMA SMA 6Ox35xl5 42x25xl5 mm
I metre York HB 1250 polarisation maintaining I metre GPT SMOg standard monomode
Normal laboratory conditions
(I) Also available for 1.5J.lm wavelength. Specification as above with exception of the drive voltage which for Y-35-5370-02 is 4.0 V and Y-35-5600-02 is 8.0 V.
(2) Electrical bandwidth ± 3 dB measured using a modulated optical microwave link. (3) Devices can be supplied with other fibres. The devices are polarisation dependant and if
monomode input fibre is chosen, a fibre Polarisation Controller is an accessory available through GEC Advanced Optical Products.
80
Appendix 4 Circuit diagrams
I Laser current control and modulator circuit
2 display cicuit temperature, laser current and -SV source
3 Temperature control circuit
4 Frequency control circuit
5 Optical/electrical converter module
6 IF section
7 Clock and data recovery circuit
8 Rear panel connector and laser connections
9 Front panels
27 > -< Eel s
R22 T R23
SK S1
2 ")
02 DF8
"¥'" ~ ~ lASER 03
lcs
OS
.17 BAV ~ ~ BAV ~ ~ } 99 ~
~ r2n 99 ~
~ ==
R1 ...:: ...; 22n
l1 ,. C3 11 22pF R6 • A 1r 4<7
C1 • 3?.. 01 .470uH 04 ~
,. J;s os ~ Ii ~B :;: 5 01 k BAV
'- BAY -' '- : ,. L.4 L.4 E K '.5 99 ~ ~ r2 99 ~ ~ 6uB 02 - 22n
R5 I~ I~
JlO2~17>-' - +
6 0 j:;IEF ) ,. BCW~
12,5Y 2
" BAY 99 608 Be B46B I C 1 03 04 R19 - - -- FIJ~ I; \i TCA 520B R12,,-20> BFR 5.6
L M LM C
G ,.5 r 93 (I
00 -
385 03
93 ~ 1K R9 ~
R20 Sk BFR
10 ,I S.
C4 • '.5 Q5
Ie2 :~ H ~> P
1Su R4 R7 RB R14 R16 R17 021
R10 R15
,.5 T6U8 33K 1k 22 " 1K 1< 560
51 - 27 "")
1 Laser current control and modulator circuit
82
LASER CURRENT
TEMPERATURE
.11V 101
101
13
R~6
101
IOn U
Cli t24 114~ R~' R~I R~1 ..
lOOn 22n 1. 10K 19K1 1[21 31 101
- 15V 25V 1
... 01 1N400S
-5 VOLT~GE SOURCE -------------~---
-11V L~ 317 0 -SV
31 21
Ie 2 R25 08 lN4005
G
-3, ]V Cl0 ell R29
4 1Su R2G 68n 101 15. 68n
2lV e12 en 1K 25V
• 6ua 68n • m 2N412" 15.
25V m ~)
1
2 display cicuit temperature, laser current and -5V source
83
2 C34
cn l8L05 0
Ie 5 R50 18n T
-L R5l G R51 "!' C30 10K
6K8 6K8 lu
3
C31 R52 lK82
R66
~JOK .BV .15V 2
31
Ai?
10K 1 C~l
1DOnF I .,. -6V -15V
3 Temperature control circuit
84
16
2' ~~.1~5~V ______ -r __________ -r~
RB2 CSO
Ie 12 '"" Ie 13 RBl
100n 410 -470 22u
in 6 out iI'r RF ~AR RF >,:::...L-l-il--. __ -I-~ SP B 7 5 5 r--,- '" /
9111 GRO GRD
MIA 1101 100n
.,. ~AR 6
o v
("2l. ,15V '-"-J
lcss
lDn 16
.. .....!... ~ 01
H
...i. S os 13
..l... 12 C DC
-1... f.!.LiI 0 DO
f.!.L RCO ,......!... ENP
10 EN1
~ 2 W CLK
~ 9 LOAD
C67 r ~ CLR
10n1 Ie 16174HC161
.,.
""v 16
1..2... flLl-A 01 I...i. 13
B DB 12.. 12
C DC 1-1... 11
0 00 74HC04 RCO ~I-
L....2.. 10
INO ENT
2 CLK
9 ~ LO~D
L-...!., CLR
Ie 17114HC161 -----.-.-----
1 o V
31 -15V
ln
..
mT 1D'l
i I
"11-
12
13
14
1S 1S
11
1S 19
hold " reset 22
.1L
.1!... ~ .2.!.. 22. 2!...
....!.
...1!.
....!. ,.!!.
RBJ
1S'
PROG
OB.O OB.l
OB.2 OS.J
DB.4
OS.5 OB.6
OS.l
P2.0
P2.1 P2.2 P2. J
P2 .•
P2.5
P2.S P2.1
-RO
iR PSEN
AlE
Ie 19 - ----
10 Ie 14
7 12 13
C65 1n C66
40
X1 WJ 187481
3 X2
4 RESET
55 h1-,..L TO ~ T1
o-!--INT
Pl.D 17
2B P1.1
P1.2 P1.3
29 3D 31
P1.4 12
P1,5 33
P1.6 P1.1
14
EA f-l-20
4 Frequency control circuit
C70 II33p
.ill 78L 10 ~ IF" E!] ,15V T C:
11 IC 19
I~on II 10n ;!r-
,~
13
~ 12 10AC 081
14 RSS 11 C14
10 10' :~ 9 ~22n B
RBS
7 f!--Q kJD 6 5 ~
10' Ie 20
RB< 2 116~~ 014 -.... ~ ~ C75
1N914
'-~ 10,
51 i :~ Cl2
1"
~
<- • .9V
FCI PC
co<
,
R1no
"
~ 100n
91&0
R1Q04
"
Rl01
"
• POC
85
A1Dl 510
"
~ ~J.. 100n
~ 100n
-,-..... " ",
10 12 ,C 21 co. "SA
22110 15 Ri09
5 IC 20 16,5
CB7
<- • - SV -'- " R1D'5
I 10k. ' 1 _____________________________________________ 1
<- A .911 ..J.... I
Rl1i Rl1l A1'" Al1' 510
" " " " R120
fel PC
eBB
~ ~...L ~ ~ " .,.. 100n
~ "T iDDn 100n 22, 100n
1 • 10 12 c •• 8T&0 poe 2200 15 1178:5 R121
5 IC 22 16,5
~oon C95
~
tt:: GA IN
<~u_=-~5V~~=--'--__I:!:jill-fu1!.J__j1 <- B A117 R119 I I
-r- 10k i
<- A .9V
Fel PC
<- • - 5V
31 -15V
C1D4
10, •
---------------------------------------------j
C10
R12-t
"
~ 100n "!'"
Bl&O
6
"
R1:?)
" ~
100n "'"'!"'
A12S
"
~ 1Don
10 12
A 127
"
poe 2200 OSA
15r----,0785
5 Ie 24
~,oon ... "
GAIN
..,..
R12!! A13l
Ie 25
"
'"
510
R132
C 102
C1aa
,
A1::!!!
: _____________ ~~ ______________________________ J
2
5 Optical/electrical converter module
s., R11D
18,5
R1ll
111,5 SO,
..,..
.. ,
R123
16,5 ""
86 1---------------------------------------------------------------
------------j IN A I lUl2 .. l3/l2f1
v' , , -----------_ ..
Lor P~SS F I L TE~
.- ...................... i IN A. I l4/l21l LS/J211
, , +:
Lor P~SS F I L TE~
100'~ RUO ,
~
10a'Q AH5 ,
or
C119
~H 10
C122 1DOn 10(]n
~GC1out H'0' e116 560
HfTlll-l ...... ~~"-' 7 1Dn ,-" .............. -,
R1H ,.
I Ie 29: 7
1--- IC29 IVA -----
... 0511B
AGC AMPLIFIER
, , ' .. I£. 11!!7
I IVA or 05118
AGC AMPLIFIER
, 4' I C 29' "8r;;'-1--"6r
C120 I ... TC121
100, .. 1 . .'00' V WIXER -:r
C12~ ::: • 10:0n *
tHC",' r",'0=,-",",:;--'
(;:136
100, ,LJ 10,===, 'f1 AGe2ou! ,......--,
7 .- ................ -. R149
1K
1SWA
e137 out B
100n or
1----------------------------------------------------- __________ ~
1- ...................... ,
IN' , L?132.'
,
or : I'" , v , ,-----------_ ..
Lor PASS FILTE~ CH2
100'~ R1S0 ,
..
, 4, , I r 00,7 -r· ~: J.. 05118 ...
AGC AMPLI F I ER
tH1
10lln
10
C 150
,LJ 10, 'fl AGC30ut
7 ________ _
IAW9tD1B
R15" ,.
C154::: C 1110n 1
or
, , , 1 ______ ----------------------------------------------- __________ _
6a IF section
m
10K
AGC offset
A163
1K
C153 1u
Ie J5
6
c- - A f-'---11
21:: 10K
R1 DB
'"
2K 10K
R19B
'"
51
R167
"
'"
87
110K
"
7 Ie 36 IIV2 SAT B5
2 015 016 017 019
AGCln'
C154
1u
GAIN
10K
3,
2
2
AGeln2
C165
I.
AGelnJ
C166 6b
1u IF section
88
La L9 L 10 L 11
SUA 120 17n+33n 56n+3911 5611+39n 1811+33/1 S~A
INPUT A
1157 1159 "='"
A176
61 ·
· LOW PASS fiLTER
S~A 120 ---------------
INPUT B
A17a
· 61
· S~A 120 lK 590
INPUT C
A1ao C160
· 61
· 10n + lK 1C163
590
- -"='" 1 On L13
Ie 31 L 12 Ie 38 1. 1. -----
I SMA SUA SUA
f:~, moo,,"", C162 C164
100. lOOn "='" "='" "='" lOOn
. .
6c IF section
R213 R211
." r-...L:' __ "',' 1
1
U U42
~~~':"r-~-~JC_+"'!1' ~ , P ',',10 ~'.;.".;2_-+-,,'<t' )" , ..... ~71 V
~ 1'" .:' l" i, .0. I A214
I H7 Lf " 5'll
IN Jl E, Li Ai
'i , , , , 'M our 1-
11215
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89
~ ,
hel 'cd
D' 10,
~ 'I ;;
" ~ '---r
r-Uff '---'
"" • .. cunl
f-A2l3
'" A212
I ~C179 " , . , 10.
I
CLOCK t'M -If"" " '/1--"",+...1.1
_ '-I 14~ c ~_
.1!..
~ 10n ~nH
'00
' .. ATA~IIII. lIel)8
"' 11 10n R229
10'
'."
~f-!:++-' IIIC10116
12 ~ 10 t-- ~ 15
11 8"~ 4 1) ,WL,
-;:-' ".
IICtlU"
U46 1
IIC1648
R2B
'"
,
,
" I~ • .!........ U
~ ~O ..!!lll... r-' 'r;H •
l~~ u~s 1
IIC12(HO
'F
'--
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~ YAII
". " I-
m
i ~ R2H
".
"" ~C1Rl ~ I
r! , I'O' R235 ,,. , ..
m
(182
'" '~H-!Eiil
-rifu:J.. Ll. .~' , 1,
- U18
11215
11.
'"
1 I F156
."
,
7 Clock and data recovery circuit
~I Rl22
" ~ R221 R226
'" S60
e17]
Hf t+ 1000p 1000p
RZ25 R22?
'" '"
H H C1H e176
lDDOp 1000p
GND 1
+1SV 2
NTC/GND 3 BROWN -4
NTC/- 5 GREEN
6
NTC31 + 7 WHITE -8
9
10
GND 11
Vamp 12
13
14
15
+5V 16
17
1B
19
+ IILD 20 WH ITE -21
22
23
-I/LD 24 BLUE
25
GND 26 BLACK -- 5V I LD 27 BROWN -PELTIER 2B WHITE
29
PELT I ER 3D PINK -- 1SV 31
GND 32
-I
-
r - ~r
-I
I
-'-~
r
I
~
90
"" 5 90
eOr-3
70 2
60 1
V l.
r-- '-___ +-__ NTC
W
8 Rear panel connector and laser connections
. · · · · SET UP VA.LLE
TEMPERATURE
D · U3
· U2 0
TEMPERATURE CONTROL
D · ._, . """ CO , · U, . . · .- ""
, CURRENT I rrVII
- . ~ I II I
. . · · · ·
· · · · . • . • . • • • • • . . . • AS< EA.-iS"lI: CLOCK8dlllTA
SET UP VALUE FREQUENCY J )( 3 COJPLER I FI PDF RECOVERY
TEMPERATURE
c=J aJNTROL
0 D 0 , ''''"'
, · U, T ....... 0 • · · U2 0
, , 1 •• - 0 1 • · 1
D 0 0 ~:
TE~ERATURE '"" • • 0 /'
CONTROL •
D 2 5 • 2 0
- · .T_ Do\TA OUTPIJ
" .. D 0 , . ON ,
· U, · 0 0 ""'M . . · 0 , ,
· ''''''" "" , - , • • 3 0
CURRENT J mA INPUT 0"""'" Wi< 0 0 " - 0
""" - . ~ I II II II II II II II I
. · · · · . . . . . . . . . . . .
9 Front panels
92
Appendix 5 Source code listings of the frequency locked loop
Listing for the zero IF case:
ORG 0
MOV R2,lOH ; stapgrootte
MOV R3,80H ;instelling voor DAC
MOV R4,80H ; tellerstand
LO: MOV A,R4 ;vorige tellerstand (RS)
MOV RS,A ;wordt tellerstand
MOV A,R3 ;R3 wordt R3 + stap
ADD A,R2
MOV R3,A
OUTL Pl,A ;DAC
MOV A,lH ; reset teller
OUTL P2,A
MOV A,3H ;enable teller
OUTL P2,A
MOV A,2H ;hold teller
OUTL P2,A
IN A,PO ; lees teller
MOV R4,A
CPL A
INC A
ADD A,RS
JC LO ;jrnp indien juiste richting
MOV A,R2 ;stap wordt -(stap)
CPL A
INC A
MOV R2,A
JMP LO
END 0
93
Listing in case the IF has to be 300 MHz
ORG 0
MOV R2,lH ; stapgrootte
MOV R4,lBH ;gewenste tellerstand
MOV R3,80H ;initialisatie DAC
MOV A,R3
OUTL Pl,A
MOV A,R2 ;berekening -(stap)
CPL A
INC A
MOV R5,A ;-(stap)
LO: ANL P2,OIB ;reset teller
ORL P2,llB ;enable teller
NOP
NOP
ANL P2,lOB ;hold teller
IN A,PO ; lees teller
CPL A
INC A
ADD A,R4
JC Ll ;verlaag stroom indien
MOV A,R3 ;teller te hoog, anders
ADD A,R2 ;verhoog stroom
MOV R3,A
OUTL Pl,A
JMP LO
Ll: MOV A,R3
ADD A,R5
MOV R3,A
OUTL Pl,A
JMP LO
END 0
Eindhoven University ,:,f Technol<xlY Re,3ean;h Reoorts 155N 0167-9708 Ccx:ien: TEUEDE
Fl1cu1ty of Electrical F.nqineering
12441 . Maril" and Ton van den Boom. Ad Haruen OF CLASSICAL AND MODERN CONTROLLER -DESIGN: A case study.
EUT Report 90-E-244. 1990. ISBN 90-6144-244-3
12451 Berg. P.H.G. vande ON THE ACCURACY OF RADIOWAVE PROPAGATION MEASUREMENTS: Olympus propagation experiment. EUT Report 90-E-245. 1990. ISBN 90-6144-245-1
12461 Maaqt. P.J.!. de A SYNTHESIS METHOD FOR COMBINED OPTIMIZATION OF MULTIPLE ANTENNA PARAMETERS AND ANTENNA PATTERN STRUCTURE. EUT Report 90-E-246 1990. ISBN 90-6144-246-1
12471 Jozoiak. L. and 1. Spmova-Kwddltaal DECOMPOSITlONAL STATE ASSIGNMENT WITH REIISE OF STANDARD DESIGNS: Using counters as sub.achines and uSln9 the method of maXimal ad]acensies to select the state chains and the state codes. ElIT Report 90-E-247. 1990. ISBN 90-6144-247-8
11481 Hoellmakers. M"'))",a:~n:d,,;,).;.~MA', ,~~~¥", DERIVATION AND UI, THE SYNCHRONOUS MACHINE WITH RECTIFIER WITH TWO DAMPER WINDINGS ON THE DIRECT AXIS. EUT Report 90-E-246. 1990. ISBN 90-0144-146-6
12491 Zhu. Y.C. and A C.P.M. Bach. P. Eykhoff MULTIYARIABLE PROCESS IDENTIFICATION FOR ROBUST CONTROL. EUT Report 91-E-249. 1991. ISBN 90-6144-249-4
mOl PfaftenhOfer. P.M. and P J.M. Cluitmans. H.M. KUlpers £MDABS: Design and formal specification of a datamodel for a clinical research database system. EUT Report 91-E-250. 1991. ISBN 90-6144-250-8
12511 EiJndhoven. J.T.J. van and G.G. de Jonq. L Stok THE ASCIS DATA FLOW GRAPH: Semantics and texiual format. EIIT Report 91-E-251. 1991. ISBN 90-6144-251-6
12521 Chen. J. and P.J.!. de Maagt. M.HA.J. Herben
1253)
(2541
12551
WIDE-ANGLE RADIATION PATTERN CALCULATION OF PARABOLOIDAL REFLECTOR ANTENNAS: A compar,tlvP study. EUT Report 91-E-252. 1m. ISBN 90-6144-252-4
Haan. S.W.H. de ! PWM CIIRRENT-SOURCE INVERTER FOR INTERCONNECTION BETWEEN A PHOTOYOLT!IC ARRIY IND THE UTILITY LINE. RUT Report 91-E-253. 1991. ISBN 90-6144-253-2
Velde. M van de and PJ.M. Cluitnans EEG ANALYSIS FOR MONITORING OF ANESTHETIC OEPTH. ElIT Report 9I-E-254 1991. ISBN 90-6144-254-0
Smolders. A. B. AN EFFICIENT METHOD FOR ANALYZING MICROSTRIP ANTENNAS WITH A DIELECTRIC COYER USING A SPECTRAL DOMAIN MOMENT METHOD. EUT Report 9H-255 1991. ISBN 90-6144-25H
Eindhoven University of Technology Research ReDOrts ISSN 0167-9708 Ccxien: TEUEDE
Faculty of Eiectrical Engineerioo
(2561
(2571
1256)
(2591
12601
12611
(2621
(2631
(2641
Bac'x. A.C.P.M. and A.A.H. Da.en IDENTIfICATION FOR THE CONTROL OF MIMO INDUSTRIAL PROCESSES EUT ~eport 9H-256. 1991. ISBN 9H>144-256-7
MOdgt. P.J.!. de and H.G. ter Morsche, J.L.N. van den Broek A SPATIAL RECONSTRUCTION TECHNIQUE APPLICABLE TO MICROWAVE RADIOMETRY. RUT Report 92-E-257. 1992 ISBN 90-6144-257-5
Vleeshouwers. J.N. DERIVATION OF A MODEL OF THE EXCITER OF A BRUSHLESS SYNCHRONOUS MACHINE. EUT Report 92-E-258. 1992. ISBN 90-6144-258-3
Orlov. V.B. DEFECT MOTION AS THE ORIGIN OF THE llF CONDUCTANCE NOISE IN SOLIDS. EUT Report 92-E-259. 1992. ISBN 90-6144-259-1
ROOllackers, J.E. ALGORITHMS FOR SPEECH CODING SYSTEMS BASED ON LINEAR PREDICTION. EUT Report 92-E-260. 1992. ISBN 90-6144-260-5
Boom, T.J.J, van den and A.I.H Damen, Martin Klompstra IDENTIfICATION FOR ROBUST CONTROL USING AN H-Infinity NORM. EUT Report 92+261. 1992. ISBN 90-6144-261-3
Groten. M. and W. van Etten LASER LINEWIDTH MEASUREMENT IN THE PRESENCE OF RIN AND USING THE RECIRCULATING SELF HETERODYNE METHOD. EUT Report 92-E-262. 1992. ISBN 90-6144-262-1
Smolders, U. RIGOROUS ANALYSIS OF THICK MICROSTRIP ANTENNAS AND WIRE ANTENNAS EMBEDDED IN A SUBSTRATE. EUT Report 92-E-263. 1992. ISBN 90-6144-263-X
auditory evoked potential pal terns. EUT Reoort 92-£-264. 1992. I5BN 90-6144-264-8
(2651 Wellen, J.S. and F. Karouta. M.F.C. Schemmann, E, Sma lbrucrqe , L.M.F. Kaufmann MANUFACTURING AND CHARACTERIZ!TJON OF G!ASIALGAAS MULTIPLE QUAHTUMWELL RIDGE WAVEGUIDE LASERS. EUT Report 92-£-26), 1992. ISBN 90-6144-26)-6
(266) CluHmans. U.N. USING GENETIC ALGORITHMS fOR SCHEDULING DATA FLOW GRAPHS. EUT Report 92-E-266 1992. ISBN 90-6144-266-4
(2671 ,J6zmk. L. and U.H. van Oil. A METHOD FOR GENERAL SIMULTANEOUS FULL DECOMPOSITION OF SEQUENTIAL MACHINES: Algoritlims and lmplenent,tl"''!. EIIT Report 92-E-267. 1992. mN 90-6144-267-2
!268j Booo. H van den and W. van Etten, W.H.C de Krom. P. van Benllekom. F. HIlI]S,ens L. Niess,n, F. de Lener ~ =
AN OPTICAL ASK !NO FSK PHASE DIVERSITY TRANSMISSION SYSTEM. EUT Report 92-E-266. 1992. ISBN 90-6144-268-0