An improved design procedure for LCC resonant filter of dimmable electronic ballasts for fluorescent lamps, based on lamp model

1186 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 20, NO. 5, SEPTEMBER 2005

An Improved Design Procedure for LCC ResonantFilter of Dimmable Electronic Ballasts forFluorescent Lamps, Based on Lamp Model

Fabio Toshiaki Wakabayashi and Carlos Alberto Canesin, Member, IEEE

Abstract—This paper presents an improved design methodologyfor the determination of the parameters used in the classical series-resonant parallel-loaded (SRPL) filter employed in the switchingfrequency controlled dimmable electronic ballasts. According tothe analysis developed in this paper, it is possible to evaluate someimportant characteristics of the resonant filter during the dimmingoperation, such as: range of switching frequency, phase shift, andrms value of the current drained by the resonant filter + fluorescentlamp set. Experimental results are presented in order to validatethe analyses developed in this paper.

Index Terms—Dimmable electronic ballasts, half-bridge in-verter, resonant filter + fluorescent lamp set, series-resonantparallel-loaded (SRPL) filter.

I. INTRODUCTION

THE proper design of an electronic ballast depends on themethodology used for the determination of the parameters

employed in the inverting stage. Usually, the half-bridge inverterconnected to a series-resonant parallel-loaded (SRPL) filter isthe most common topology used as high frequency inverter forelectronic ballasts [1]–[3]. In addition to its simple circuitry andreduced costs, this structure is capable to provide desired char-acteristics for electronic ballasts, such as: high voltage duringthe lamp ignition process, preheating for the electrode filamentsbefore the lamp ignition, soft-switching for the semiconductordevices employed in the half-bridge inverter, and suppression ofcurrent and voltage dc components over the fluorescent lamp.

Several design methodologies have been proposed for this cir-cuit, focusing their analyses on the SRPL filter, in order to pro-vide an optimized choice of its parameters [2]–[13]. In practice,these methodologies are based on analyses of equivalent circuitimpedances, considering the fundamental approximation tech-nique [3], and assuming that the fluorescent lamps behave asvariable resistances [4]–[13].

In spite of the large application of these methodologies, noneof them provides a complete analysis of the SRPL filter, usu-ally omitting some important aspects during dimming operation,such as the obtaining of the phase-shift in the current drained bythe set of resonant filter fluorescent lamp, and the rms value ofthe current through the resonant inductor, which can indicates

Manuscript received September 2, 2003; revised December 14, 2004. Thiswork was supported by the State of São Paulo Research Foundation (FAPESP)and the Brazilian National Council for Scientific and Technological Develop-ment (CNPq). Recommended by Associate Editor R.-L. Lin.

The authors are with the São Paulo State University, Ilha Solteira (SP) 15385-000, Brazil (e-mail: [email protected]).

Digital Object Identifier 10.1109/TPEL.2005.854058

the levels of current stresses in this inductor and in the semicon-ductor devices.

The power processed through the lamp presents an almostlinear dependence on the phase-shift of the current processedthrough the set of resonant filter lamp. Due to this fact,an integrated circuit (IC) for closed-loop dimming controlwas presented in [14], providing sophisticated features forballasts applications. However, the prediction of phase-shiftduring dimming operation has not been clearly presented in theliterature. Also, it is very important to highlight that, usually, theswitches employed in the Half-Bridge inverter are MOSFETs,and these devices are sensitive to the square of the rms valuesof processed currents. So, the prediction of current stressesis useful in the specification of optimized parameters for theSRPL filter.

According to this context, this paper presents an improved de-sign methodology for the determination of the resonant parame-ters of SRPL filter employed in switching frequency controlleddimmable electronic ballasts, incorporating new analyses ca-pable to provide more details about the operation.

In addition, in order to develop an accurate design method-ology, it is necessary to admit a proper resistive lamp model.Regarding to this subject, the effects of ambient temperature onthe electrical characteristics of fluorescent lamps must be con-sidered [15]–[17]. Thus, the required analyses are based on thelamp model presented in [16].

II. ANALYSIS OF THE SRPL FILTER

The analysis of the SRPL filter is developed in two distinctparts, namely: ignition process, and dimming operation.

In the ignition process, the lamp is considered as an open-circuit. Thus, during this operational stage, the function of theinverter is to provide a proper preheating for the electrodefilaments, and also enough voltage levels over the lamp forthe occurrence of the first arc through the gas column.

On the other hand, during the dimming operation, the lampis admitted as a variable resistance. In this context, severaldifferent models have been proposed to describe the behaviorof this equivalent resistance. Because of its simplicity andaccuracy, the model presented in [16], for a F40T12 fluorescentlamp, will be used to represent the lamp equivalent resistance,assuming a fixed ambient temperature (T). In this way, from[16], assuming an ambient temperature of 24 C, the equivalentresistance can be written as

(1)

0885-8993/$20.00 © 2005 IEEE

WAKABAYASHI AND CANESIN: LCC RESONANT FILTER OF DIMMABLE ELECTRONIC BALLASTS 1187

TABLE ICOEFFICIENTS EMPLOYED IN (2)

where

(2)

1) value of voltage over the fluorescent lamp asa function of P;

2) value of power processed through gas column ofthe fluorescent lamp (without considering the filaments).

Table I shows the values of coefficients of (2). It is informedthat, according to [16], these coefficients are determined usinga set of experimental data.

Fig. 1 shows the curve of equivalent resistance, as a functionof the power processed through the fluorescent lamp.

After assuming the conditions cited before, it is possible toproceed with the required analyses. In addition, it is noticed thatthe IR2159x IC was used for controlling the semiconductors ofthe half-bridge inverter.

A. Analysis of Ignition Process

The analysis of the SRPL filter during the ignition processis developed according to Fig. 2. As commented before, thelamp is considered as an open circuit and the resonant elements( C and C ) are connected in series. Thus, the followingequation can be defined:

CC C

C CC

C(3)

where

CCC

(4)

The circuit presented in Fig. 2 can be represented by (5), as-suming the fundamental approximation of the input voltage

(5)

where

HBfundamental component of the

voltage applied over A and B (6)

peak value of

HB HB (7)

HB switching frequency

during ignition process.

Fig. 1. Lamp equivalent resistance during dimming operation, considering t =

24 C.

Fig. 2. Ballast equivalent circuit before lamp ignition.

In addition, it is possible to write the following equations:

(8)

C (9)

From (8) and (9)

C (10)

Replacing (6) and (10) into (5)

HB (11)

where

C(12)

1) C is defined according to (3);2) is the frequency related to , expressed in Hz.Thus, applying the Laplace transform in (11) and considering

the initial conditions equal to zero, it is possible to determinetwo different sets of equations, capable to describe the behaviorof instantaneous values of current through and voltage overC , depending on the values of HB and .

When HB

HB

HB HB (13)

C HB HB (14)


CC HB

HB HB (15)

When HB

HB

HBHB

(16)

CHB

HB

HB (17)

CC HB

HBHB

(18)

Fig. 3 presents the waveforms of , depicted for dif-ferent values of HB and . According to this figure,it is possible to note that , which can be considered prac-tically equal to the voltage over the fluorescent lamp, can reachextremely high values, when HB .

However, when HB , the maximum values ofcan be limited, and the peak value of is obtained

according to

CC

HB

HB

(19)

In addition, from (17), the rms value of can be obtainedwith specific softwares, such as MathCAD or MATLAB. Thisvalue is important because it flows through the lamp filaments,before the establishment of the arc through the gas column. Thewaveforms of are similar to the waveforms of de-picted in Fig. 3. Thus, the difference between HB and

is also responsible for the rms values of , and it canbe used to provide a proper strategy for the preheating of thelamp filaments.

According to Fig. 3, it is possible to establish a satisfactorypreheating process, through a proper switching frequency con-trol during the ignition, considering a strategy to slow down theincrease of until a satisfactory evolution of the tem-perature in the filaments.

Nevertheless, the variation of the switching frequency mustbe wisely chosen, in order to provide enough current for thepreheating, without excessive glow current due to the voltageapplied over the lamp.

Fig. 3. Theoretical waveforms of v (t). (a) !HB(ign)=0.908!Ceq.(b)!HB(ign)=0.952!Ceq, (c)!HB(ign)=!Ceq, (d)!HB(ign)= 1.053!Ceq,(e) !HB(ign)=1.110!Ceq, and (f) !HB(ign)=1.255!Ceq.

B. Analysis of Dimming Operation

During the dimming operation, the ballast equivalent circuitcan be represented according to Fig. 4. From this figure, it ispossible to define

HBHB C

(20)

HB C

HB C(21)

(22)

where

HB HB (23)

HB switching frequency required to process P watts,assuming the phase-shift control strategy [14].

One can define the parameter as

HB (24)

where is defined according to (12).


Fig. 4. Ballast equivalent circuit during dimming operation.

From (20)–(22), the equivalent impedance of the resonantfilter fluorescent lamp set , during the dimming oper-ation, can be expressed by

(25)

where

C (26)

C (27)

HB C (28)C C

C(29)

C HB C C (30)

HB C C (31)

From (25), the magnitude of is obtained through

(32)

For the voltage over the fluorescent lamp, it is possible toconsider that

(33)

From (21), (25), and (33) the following expression can beobtained:

(34)

where

C

C(35)

C(36)

The absolute value of (34) can be interpreted as a relationbetween the rms values of v and , resulting in

(37)

whereis defined according to (2);is the rms value of .

According to Fig. 4, the current through can be expressedas

(38)

From (25) and (38), the following equation can be written:

(39)

where

C

C(40)

C

C(41)

From (39), the rms value of during dimming operationcan be defined by

(42)

In addition, from (39), the phase-shift of the current canbe obtained through

(43)

The value of the phase-shift of , provided by (43), permitsto verify the conditions for the obtaining of ZVS turn-on in thesemiconductor devices employed in the Half-Bridge, as shownin Fig. 5. Analyzing the branch containing , the direction ofthe current (“source-to-drain”) imposes the turn-on of the in-trinsic diode of , at . After a given time interval (de-limited by the phase-shift), the resonance will invert the currentthrough this branch, imposing a “drain-to-source” direction. Atthis moment, when , it is assumed that a ZVS turn-onis performed in , reducing the turn-on losses associated tothis device (MOSFET). In addition, this analysis is valid for ,during the second half of the switching period.

In the classical configuration of the Half-Bridge inverter, ac-cording to Fig. 5, the capacitor C is supposed to act as a voltagesupply during the half-cycle where . Thus, the valueof C can be determined according to constraints imposed tothe maximum value of voltage ripple over its terminals. Fromthe analysis of impedances, in Fig. 4, one can obtain

(44)

where

HB C(45)

From (38), and (44), it can be obtained

(46)


Fig. 5. Topological stages, and main theoretical waveforms of the classicalresonant half-bridge inverter.

Thus, from (25), (45), and (46), it is possible to write

(47)

where

HB C(48)

HB C(49)

is defined according to (40) and is definedaccording to (41).

From (47), it is possible to establish

(50)

In this way, the voltage ripple (peak-to-peak) over C can berepresented by

(51)

The switching frequency during dimming operation can beobtained using (52), derived from (37)

(52)

In (52), and are replaced by their equivalentexpressions [(35) and (36), respectively]. In addition, the param-eter can also be derived from (12) and (24), resultingin

HB C (53)

Thus, (53) is replaced in (29) and (30) ( and C ,respectively). After it, the equation resulting from (52) is ex-panded, leading to

HB HB

HB (54)

where

C C C

C (55)

C C C C C

(56)

C C (57)

From (54), six different mathematical solutions can be ob-tained. However, only two solutions present physical meaning.Furthermore, analyzing these two remaining solutions, only onecan be considered as a proper solution, because of the dim-ming strategy chosen before, where higher switching frequen-cies produce lower output power in the fluorescent lamp. Thus,(58), shown at the bottom of the page, represents the requiredswitching frequency during dimming control, where you have

and defined according to (59) and (60), respec-tively, shown at the bottom of the next page.

From (52), using (35) and (36) without replacing byits equivalent expression, it is possible to write

C(61)

HB (58)


where

HB C C (62)

HB C (63)

CCC HB C C

(64)

CC HB C C (65)

1) is defined according to (1);2) is defined according to (2);3) HB is defined according to (23).Thus, the value of can also be expressed as (66)

shown at the bottom of the page.The value of is obtained for the nominal value of power

processed through the lamp. Thus, from (3) and (53)

HB

C CC C

(67)

In order to demonstrate the improved methodology for the de-sign of SRPL filter in dimmable applications, a complete designprocedure, based on the developed analyses, will be presented.

III. DESIGN PROCEDURE AND EXAMPLE

The improved design procedure will be presented assumingthe following input and output data:

40 W 2.5 W HB 310

HB 40 kHz and ambient temperature 24 C

: The rms value of the fundamental component ofis determined according to the following

expression:

HB (68)

: Using the nominal values in (2), and (68), it is pos-sible to obtain the value of the relation between

and

HB101.44

3100.727

Fig. 6. (a) v for nominal load, as a function of C , for different valuesof C and (b) i for nominal load, as a function of C , assuming C =

180 nF.

In addition, the following parameters must bedetermined:

HB HB HB251 327 41 rad

257.24

: The value of C is adopted assuming that it mustbe at least 10 times higher than C , i.e., C .This statement is established because C is de-signed to suppress the dc component of ,without changing the resonance between andC . The value of C is specified according to itsinfluence on the rms value of the current through

, for nominal load.From (66), assuming , it is possible

to obtain the value of , after the adop-tion of C and C . Then, (42) and (51) can be usedto guide the choice of the values of C and C .

As commented before, the value of C canbe determined through an analysis in the voltageripple in its terminals, according to (51). Fig. 6(a)shows the value of as a functionof C , for different values of C . In this figure,it can be verified that reduced values of C areresponsible for higher values of ,which can result in improper operating points forthe electronic ballast.

Fig. 6(b) shows the value ofas a function of C , for C nF. From thisfigure, it is possible to note that the value of

is directly proportional to C .Thus, the following values are adopted:

C 180 nF and C 6.8 nF

(59)

(60)

C C(66)


: The value of is determined using (66) and (67)

0.769 1.43 mH

: The phase-shift of for nominal load is calcu-lated through (43)

48.29

It is important to notice that the phase-shiftof is responsible for the occurrence ofZVS turn-on processes for the switches of thehalf-bridge inverter. So, if (43) results in positivevalues or is considered too low to ensure the re-quired ZVS turn-on processes, it is necessary torestart the design, changing the value of C or

HB .: After the definition of C and C for the op-

eration at nominal condition, it is necessary to an-alyze the ignition process. From (12)

327,000.54 rad s

The frequency related to can be calculatedusing (12)

52.04 kHz

In (19), admitting that HB (and HB )can assume different values due to the controlstrategy, it is possible to obtain the graph of

depicted in Fig. 7(a). Fig. 7(b) is ob-tained through the rms value of (17), computed inMathCAD. From the results presented in Fig. 7,it is possible to observe that higher values of

HB are responsible for lower values ofand , during the ignition process.

As commented before, the value of HBmust be wisely chosen, because it is necessary toprovide enough current for the preheating whilethe voltage over the lamp is sustained in reducedvalues, in order to avoid the occurrence of exces-sive glow current, and the ignition itself.

As an example of control strategy for a proper ig-nition process, it could be assumed that HB

kHz, from Fig. 7. This frequency should besustained during a given time interval, until thefilaments reach a suitable temperature. After it,

HB should be reduced (e.g., 65 kHz), pro-viding the increase of until the estab-lishment of the arc through the gas column. Then,the control should impose the nominal value ofswitching frequency (in this case, 40 kHz), com-pleting the ignition process.

Some specific ICs, such as IR2159x, are capableto provide a fully programmable ignition process,based on the control of the switching frequency

Fig. 7. (a) v as a function of fHB and (b) i as a functionof fHB .

Fig. 8. Values of jZ (P )j for the input and output data of this design example,during dimming control.

[18]. Additionally, a soft-transition between the ig-nition and the stable operation can be achieved withthis IC, minimizing the risk of failure in the estab-lishment of stable operation.

: Finally, it is possible to evaluate the equivalentimpedance of the resonant filter + fluorescentlamp set, during dimming control. The values of

HB and can be obtained through(58) and (53), respectively. After it, these valuesare employed in (32), generating the graph showedin Fig. 8. According to this figure, it is pos-sible to observe that the value of the magnitudeof is practically constant during the op-erating range specified in this design example2.5 W 40 W . This fact occurs because

the variation of the switching frequency, per-formed to provide dimming control, is responsiblefor changes in the reactances of C , and C .

Based on this result, it can be assumed that the valuesof , defined by (42), will also remain practicallyconstant, even for low values of power processed throughthe fluorescent lamp. In other words, the switching frequencyvariation permits a rearrangement in the flux of reactive energythrough the resonant filter fluorescent lamp set. So, in spite ofthe decrease of active power processed through the set, for lowdimming conditions (higher values of switching frequency),the reactive power is increased, sustaining the total amount of


TABLE IIPARAMETERS USED TO IMPLEMENT THE PROTOTYPES

power in high levels, near to the nominal ones, obtained for40 W. Thus, it is possible to conclude that the conduction

losses in the MOSFETs of the Half-Bridge inverter will bepractically constant during the dimming operation, possiblyresulting in a significant reduction in the global efficiency ofthis structure, when operating in low dimming conditions.

After specifying the values of C and C , it is possibleto analyze the behavior of the Half-Bridge switching frequency,the phase-shift of , and the value of , during thedimming process.

The values of C , and C obtained with this design pro-cedure ensure the proper operation at the nominal point. Con-cerning the analyses of operating points other than the nom-inal one, they can be developed with the equations presented inthis paper. It should be highlighted that, usually, these analysesare not presented in the literature, especially the evaluation of

, which can indicate the stresses of current in the cir-cuit, during dimming operation.

IV. EXPERIMENTAL RESULTS

In order to validate the analysis and the improved design pro-cedure developed in this paper, different laboratorial prototypeswere implemented, according to the parameters presented inTable II.

As commented before, the IR2159x is used to control theHalf-Bridge inverter, providing conditions to the developmentof a proper ignition process, obeying the recommendations ofANSI standards [19].

The experimental results from the implemented prototypesare compared to theoretical results provided by (42), (43), and(58), based on the lamp model described by (1). It is informedthat overloads ( 40 W) were imposed to the prototypes,in order to verify the behavior of the analyzed parameters underthese conditions. In addition, it is informed that the experimentaldata presented in the following figures are average values of sev-eral sets of measurements, obtained for each different operatingpoint.

The voltage and current waveforms from are depicted inFig. 9, considering two different operating points for the Ballast

Fig. 9. Commutation details of S . (a) P '40 W, f = 40 kHz. (b) P '=5 W, f ' 75 kHz. (c) Turn-on (intrinsic diode),P ' 40 W, f '= 40 kHz.(d) Turn-off, P ' 5 W, f = 5 W, f ' 75 kHz. (e) turn-off, P ' 40 W,f ' 40 kHz. (f) Turn-off, P ' 5 W, f ' 75 kHz.

2 ( HB 40 kHz). From Fig. 9(a) and (b), one can observethe changes in the phase-shift because of the dimming operation.Fig. 9(c) and (d) show details of the turn-on processes of theintrinsic diode of . The details related to the turn-off processesof are shown in Fig. 9(e) and (f). Based on Fig. 9(c) until (f), itis possible to verify that hard-switching processes are performedin the turn-on of the intrinsic diode and in the turn-off of .However, it is important to emphasize that the ZVS turn-on ofthe MOSFETs permits the suppression of the major switchinglosses associated to these devices.

Fig. 10 shows the ballasts switching frequencies as a func-tion of power processed through the lamp. The dots arethe experimental results, obtained with a digital oscilloscope(Tektronix—TDS420A), and the solid lines are derived fromthe theoretical analysis provided by (58). According to thisfigure, it is possible to verify that (58) properly representsthe ballast switching frequency required for dimming control,which permits to evaluate the operating points concerned tothe ballasts before implementing prototypes.

The phase-shift of for each different implemented ballastsis depicted in Fig. 11. In this figure, it is possible to note somedifferences between theoretical (solid lines) and experimental


Fig. 10. Ballast switching frequency during dimming operation.(a) f = 30 kHz. (b) f = 40 kHz. (c) f = 50 kHz.(d) f = 60 kHz

Fig. 11. Phase-shift of i during dimming operation. (a) f = 30 kHz.(b) f = 40 kHz. (c) f = 50 kHz. (d) f = 60 kHz.

results (dots). These differences can be explained through thefollowing two comments.

Firstly, due to the low-pass characteristic of the SRPL filter,most of the energy transferred to the fluorescent lamp is pro-cessed through the fundamental component of . In thiscontext, the fundamental approximation technique can be em-ployed for obtaining the equations of the analyzed system. So,the analysis developed with rms values of currents and voltages

Fig. 12. rms values of i during dimming operation. (a) f = 30 kHz.(b) f = 40 kHz. (c) f = 50 kHz. (d) f = 60 kHz.

are very accurate. However, the phase-shift depends on the in-stantaneous values of . Since the instantaneous values of

are different of the instantaneous values of , theexperimental and theoretical waveforms of the current through

are slightly different, resulting in small inaccuracies.Secondly, it is important to inform that the experimental re-

sults concerned to phase-shift were determined using the par-allel cursors of the digital oscilloscope, which means that thesemeasurements are less accurate than those ones related to theballast switching frequency.

Despite these inaccuracies, it is possible to note that the theo-retical values of phase-shift are very similar to the experimentaldata, confirming the validity of this analysis.

Fig. 12 shows the rms values of during dimming opera-tion, for each designed prototype. The solid lines represent the-oretical values and the dots are experimental data. In this figure,it is very important to observe that the values of this parameterremain practically constant during the operating power range(2.5 W 40 W), considering the F40T12 lamp employedin the prototypes.

This fact is very interesting because the rms values ofcurrents processed through the semiconductor devices areproportional to . So, the rms currents through the semi-conductors (MOSFETs), and, consequently, the conductionlosses associated to these devices (proportional to the squarevalue of their rms currents), are also practically constant duringthe dimming operation. It means that these conduction losseswill assume significant values when the ballasts are operatedwith low levels of power processed through the lamps, whichwill obviously reduce the Half-Bridge efficiency. This analysisis usually neglected in the design methodologies presented inthe literature. Moreover, it is important to emphasize that, inlow dimming conditions, the power supplied to the filamentswill assume significant values when compared to the power


Fig. 13. Experimental results of efficiency for the ballast designed to processnominal load at 40 kHz, during dimming operation.

processed in the gas column (arc). So, it is expected that thelow dimming operation will result in low efficiency.

In order to confirm these statements, Fig. 13 shows exper-imental results of efficiency, measured in the ballast designedfor nominal load at 40 kHz. It is important to observe that theseresults are only concerned to the ballast output stage circuitry,i.e., half-bridge inverter resonant filter fluorescent lamp. Inorder to perform the required measurements, the input rectifierwas replaced by a dc regulated voltage supply.

From Fig. 13, the decrease in the ballast efficiency for lowdimming conditions can be clearly observed. In addition to theconduction losses in the MOSFETs, it is necessary to observethat the higher switching frequency, imposed for low dimmingoperation, is responsible for increasing the reactance of ,and decreasing the reactance of C . Thus, the losses associatedto and to the intrinsic series resistance of C are also in-creased when compared to losses verified in the nominal con-dition ( 40 W). The operation in low dimming conditions(high switching frequencies) can also increase the “skin effect”losses, when compared to the nominal operating point, reducingeven more the efficiency of this circuit. Finally, for low dim-ming operation, the power provided to the lamp filaments willbe proportionally high, when compared to the power processedthrough the gas column.

The losses associated to the low dimming operating pointcan be minimized by the employment of MOSFETs with low

(drain-to-source resistance, during conduction), andlitz wire for the winding of . However, before implementingthese modifications, their impact on the costs associated to thiscircuit should be evaluated by the manufacturers, in order tosustain its attractiveness for the market.

V. CONCLUSION

This paper presented an improved design procedure for spec-ifying the parameters of the classical SRPL filter employed indimmable electronic ballasts.

The theoretical analyses presented in this paper are based onequivalent circuits, and on the fundamental approximation of theinput voltage of inverter stage, like most of analyses presentedin literature. These analyses permit to evaluate the behavior of

some important parameters during the dimming operation, suchas: ballast switching frequency, phase-shift and rms value of thecurrent through the resonant inductor.

It is important to inform that the analyses of the last twoparameters cited above are usually neglected in the designmethodologies presented in literature. However, the predictionof the phase-shift can be used to define a sophisticated dimmingcontrol strategy, according to [14], and it should be evaluated.Moreover, the rms value of current through the resonant in-ductor can indicate the level of current stresses through thesemiconductor devices, and through the components of theresonant filter, providing conditions to develop a qualitativeevaluation of the ballast efficiency.

The accuracy verified in the comparison among theoret-ical and experimental results validates the proposed designprocedure.

Finally, the experimental results for the efficiency of the bal-last output stage were measured, during dimming operation.From the obtained results, it can be concluded that, from thepoint of view of energy saving, low dimming operating pointsare not always recommended, due to the reduced levels of effi-ciency verified in these conditions.

REFERENCES

[1] R. L. Steigerwald, “A comparison of half-bridge resonant convertertopologies,” IEEE Trans. Power Electron., vol. 3, no. 2, pp. 174–182,Oct. 1988.

[2] M. K. Kazimierczuk and W. Szaraniec, “Electronic ballast for fluores-cent lamps,” IEEE Trans. Power Electron., vol. 8, no. 4, pp. 386–395,Oct. 1993.

[3] M. C. Cosby, Jr. and R. M. Nelms, “A resonant inverter for electronicballast applications,” IEEE Trans. Ind. Electron., vol. 41, no. 4, pp.418–425, Aug. 1994.

[4] C. S. Moo, Y. C. Chuang, Y. H. Huang, and H. N. Chen, “Modelingof fluorescent lamps for dimmable electronic ballasts,” in Proc. IEEEIAS’96 Annu. Meeting, 1996, pp. 2231–2236.

[5] Z. Li, P. K. T. Mok, W. H. Ki, and J. K. O. Sin, “A simple method todesign resonant circuits of electronic ballast for fluorescent lamps,” inProc. IEEE-ISCAS’97, 1997, pp. 1744–1747.

[6] C. S. Moo, H. L. Cheng, H. N. Chen, and H. C. Yen, “Designing adimmable electronic ballast with frequency control,” in Proc. IEEEAPEC’99, 1999, pp. 727–733.

[7] C. S. Moo, H. L. Cheng, H. N. Chen, and H. C. Yen, “Designing adimmable electronic ballast with frequency control,” in Proc. IEEEAPEC’99, 1999, pp. 727–733.

[8] R. N. Prado, A. R. Seidel, F. E. Bisogno, and M. A. D. Costa, “A de-sign method for electronic ballast for fluorescent lamps,” in Proc. IEEEIECON’00, 2000, pp. 2279–2284.

[9] T. J. Liang, C. A. Cheng, W. B. Shyu, and J. F. Chen, “Design procedurefor resonant components of fluorescent lamps electronic ballast basedon lamp model,” in Proc. IEEE PEDS’01, 2001, pp. 618–622.

[10] C. A. Cheng, T. J. Liang, C. M. Chuang, and J. F. Chen, “A novel methodof using second-order lamp model to design dimmable fluorescent lampselectronic ballast,” in Proc. IEEE IECON’01, 2001, pp. 1033–1037.

[11] Y. K. E. Ho, S. T. S. Lee, H. S. H. Chung, and S. Y. Hui, “A comparativestudy on dimming control methods for electronic ballasts,” IEEE Trans.Power Electron., vol. 16, no. 6, pp. 828–836, Nov. 2001.

[12] T. J. Ribarich and J. J. Ribarich, “A new procedure for high-frequencyelectronic ballast design,” IEEE Trans. Ind. Applicat., vol. 37, no. 1, pp.262–267, Jan./Feb. 2001.

[13] F. E. Bisogno, A. R. Seidel, R. Holsbach, and R. N. Prado, “Resonantfilter applications in electronic ballast,” in Proc. IEEE IAS’02 Annu.Meeting, 2002, pp. 348–354.

[14] J. Adams, T. J. Ribarich, and J. J. Ribarich, “A new IC for dimmablehigh-frequency electronic ballasts,” in Proc. IEEE APEC’99, 1999, pp.713–719.

[15] C. S. Moo, Y. C. Hsieh, H. C. Yen, and C. R. Lee, “Fluorescent lampmodel with power and temperature dependence for high-frequency elec-tronic ballasts,” IEEE Trans. Ind. Applicat., vol. 39, no. 1, pp. 121–127,Jan./Feb. 2003.


[16] F. T. Wakabayashi and C. A. Canesin, “A new model for tubular fluo-rescent lamps operated at high frequencies for dimmable applications,”in Proc. IEEE ISIE’03, 2003. CD-ROM.

[17] F. T. Wakabayashi and e C. A. Canesin, “An improved design procedurefor LCC resonant filter of dimmable electronic ballasts for fluorescentlamps, based on lamp model,” in Proc. IEEE IECON’03, Nov. 2003. tobe published.

[18] Data Sheet n PD60169-C, International Rectifier. [Online]. Available:http://www.irf.com

[19] Y. Ji, R. Davis, C. O’Rourke e, and E. W. M. Chui, “Compatibility testingof fluorescent lamp and ballast systems,” IEEE Trans. Ind. Applicat., vol.35, no. 6, pp. 1271–1276, Nov./Dec. 1999.

Fabio Toshiaki Wakabayashi was born in Jales(SP), Brazil, in 1974. He received the B.S., M.S.,and Ph.D. degrees in electrical engineering from SãoPaulo State University, Ilha Solteira, in 1996, 1998,and 2003, respectively.

Currently, he is an Associate Researcher of thePower Electronics Laboratory, UNESP, Ilha Solteira,participating in a R&D project concerning powerquality issues. His interests include electronicballasts for fluorescent lamps, dimming control,power-factor-correction techniques, soft-switching

techniques, dc-to-dc converters, and switching-mode power supplies.

Carlos Alberto Canesin (S’87–M’97) receivedthe B.S. degree from Paulista State University, IlhaSolteira (SP), Brazil, in 1984 and the M.S. andPh.D. degrees from the Federal University of SantaCatarina, Florianópolis (SC), Brazil, in 1990 and1996, respectively, all in electrical engineering.

He started the Power Electronics Labora-tory—LEP, UNESP, São Paulo State University,Ilha, Brazil, where he is currently an AssociateProfessor. His interests include soft-switching tech-niques, dc-to-dc converters, switching-mode power

supplies, solar/photovoltaic energy applications, electronic fluorescent ballasts,and active power-factor correction techniques.

Dr. Canesin is an Associate Editor for the IEEE TRANSACTIONS ON POWER

ELECTRONICS.

Documents

An improved design procedure for LCC resonant filter of dimmable electronic ballasts for fluorescent lamps, based on lamp model