An electrically small antenna with integrated DGS filter

second stopband can be kept constant when the first stopband

changes by tuning l3 and l6 (as shown in Figure 6), while the

first stopband is constant when the second stopband changes by

tuning l4 and l5 (as shown in Figure 7). Obviously, both stop-

bands can be controlled independently. The open-circuited stub

shunted in the middle of the filter can improve the impedance

matching in the passband. Figure 8 shows the simulated S11

with and without the open-circuited stub. It can be seen that the

return loss of the upper passband is greatly improved by adding

the open-circuited stub.

3. SIMULATION AND EXPERIMENTAL RESULTS

According to the above analysis, the proposed dual-bandstop fil-

ter is designed, simulated, and optimized using electromagnetic

simulation tool IE3D. Note that the shorted stubs and the T-

shaped microstrip line are bended in the final fabrication for fur-

ther size reduction. The final dimensions of the dual-bandstop

filter are as follows: L ¼ 16.2 mm, L1 ¼ 3.6 mm, L2 ¼ 3.0 mm,

L3 ¼ 10.1 mm, L4 ¼ 6.4 mm, L5 ¼ 6.6 mm, L6 ¼ 10.5 mm,

L7 ¼ 3.9 mm, L8 ¼ 0.2 mm, W ¼ 0.8 mm, W1 ¼ 0.15 mm, W2

¼ 1 mm, W3 ¼ 0.2 mm, W4 ¼ 0.9 mm, s ¼ 0.1 mm, and d ¼0.2 mm. Using a print-circuit-board (PCB) technique, the filter

was fabricated on a microstrip substrate with a relative dielectric

constant of 3.5 and thickness of 0.508 mm. The fabricated dual-

bandstop filter is shown in Figure 7. Although a low dielectric

constant has been used, the filter is extremely compact with a

size of 11.2 mm � 8.05 mm. Figure 8 demonstrates the S-pa-

rameters of the dual-bandstop filter. It can be seen that the simu-

lated and measured results are in reasonable agreement with

each other. As can be observed from Figure 9, two stopbands

centered at 2.95 GHz and 4.05 GHz are realized. The first stop-

band has a measured maximum attenuation of 36 dB at 2.99

GHz, while the second stopband has a largest attenuation of 31

dB at 4.02 GHz. Furthermore, the insertion loss in the lower

passband from DC to 2.5 GHz is less than 0.5 dB, while the

insertion losses in the middle passband (from 3.25 GHz to 3.67

GHz) and upper passband (from 4.37 GHz to 6.67 GHz) are all

less than 1 dB.

Table 2 gives a comparison of some typical dual-bandstop

filters with the proposed design. Compared with the dual-band-

stop filter in [12], which has a similar performance with the pro-

posed work, the latter is only 13% of their overall size.

5. CONCLUSION

A novel compact dual-bandstop filter based on CRLH-TL is pre-

sented. The related circuit model has also been developed,

which can be used to design the desired dual-bandstop filter.

Each of the central frequencies of the stopbands can be con-

trolled independently. A compact dual-bandstop filter is

designed, fabricated, and tested. The simulated and measured

results are found in good agreement with each other.

ACKNOWLEDGMENTS

This work is supported by grants from the National Natural Science

Foundation of China-NASF (Grant No: 10976005), the Program

for New Century Excellent Talents in University (Grant No:

NCET-11-0066), and the Research Fund of Shanghai Academy of

Spaceflight Technology.

REFERENCES

1. F.-C. Chen and Q.-X. Chu, Design of dual-band CT filter with

sourceload coupling, J Electromagn Waves Appl 25 (2011), 15–22.

2. K. Song and Y. Fan, Compact ultra-wideband bandpass filter using

dual-line coupling structure, IEEE Microwave Wireless Compon

Lett 19 (2009), 30–32.

3. K. Song and Q. Xue, Novel broadband bandpass filters using Y-

shaped dual-mode microstrip resonators, IEEE Microwave Wireless

Compon Lett 19 (2009), 548–550.

4. S.-C. Lin and C.-Y. Yeh, Stopband-extended balanced filters using

both k/4 and k/2 SIRS with common-mode suppression and

improved passband selectivity, Progr Electromagn Res 128 (2012),

215–228.

5. K. Song and Q. Xue, Inductance-loaded Y-shaped resonators and

their applications to filters, IEEE Trans Microwave Theor Tech 58

(2010), 978–984.

6. J.-Q. Huang, Q.-X. Chu, and C.-Y. Liu, Compact UWB filter based

on surface-coupled structure with dual notched bands, Progr Elec-

tromagn Res 106 (2010), 311–319.

7. K. Song and Q. Xue, Compact ultra-wideband (UWB) bandpass fil-

ters with multiple notched bands, IEEE Microwave Wireless Com-

pon Lett 20 (2010), 447–449.

8. S. Barbarino and F. Consoli, UWB circular slot antenna provided

with an inverted-L notch filter for the 5 GHz WLAN band, Progr

Electromagn Res 70 (2007), 297–306.

9. K. Chin, J. Yeh, and S. Chao, Compact dual-band bandstop filters

using stepped-impedance resonators, IEEE Microwave Wireless

Compon Lett 17 (2007), 849–854.

10. S. Han, X. Wang, and Y. Fan, Analysis and design of multiple-

band bandstop filters, Progr Electromagn Res 70 (2007), 297–306.

11. H.-K. Chiou and C.-F. Tai, Dual-band microstrip bandstop filter

using dual-mode loop resonator, Electron Lett 43 (2007), 507–509.

12. X.-Y. Zhang, X. Huang, H.-L. Zhang, and B.-J. Hu, Novel dual-

band bandstop filter with controllable stopband frequencies, Micro-

wave Opt Technol Lett 54 (2012), 1203–1206.

13. G. Monti, R. De Paolis, and L. Tarricone, Design of a 3-state

reconfigurable CRLH transmission line based on mems switches,

Progr Electromagn Res 95 (2009), 283–297.

14. J.L. Jimenez Martn, V. Gonzalez-Posadas, J.E. Gonzalez-Garca,

F.J. Arques-Orobon, L.E. Garcia-Munoz, and D. Segovia-Vargas,

Dual band high efficiency class CE power amplifier based on

CRLH diplexer, Progr Electromagn Res 97 (2009), 217–240.

15. X.Q. Lin, P. Su, Y. Fan, and Z.B. Zhu, Improved CRLH-TL with

arbitrary characteristic impedance and its application in hybrid ring

design, Progr Electromagn Res 124 (2012), 249–263.

16. T. Yang, P.L. Chi, and T. Itoh, Compact quarter-wave resonator

and its applications to miniaturized diplexer and triplexer, IEEE

Trans Microwave Theor Tech 59 (2011), 260–269.

17. C. Caloz and T. Ihoh, Electromagnectic metamaterials, transmis-

sion line theory and microwave applications, Wiley, New York,

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VC 2013 Wiley Periodicals, Inc.

AN ELECTRICALLY SMALL ANTENNAWITH INTEGRATED DGS FILTER

Sofiane Oukil,1 Hocine Kimouche,1 and Arab Azrar21Microwaves/Radar Laboratory, Ecole Militaire Polytechnique, BP17 Bordj El-Bahri Algiers, Algeria2 Institute of Electrical and Electronic Engineering, University ofBoumerdes, Boumerdes, Algeria; Corresponding author:[email protected]

Received 24 August 2012

ABSTRACT: In this letter, a dumbbell-shaped defected groundstructure is proposed to suppress higher order harmonics of an

electrically small antenna. A slot loading technique is applied to a well-known low-profile antenna, and the area miniaturization of a

962 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 5, May 2013 DOI 10.1002/mop

rectangular microstrip patch antenna by inserting a number of slitsparallel to the radiating edges is investigated in relation to the quality

factor. The antenna was designed to operate at 0.95 GHz (GSM).Measurement results on a prototype antenna fully demonstrate theperformance of the proposed antenna. VC 2013 Wiley Periodicals, Inc.

Microwave Opt Technol Lett 55:962–966, 2013; View this article online

at wileyonlinelibrary.com. DOI 10.1002/mop.27498

Key words: DGS; small antenna; integrated DGS filter; electrically

small antenna

1. INTRODUCTION

Electrically small antennas, i.e., antennas that are small com-

pared to the free-space wavelength at their frequency of opera-

tion, have been a subject of research for many years [1, 2]. And

the challenges of these antennas are well known. They are uti-

lized in many applications, from mobile phones to GPS

receivers and wireless computer links. Moreover, the advance in

the electronic technology leads to an extensive decrease in the

overall size of the electronic parts needed for such mobile appli-

cations. Based on this fact, there is a demand in modern com-

munication systems for smaller antennas, with size significantly

less than the usual half wavelength.

In this letter, we present a microstrip small antenna inte-

grated with defected ground structure (DGS) jointly in ground

plane. The area miniaturization is achieved due to the presence

of the slits. The slits force the surface currents to meander, thus

artificially increasing the antenna’s electrical length without

modifying the patch’s global dimensions. A significant decrease

in the resonant frequency is observed, depending on the slit’s

length and width. The resultant antennas can be characterized as

small antennas in accordance to the relevant fundamental

limitations.

A new technique to reduce the higher order harmonics is

proposed using a dumbbell-shaped DGS etched in the ground

plane is presented and compared with a small antenna without

the DGS unit cell. It is well known that the DGS unit cell acts

as the stop-band filter at some frequency bands.

2. FUNDAMENTAL LIMITS AND COMPUTATION OF THEQUALITY FACTOR Q

The fundamental limitations of small antennas were first

addressed by Wheeler [1] and Chu [3]. It has been shown that

when an antenna becomes electrically small, its bandwidth

decreases. Chu derived an expression relating the antenna’s radi-

ation quality factor Q with the space the antenna fills, called

the radian sphere where the radius of this radian sphere is the

largest linear antenna dimension. Chu established a fundamental

Q limitation, which later reexamined by McLean [4], and is

given by:

Q ¼ 1

Kaþ 1

ðKaÞ3(1)

where K ¼ 2pk and k is the operating wavelength and ‘‘a’’ is the

radius of the radian sphere.

The above expression establishes a fundamental limit which

means that no antenna can ever exceed this threshold. If the

antenna presents losses, meaning the radiation efficiency is

lower than unity, then the quality factor is given as Q ¼ grQr.

The more efficiently an antenna occupies its radian sphere the

closer its Q would be to the Chu limit. An antenna is said to be

a miniature if the radius a <k/2p or alternatively if Ka < 1.

An approximate expression for the computation of the quality

factor of an antenna is presented by Yaghjian and Best in [5, 6],

and it is given by:

QzðxÞ ¼x

2RðxÞ

ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi½R0ðxÞ�2 þ X0ðxÞ þ jXðxÞj

x

� �2s

(2)

where R(x) and X(x) are the resistance and reactance of the

antenna. This expression is used to calculate the quality factor

of the presented antenna configurations and it is compared with

the Chu limitation and the miniature antenna limit.

3. ANTENNA STRUCTURE AND DESIGN

The configuration of the proposed electrically small antenna is

illustrated in Figure 1. A low-cost Epoxy glass substrate with

relative permittivity of 4.32 and thickness of 1.6 mm is used. To

achieve a good match to a 50 X line, the width WL of microstrip

feed-line was set as 3.1 mm. The original patch that was used is

a rectangular microstrip patch antenna of resonant length L ¼36 mm and width W ¼ 44 mm. The slits are placed in parallel

with the radiating edges as shown in Figure 1.

Figure 1 The layout of the resultant small antenna. [Color figure

can be viewed in the online issue, which is available at

wileyonlinelibrary.com]

Figure 2 Input reflection coefficient for various values of Ls. [Color

figure can be viewed in the online issue, which is available at


DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 5, May 2013 963

The slit width (Ws) was set equal to 1 mm. Simulation for

various lengths, Ls, as shown in Figure 2, shows that the reso-

nant frequency decreases with increasing the length Ls of the

antenna.

The quality factor Q for each one of the aforementioned slit-

ted configurations was calculated using Eq. (2) and is given in

Table 1. In Figure 3, the typical simulated Q curve for a specific

slit length shows that the minimal value for Q is at resonant

frequency.

The marker corresponds to the resonant frequency and the

relevant value of the quality factor Q. As can be seen for Ls >20 mm, all the demonstrated cases, the ratio of the antenna max-

imum dimension to the resonant wavelength is such that the

product Ka is below the Chu limit concerning the small antenna

definition (Ka < 1). Thus, the designed antennas can be labeled

as ‘‘small antennas’’.

4. SIMULATION AND EXPERIMENTAL RESULTS

Simulations are performed using commercially available pack-

age IE3D from Zeland. The measurement of the return

loss was performed using an Agilent 8719ES Vector Network

Analyzer. A photograph of the realized antenna is shown in

Figure 4.

The slit length (Ls) was set equal to 32 mm. The obtained

measured and simulated reflection coefficients are shown in Fig-

ure 5. According to this figure, it may be noticed that a good

agreement between simulated and measured resonant frequencies

is achieved, except the shift location of the resonant frequency.

This difference is probably caused by fabrication tolerances,

because of the uncertainty in the thickness and/or the dielectric

constant of substrate.

Obviously, the bandwidth achieved is sufficient for the Mo-

bile applications GSM (0.89–0.96 GHz). The currents on the

surface of the patch at 0.95 GHz are also presented in Figure 6.

It can be seen that the slits force the surface currents to mean-

der, thus artificially increasing the antenna’s electrical length

without modifying the patch’s global dimensions.

Figure 7 shows both the simulated and measured input reflec-

tion coefficient for the antenna without DGS for frequencies

from 0.8 GHz to 4.5 GHz. Clearly, the two figures are in good

agreement.

TABLE 1 Performance of the Slitted Configured Antenna

Ls (mm) F0 (GHz) Q QChu Ka

34 0.91 34.77 3.94 0.73

32 0.95 32.74 3.59 0.76

28 1.05 31.70 2.87 0.84

24 1.17 25.27 2.26 0.94

20 1.32 15.82 1.78 1.06

18 1.39 12.28 1.60 1.12

Figure 3 The simulated Q curve for the 32 mm slit length case.

[Color figure can be viewed in the online issue, which is available at


Figure 4 Photograph of the realized antenna. [Color figure can be

viewed in the online issue, which is available at wileyonlinelibrary.com]

Figure 5 Measured and simulated return loss of the small antenna.



Figure 6 Current distributions on the antenna surface at 0.95 GHz.




5. SMALL ANTENNA WITH DGS FILTER INTEGRATED

The proposed stop-band harmonic suppression antenna configu-

ration is shown in Figure 8. The creation of a dumbbell-shaped

DGS [7] in the ground plane of the antenna is used for the sup-

pression of undesired frequencies. To study the effect of the

integrated filter on the antenna performances, the following sim-

ulations and measurements are performed.

It has been remarked that the reflection coefficient of the

DGS patch antenna depends on the dimension of the DGS unit

cell. An optimal values has been obtained by choosing a ¼ h ¼4 mm and d ¼ 16 mm. The fabricated antenna is depicted in the

photograph shown in Figure 9.

The simulated and measured input reflection coefficients of

the small patch antenna and small patch antenna with the DGS

unit cell are shown in Figures 10(a) and 10(b). The measured

results are in good accordance with the simulation ones.

It is clear from the results that the resonant frequency

(fundamental) of the proposed small antenna is 0.95 GHz.

The insertion of the filter leads to suppressing the higher

Figure 8 Small antennas with defect in the ground plane. [Color

figure can be viewed in the online issue, which is available at


Figure 9 Photograph of the proposed antenna: (a) front side and (b)

back side. [Color figure can be viewed in the online issue, which is

available at wileyonlinelibrary.com]

Figure 7 Measured and simulated return loss of the small antenna for

0.8–4.5 GHz. [Color figure can be viewed in the online issue, which is

available at wileyonlinelibrary.com]

Figure 10 (a) Simulated return loss of the small antenna and small

antenna with the DGS. (b) Simulated and measured return loss of the

small antenna with the DGS. [Color figure can be viewed in the online

issue, which is available at wileyonlinelibrary.com]

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 5, May 2013 965

order harmonics that correspond to 1.4 GHz, 1.7 GHz, 3.2

GHz, etc.

6. CONCLUSION

A basic rectangular microstrip patch antenna can be miniatur-

ized by inserting a number of slits parallel to the radiating

edges. A frequency reduction of up to 90% and an area reduc-

tion up to 55% can be observed, depending on the length of the

slit. For all the above cases, the resultant antennas can be char-

acterized as ‘‘small antennas’’ since the Chu and Wheeler limit

is satisfied.

A stop-band harmonic suppression microstrip small antenna

based on the dumbbell-shaped DGS has been presented. The

compact antenna with ultra-low harmonics is obtained by insert-

ing the filter, and it has potential application in the future in

next-generation mobile communication system.

REFERENCES

1. H.A. Wheeler, Fundamental limitations of small antennas, Proc

IRE 35 (1947), 1479–1484.

2. H.A. Wheeler, Small antennas, IEEE Trans Antennas Propag 23

(1975), 462–469.

3. L.J. Chu, Physical limitations of omni-directional antennas, J Appl

Phys 19 (1948), 1163–1175.

4. J.S. McLean, A re-examination of the fundamental limits on the

radiation Q of electrically small antennas, IEEE Trans Antennas

Propag 44 (1996), 672–676.

5. A.D. Yaghjian and S.R. Best, Impendence, bandwidth, and Q

of antennas, IEEE Trans Antennas Propag 53 (2005), 1298–

1324.

6. S.R. Best, The inverse relationship between quality factor and

bandwidth in multiple resonant antennas, IEEE Antennas Propag

Soc Int Symp (2006), 623–626.

7. D. Ahn and J.S. Park, A design of the low-pass filter using the

novel microstrip defected ground structure, IEEE Trans Microwave

Theory Tech 49 (2001), 86–93.

VC 2013 Wiley Periodicals, Inc.

PLANAR MONOPOLE ANTENNA FORMULTIBAND WIRELESS APPLICATIONS

Seyed Ehsan Hosseini,1,2 Amir Reza Attari,1

and Mohammad Saeed Majedi1,21 Electrical Engineering Department, Ferdowsi University ofMashhad, Mashhad, Iran2Communications and Computer Research Center, FerdowsiUniversity of Mashhad, Mashhad, Iran; Corresponding author:[email protected]

Received 26 August 2012

ABSTRACT: In this article, a new multiband planar monopoleantenna for mobile devices application is presented. The proposed

antenna operation is on the basis of using two radiation spiral stripsand two slots on the ground plane. This antenna covers eight standard

frequency bands including: GSM900, DCS1800, PCS1900, UMTS,WiBro, Bluetooth, WiMax, and WLAN. It occupies the area of 40 mm� 18 mm excluding the ground plane. A prototype of this antenna has

been fabricated and its parameters have been measured. There is agood agreement between the measured and simulated results. VC 2013

Wiley Periodicals, Inc. Microwave Opt Technol Lett 55:966–970,

2013; View this article online at wileyonlinelibrary.com. DOI 10.1002/

mop.27497

Key words: monopole antennas; multiband antennas; planar antennas

1. INTRODUCTION

With the progress of wireless communication systems which

operate in multi-frequency bands, there is a growing demand

for multiband antennas and microwave devices. For mobile

phones, planar antennas are very attractive, because these

antennas have low profile and also can be simply fabricated on

the system circuit board of the mobile phone for practical

applications.

In recent years, many planar antennas with multiband capa-

bility are suggested, including multiband planar inverted-F

antennas (PIFA) [1, 2] and planar monopole antennas [3–7]. The

proposed antenna in Ref. [3] covers four standard frequency

bands using a planar rectangular patch and a folded slit at the

patch’s bottom edge. In Ref. [4], a T-shaped planar monopole

antenna for covering five frequency bands is proposed. In Ref.

[5], a dual band monopole antenna is presented which is com-

posed of two inverted-L branches. A planar dual U-shaped

monopole antenna with multiband operation for the WiMax sys-

tem is proposed in Ref. [6]. Finally, a new monopole antenna

for penta-band WWAN operation in the mobile phone using

chip inductor is presented in Ref. [7].

In this article, a planar monopole antenna with two spiral

strips and two slots in the ground plane is proposed. Similar to

the presented antennas in Refs. [8] and [9] using slots in the

ground plane leads to the improvement of impedance matching.

The presented antenna can cover eight standard frequency

bands including the global system for mobile communication

(GSM900, 880–960 MHz), digital communication system

(DCS, 1710–1880 MHz), personal communication system

(PCS, 1850–1990 MHz), universal mobile telecommunication

system (UMTS, 1920–2170 MHz), wireless broadband (WiBro,

2300–2390 MHz), Bluetooth (2400–2484 MHz), worldwide

interoperability for microwave access (WiMax, 3400–3600

MHz), and wireless local area network (WLAN, 5150–5350

MHz, 5725–5825 MHz). In contrast to the planar inverted-F

antennas [1, 2], the proposed antenna has low profile. In addi-

tion, it covers more frequency bands in comparison with the

antennas in Refs. [3–7].

2. ANTENNA STRUCTURE

Figure 1 illustrates the geometry with detailed dimensions of

the proposed multiband planar monopole antenna which has

two spiral strips and two slots in the ground plane. A 1.6-mm

thick FR4 dielectric with size of 40 � 110 mm2, relative per-

mittivity of 4.4 and loss tangent of 0.0245 is used as the sub-

strate of the antenna. This structure is fed by a 50 X microstrip

line which is printed on the circuit board and connected

through a via-hole to a 50 X SMA connector on the back side

of the circuit board.

As it will be shown in the next section, two spiral strips

are the radiation elements which excite proper resonance

modes for achieving desired frequency bands. Using slots on

the ground plane also leads to improvement of the input

matching of the antenna in some considered frequency

bands.

3. RESULTS AND DISCUSSION

The proposed antenna in the previous section was fabricated.

Figure 2 shows the photograph of a prototype. Figure 3 shows

the simulated and measured reflection coefficients versus the fre-

quency for this antenna. HFSS [10] has been used to analyze

and optimize the output characteristics of the antenna, and

E8364 network analyzer is used to measure the reflection


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An electrically small antenna with integrated DGS filter