A Very Low-Profile Dual Output LLC Resonant Converter for LCD/LED TV Applications

3514 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 7, JULY 2014

A Very Low-Profile Dual Output LLC ResonantConverter for LCD/LED TV Applications

Ibrahim Demirel and Burcu Erkmen, Member, IEEE

Abstract—The concept of low-profile power supply becomes acurrent issue because of the recent trends in LCD/LED TV con-sumer market. Realizing a low-profile power supply with highpower density is a challenging work due to the size limitation of pas-sive components. In this paper, a very low-profile dual output LLCresonant converter proposed by considering the criteria of 10 mmcomponent height limit without any drop-in cuts on the surfaceof printed circuit board. The simulation model of the converter isapplied to the SPICE tool in order to estimate the performanceand borders of the passive components. On the basis of theoreticalapproach and simulation results, two low-profile transformers areutilized in order to acquire a prototype with 200 W output power(24 V/6 A and 12 V/4.5 A). LLC series resonant converter proto-type with the dimension of 140 mm × 291 mm × 10 mm achieves89.4% efficiency at nominal load conditions. The input voltagerange is between 330 to 400 V and the nominal input voltage isspecified as 400 V. Operating frequencies are obtained as 65 and81 KHz at 330 and 400 V, respectively, under the nominal load con-ditions. Thermal and electromagnetic interference performancesof the proposed converter are included in the paper to validateexperimental study.

Index Terms—Dual Output, leakage reduction method, LLC res-onant converter, low-profile power supply.

NOMENCLATURE

LR Resonant inductance of a transformer.LM Magnetizing inductance of a transformer.CR Resonant capacitor.D Output rectifier diode.CO Output capacitor.RL Output load resistance.n Turn ratio of a transformer.n′ Equivalent turn ratio of transformers.NP The number of turns on the primary side.NS The number of turns on the Secondary side.Vin RMS value of the input voltage.VDC Maximum value of the input voltage.Vout Output voltage.Vo Fundamental RMS output voltage.Iout Output current.

Manuscript received April 17, 2013; revised July 24, 2013; accepted August5, 2013. Date of current version February 18, 2014. This work was supportedby the electronic division of Arcelik Company Ltd. of Turkey under GrantAR20D11T04. Recommended for publication by Associate Editor M. Ponce-Silva.

The authors are with the Department of Electronics and CommunicationEngineering, Yildiz Technical University, 34330 Istanbul, Turkey (e-mail:[email protected]; [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TPEL.2013.2278715

Io Fundamental RMS output current.RAC Equivalent ac resistance.RE Total ac resistance.MAC Voltage gain of the converter.fn Normalized frequency.m Inductance ratio: LM /LR.QE Quality factor.M Minimum gain value.VF Forward voltage drop of rectifiers.Vgs Gate–source voltage of switching MOSFETs.ILM Magnetizing current of primary side.VC R Voltage across the resonance capacitor.Vd Output voltage of secondary side rectifiers.ID Output current of secondary side rectifiers.

I. INTRODUCTION

R ECENTLY, slimming trend in consumer electronicsemerged the necessity of power supplies to go on a di-

rection in the same way to follow these product trends. Techni-cal specifications such as low profile, high power density, highefficiency, thermal reliability, etc., make power supplies diffi-cult to design. To overcome these difficulties, an LLC seriesresonant converter is widely preferred among other topologiesbecause of the features that offered to the designer. In this topol-ogy, soft switching methods [zero-voltage switching (ZVS) andzero-current switching (ZCS)] used at main switches, reduceswitching losses at high frequencies, provide high efficiency,and eliminate reverse recovery problem of the output rectifica-tion diodes [1]–[5]. In [6], the necessary dead time and maxi-mum switching frequency are investigated for realizing the ZVSoperation of the LLC resonant converter. A LLC converter withZVS operation of the primary switches and ZCS operation ofthe rectifier diodes is introduced in [7]. Furthermore, controlease of output regulation over a wide load range and cost effec-tive approach for high power density applications are the otherimportant features of LLC resonant converters [4]–[15]. On theother hand, while designing a very low-profile LLC resonantconverter with high power and dual output, slimming down theheight of the transformer and optimizing the cross regulationare very difficult matters to compete with [13], [14], [16]. Inthis issue, using smaller size transformers make the thermalproblems become apparent. In addition, less winding area andmore output power request are the main handicaps for thesetransformers. Moreover, limitations of the core material of thetransformers restrain using a single transformer in order to get ahigh-output power. The output regulation of LLC series res-onant topology becomes harder, especially when the output

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DEMIREL AND ERKMEN: VERY LOW-PROFILE DUAL OUTPUT LLC RESONANT CONVERTER FOR LCD/LED TV APPLICATIONS 3515

numbers increase, because of the magnetic coupling betweenwindings [13], [14]. Abovementioned limits of the transformersrestrict perfect cross-regulation. In prior literatures, in order tosolve this problem, two transformers are connected in differentways to derive desired output value [8]–[10]. But the heightlimit has not been too influential for these approaches.

In LLC resonant topology, caused by the inequality of sec-ondary windings of the transformers, current imbalance prob-lems occur. In each switching cycle, series resonant frequencychanges and results in transition of secondary leakage induc-tance to the primary. For this reason, total resonant inductancevalue increases and this increase can cause rectifying currentimbalance and rectifier stress. In [17], [18], by changing theconnection method of transformers windings, some resonantinductance reduction methods have been proposed for single-output LLC modules. For multiple outputs, leakage inductancebalance becomes significant. With the connection of two trans-formers series in primary side and parallel in secondary side,leakage inductance balance can be compensated by swappingsecondary windings optimally [19], [20].

Further decreased demands on the height level, have revealeda printed circuit board (PCB) cutting method [21]. Passive com-ponents such as transformers or electrolytic capacitors and thecomponents that need a heat-sink to meet thermal specificationssuch as main switches and output rectification diodes, cover alarge volume upon the PCB. In order to achieve a low-profilemodule, the heights of these components need to be reduced. InPCB cutting method, the placement areas of abovementionedcomponents are cut and the components dropped-in the holesthat formed via PCB cuts. By the aid of these holes on PCB, theheight limit on top of the PCB can be reduced [21]. However,by the reason of the cuts on PCB, usable PCB space decreases.This space reduction affects the layout of the components andemerges the necessity of placing the components closer. For thisreason, the thermal and EMI performance of these power mod-ules diminish. To overcome these problems, the surface area ofthe PCB is enlarged and extra components are added to managereliability standards. However, the enlargement of the PCB de-creases the power density and the cost of the converter increaseswith the extra component usage.

In this paper, without any cuts on PCB, a 10-mm-heighteddual output LLC resonant converter utilized by two transformersis proposed. Two 10-mm transformers are connected in serieson the primary side and in parallel on the secondary side to over-come current imbalance problems. The turn numbers of trans-formers are calculated considering the connection method oftransformers. Besides, this paper presents a high power densityLLC power module application convenient for LCD/LED TVproducts. The proposed converter offers strictly regulated dualoutput with optimal thermal performance and appropriate EMIresults. The simulation models of the converter are provided bySPICE. Based on the simulation results, an experimental circuitis designed. Output voltage regulation is demonstrated underwide range load conditions (no load – nominal load). Compari-son of simulation and experimental results are presented alongwith the thermal performance and EMI results. In addition, us-ing appropriate secondary windings connection, a current imbal-

ance analysis of dual output LLC module is presented along withthe effects of reducing leakage inductances. Also efficiency of10-mm LLC power module is compared against with the13.5-mm power module using the same structure of two trans-formers. Design consideration for proposed LLC converter ispresented in Section II. In this section, analysis and simulationmodel of the proposed converter will be discussed. In Section III,design example and experimental results are given to validateproposed converter. Section IV gives the conclusion.

II. DESIGN CONSIDERATION FOR PROPOSED LLC

By analyzing resonant transformer core materials that dis-played the fact that one transformer is not enough to get highpower output with the small size cores. To achieve high powerdensity purpose, most effective method is the series primaryand parallel secondary connection of the two transformers [10].In addition by using this connection method, secondary leak-age inductance reduction is achievable. The topology and keywaveforms of conventional LLC resonant converter have beendiscussed in [1]–[3], [16], and [22] together with working prin-ciples and operation modes of the conventional converter. Thelow-profile LLC resonant converter with two transformers, se-ries in primary side and parallel in secondary side structureis proposed for considering high power density purpose andtogether with fulfilling the thermal and electromagnetic com-patibility (EMC) standards without using any extra filter com-ponent. The profile of the series–parallel-connected converteris optimized for 10-mm limit without any drop-in cuts on thesurface of PCB.

A. Analysis

The most common method to analyze LLC resonant con-verters is fundamental harmonic approximation (FHA). In LLCresonant converters, by using two switching components witha 50% duty cycle, a square- wave input voltage is generated.In FHA method, input-to-output power transfer is representedby the fundamental component of this generated input source.Input-to-output voltage functions and gain calculations can berealized by the aid of FHA [1], [16]. These functions and cal-culations are used to determine the design process of LLCconverters.

The low-profile series–parallel-connected LLC converter pro-posed for high power density is shown in Fig. 1. M1 and M2 arethe switching components used for generating the voltage of aresonant tank. As mentioned before, two transformers are usedto obtain 200 W (24 V/6 A–12 V/4.5 A) output power. LR1 is theresonant inductance and LM 1 is the magnetizing inductance ofthe first transformer. LR2 and LM 2 are the resonant and magne-tizing inductances of the second transformer, respectively. CR

is the resonance capacitor of the LLC converter. D1 and D2 arethe output rectifiers of the first output voltage and D3 , D4 arethe output rectifiers used for the second output voltage. CO1and CO2 are the output capacitors. RL 1 and RL2 resistances areused to represent output loads. Aforementioned FHA method isapplied to the series–parallel-connected LLC converter to obtainthe input-to-output voltage function.


Fig. 1. Series–parallel-connected LLC converter proposed for high powerdensity.

Fig. 2. (a) AC equivalent model. (b) Simplified equivalent model.

Considering the transformers individually, because of the se-ries connection in the primary side, the Vin input voltage isshared between the transformers. Under this condition, eachtransformer outputs the half of the total output power whichcorresponds to 100 W (24 V/3 A–12 V/2.25 A). In Fig. 2(a),equivalent model of one transformer providing half of the outputpower is shown. n1 = (NP /NS 1) and n2 = (NP /NS 2) are theturn ratios of the transformer for both outputs. RL1 and RL2 are

the output load resistances. RAC1 and RAC2 are the equivalentac load resistances shown in primary considering the turn ratiosof the transformer. Assuming the two transformers are identi-cal, we can simplify the equivalent model and rewrite the circuitparameters as follows:

LM = LM 1 + LM 2 (1)

LR = LR1 + LR2 (2)

CR = CR1 + CR2 . (3)

Fig. 2(b) shows the simplified equivalent model of transform-ers connected in series in the primary side and in parallel inthe secondary side. Because of the series primary connection,the number of the primary turns is duplicated due to the paral-lel connection in the secondary side, secondary turn numbersremain unchanged. Therefore, the RL1 and RL2 output load re-sistances are divided in half because input voltage is applied toLM = LM 1 + LM 2 and assuming one transformer outputs thetotal power. For these reasons, the turn ratios of the transform-ers and the value of output load resistances can be obtained asfollows:

n′1 = 2n1 , n′

2 = 2n2 (4)

R′L1 =

RL1

2, R′

L2 =RL2

2. (5)

By using the above assumptions, equivalent model of convertercan be obtained. Fundamental root mean square (RMS) value ofthe input square-wave voltage Vin can be obtained by followingequations:

Vin =√

2π

× VDC (6)

where VDC is the maximum input voltage. Fundamental RMSoutput voltages Vo1 , Vo2 are given as follows:

VO1 =2√

2π

× n′1 × Vout1 (7)

VO2 =2√

2π

× n′2 × Vout2 . (8)

RMS output currents Iout1 , Iout2 are shown as following:

IO1 =π

2√

2× 1

n′1× Iout1 (9)

IO2 =π

2√

2× 1

n′2× Iout2 . (10)

From (7) and (9), the expression of equivalent ac resistance forthe first output is given by

RAC1 =VO1

IO1=

8n′21

π2 × R′L1 . (11)

From (8) and (10), the equivalent ac resistance can be derived

RAC2 =VO2

IO2=

8n′22

π2 × R′L2 . (12)


Fig. 3. Asymmetric connection of the proposed converter.

As shown in Fig. 3(b) for dual output the equivalent ac resis-tances are connected in parallel. Both the models of the trans-formers can be represented with the same RAC1 //RAC2 From(4), (5), (11), and (12) total equivalent resistance RE can bederived as follows:

RE =16n2

1n22RL1RL2

π2(n21RL1 + n2

2RL2). (13)

After obtaining (13), the voltage gain MAC and characteristicparameters of the converter can be derived. The complex acanalysis method is used to derive these parameters. The voltagegain of the proposed converter can be obtained as follows:

MAC(S ) =RE × sLM

RE + sLM

1sCR

+ sLr + RE × sLM

RE + sLM

(14)

(14) shows input–output relation of the LLC converter ins-domain. Frequency domain analysis (s = jω) can be definedto express normalized version of (15). Input to output voltagegain functions becomes

|MAC | =∣∣∣∣

mf 2n

[(m + 1) f 2n − 1] + j [(f 2

n − 1) fnQE m]

∣∣∣∣

(15)

where fn is used for frequency normalization, m = LM /LR isthe inductances ratio value and QE =

√LR/(CR × RE ) is the

quality factor of the proposed resonant converter. Gain curvesfor the proposed converter can be optimized from (15) for de-termining the design parameters.

In Fig. 3, asymmetric connection of the series–parallel-connected LLC converter is shown. In this connection method,

secondary windings are swapped to reduce secondary leakageinductances. Because of the secondary leakage inductance ig-norance in the FHA model [3], [11], [23], the input-to-outputtransfer function is the same as (15).

B. Modeling and Simulation

Simulation model is used to obtain an idea about the behaviorof the converter proposed for high power density at differentinput voltages and output loads in order to observe the nonidealconditions before implementing the circuit. In series primary,parallel secondary connection method, because of the diversitybetween the transformers and undesirable parasitic caused bythe imperfection of windings, some adverse conditions may oc-cur. For this reason, a model of the converter is composed. Inthis model, leakage and parasitic components are added to theproposed converter scheme to observe abovementioned draw-backs of using two transformers. By the aid of simulation model,frequency deviation of the converter is evaluated under differentloads and nominal/minimum input voltage range. In addition,the effect of the equivalent series resistance of the output capac-itors is examined. Tolerance analysis of the inductances withina specified margin is realized in order to evaluate the workinglimits of the converter limits. LTSPICE is used for preestimationanalysis of abovementioned criteria. The simulation results aregiven in the Section III along with the experimental results.

C. Transformer Mismatch

Mismatches of transformers between windings are unavoid-able caused by such occurrences as ampere-turn voids and resultin a net axial force [24]. Two parallel transformers affected bymismatch result an unequal distribution of the load currentsbetween two transformers. Transformers could be mismatchedbecause of several things. The most prominent causing effectsare presented below.

1) Core material caused by the imperfection, while construct-ing the ferrite material which is used in the transformers;

2) Air gap difference between two transformers;3) The imperfection of the wires which is used for generation

of windings, causes inequality for the inductance value oftransformers.

In order to prevent undesired performance drop, some toler-ance limits for transformers are defined in the simulation.

III. DESIGN EXAMPLE AND EXPERIMENTAL RESULTS

The design procedure of the converter proposed for highpower density based on theoretical analysis and simulationmodel was discussed in Section II. To validate the designedconverter, a prototype with 200-W power output is realized. Asmentioned before, 10-mm height two transformers are utilizedto design a 200-W output power. The output voltages and loadcurrents are specified as 24 V/6 A and 12 V/4, 5 A respectively.The nominal input voltage is specified as 400 V and the inputvoltage range is between 330 and 400 V. The design steps arecalculated with the assumption of two transformers combined


Fig. 4. Switching waveforms, primary side current and resonance capacitor voltage at nominal input voltage. (a) Vin = 400 V and nominal load (24 V/6 A−12 V/4,5 A). (b) Vin = 330 V and nominal load (24 V/6 A−12 V/4, 5 A). (c) Vin = 400 V and no load. (d) Vin = 330 V and no load.

in a one transformer structure. After the parameter extractionsare completed, the values of parameters distributed to the trans-formers equally.

The turn ratio of the transformer is calculated using the input–output voltage relation and minimum gain M in (16) and (17).From (15) and using peak gain curves, M value is estimated.Since we have two outputs, the turn ratios can be shown asfollowing:

n′1 =

Np

Ns1=

Vin

2(Vo1 + VF )M (16)

n′2 =

Np

Ns2=

Vin

2(Vo2 + VF )M (17)

where n′1 is calculated as n′

1 = 9.62 and n′2 is calculated as n′

2 =19.25. Here, VF is the forward voltage drop of the rectifier usedfor both outputs. Using (13) RE can be calculated as RE =218.1 Ω where R′

L1 = 4 Ω and R′L2 = 2.66 Ω. Based on peak

TABLE IDESIGN SPECIFICATIONS AND CIRCUIT COMPONENTS

gain curves [25], [26] Q is selected as Q = 0.4 and resonancefrequency fR determined as 81 kHz. Using Q and fR , resonancecapacitance CR can be calculated as CR = 1/(2πQfR RE ). Oncethe CR is determined, LR resonance inductance can be writtenas LR = 1/4π2fR

2CR .m value determines the ratio between LR and LM . Decreased

m values resulted in higher circulating current caused by the


TABLE IICALCULATED AND MEASURED CIRCUIT PARAMETERS OF THE CONVERTER

Fig. 5. Secondary side rectifiers current and voltage waveforms under differentinput voltages at nominal load. (a) Vin = 400 V, Vd , and ID of 12 V and 24 V.(b) Vin = 330 V, Vd , and ID of 12 V and 24 V.

lower magnetizing inductance. m is selected as m = 3, 9 by afew iterations for optimizing design procedure.After determin-ing m, we can obtain the magnetizing inductance. The design

Fig. 6. ILM and output voltages Vo1 = 24 V and Vo1 = 12 V.

Fig. 7. Output ripple level of the output voltages Vo1 and Vo2 at nominalinput voltage.

Fig. 8. Output ripple level of the output voltages Vo1 and Vo2 at minimuminput voltage.

specifications and the circuit components of the proposed con-verter are summarized in Table I. Calculated and measured pa-rameters of the converter are presented in Table II. The value ofinductances and number of the turn ratios are given for all thestructures and each transformer, respectively.


Fig. 9. Switching waveforms of low side MOSFET, primary side current, secondary side rectifying current, and resonance capacitor voltage at nominal andminimum input voltage. (a) Vin = 400 V, and nominal load 24 V/6 A−12 V/4.5 A. (b) Vin = 330 V, and nominal load 24 V/6 A−12 V/4.5 A. (c) Vin = 400 V,and no load. (d) Vin = 330 V, and no load.

To verify the theoretical analysis of the converter proposedhigh power density, the simulation results under different inputvoltages and output loads are given in Fig. 4. The simulationresults are realized using LTSPICE. Primary side waveforms ofthe proposed converter are shown in the waveforms shown inFig. 4. Vgs1 and Vgs2 are the gate voltages drive the switchingmosfets M1 and M2 . ILM is the magnetizing current of theprimary side and VC R is the voltage across the resonance ca-pacitor. From Fig. 4(a) and (c) frequency variation of M1 ,M2and changes on ILM and VC R , can be seen under different out-put load conditions. Fig. 4(a) and (b) shows the effect of inputvoltage change. As the input voltage decreases, the operationfrequency also decreases and the behavior of the converter ap-proaches to the capacitive region. This effect can be seen by theshape of ILM .

In Fig. 5, secondary side rectifiers voltages Vd and currentsID at nominal input voltage are prensented.

From Fig. 5, it is clear that the minumum input voltage, causedby the decreased frequency, secondary rectifiers peak currentlevel is increased compared to nominal input voltage appliedto the resonant tank of the converter. The experimental resultsof the converter are presented to validate theoretical approachand simulation results. In Fig. 6, the output voltage levels andILM primary magnetizing current of the transformers are given.Because of the cross regulation of the windings, the control ofoutput voltage ripple is a challenging work. Fig. 7 and Fig. 8show the voltage ripple of the output voltages at nominal andminunum input voltage. It is clear that ripple levels for bothoutputs are lower than 500 mV.

Fig. 10. Output rectifier waveforms. (a) at nominal voltage Vin = 400 V and(b) at minimum input voltage Vin = 330 V.


Fig. 11. Most heated parts of the low-profile converter and maximum temper-ature values tested under 25 ◦C ambient temperature.

Fig. 12. Conducted emissions test result of the converter.

Fig. 13. Radiated emissions test result of the converter.

The waveforms of resonance capacitor voltage, 24 V outputrectifier, magnetizing current on the primary side, and the gatevoltage of the low side MOSFET are shown in Fig. 9. Fig. 9(a)and (b) shows the waveforms of abovementioned circuit pa-rameters under the nominal load condition and different inputvoltages. Decreasing the input voltage affects the switching fre-quency. Peak levels of VC R and ID increases because of the de-creased frequency in order to provide the output power. Underno load condition, waveforms of the low-profile converter are

Fig. 14. Efficiency comparison of a 13.5 mm LLC converter and proposed10 mm converter at nominal input voltage Vin = 400 V, and nominal load24 V/6 A−12 V/4.5 A.

Fig. 15. 24 V output currents of the rectifiers. (a) without leakage reductionand (b) with leakage reduction.

given in Fig. 9(c) and (d). The variation of the circuit parameterscan be observed apparently. Operating frequencies are obtainedas 65 and 81 KHz at 330 and 400 V, respectively, under the nom-inal load condition. At the nominal input voltage with no loadcondition, the operating frequency is 186 KHz which is limitedby controller IC which includes an accurate current-controlledoscillator. The switching conditions of the output rectifiers areshown in Fig. 10 at the nominal output load. The impact of inputvoltage on rectifiers can be seen in Fig. 10. Fig. 11 shows thethermal performance of the low-profile converter by display-ing the temperatures of the most heated parts. The converteris tested at nominal load and 25 ◦C ambient temperature. Themaximum observed temperature value is 69.2 ◦C. In addition,the converter for high power density is applied to LCD/LED


Fig. 16. Efficiency comparison between two secondary connection structuresat nominal input voltage Vin = 400 V, and nominal load 24 V/6 A−12 V/4.5 A.

Fig. 17. Output rectifier currents (ID1 and ID2) based on transformer’s turnratio mismatch condition.

TV module to validate EMC criteria. In Figs. 12 and 13, theresults of the EMC tests for conducted and radiated emissionsare given. The EMC standard limits are obtained with the heightlimit of 10 mm without using any extra filter component.

A 13.5-mm LLC module is also designed with the same circuitparameters to verify and compare the efficiency results of theproposed converter. Fig. 14 shows the efficiency comparisonof 10 and 13.5-mm LLC modules. The low-profile converter is0.04% more efficient at 90% load condition and above. At 100%load condition, improvement made by the converter is 0.22%.Over 89.4% efficiency is achieved at nominal load condition.In other load conditions, height impact is more obvious andefficiency of 13.5 mm converter is higher.

In order to reduce the current imbalance caused by leakageinductances, secondary windings are connected asymmetrically.In Fig. 3, asymmetric connection for dual output series, primaryand parallel secondary connection is shown. By the aid of this

Fig. 18. Illustration of the low-profile transformers. (a) side view and (b) topview.

connection method, the voltage and current waveforms of theoutput rectifiers are obtained.

Fig. 15(a) shows the output currents of D1 and D2 rectifiers.Because of the leakage inductances, the secondary side currentis not shared equally between the output rectifiers. In Fig. 15(b),the effect of the asymmetrical connection is shown. The currentpeak levels of the rectifiers become closer. To estimate the per-formance of this connection, the efficiency of the asymmetricalconnected converter is shown in Fig. 3 is compared with theefficiency of the converter shown in Fig. 1. In Fig. 16, the com-pared efficiency results of the leakage inductance reduction aregiven. From the results, maximum 0.09% efficiency differenceis obtained. It is obvious that for 200-W output power range,efficiency is not sensitive to the leakage inductances.

Peak currents imbalances of the output rectifier diodes oc-curred due to the turn-number mismatches of the transformerwhich causes difference in the leakage inductances at the sec-ondary side. Simulations are carried out by setting 10% variationon transformer’s turn ratio. There is maximum 4.3% differencefor the peak levels of ID1 and ID2 shown in Fig. 17. Unevendistribution currents of the output rectifier diodes also occurred


in Fig. 15(a). Maximum 17% difference for the same rectifiercurrents is obtained without using the leakage reduction method.

Fig 18 shows the realized protoype of the low-profile LLCresonant converter using primary series and secondary parallel-connected magnetic components and the criteria of 10 mm.

A side view and top view of a low-profile transformers utilizedin 200 W LLC resonant converter was illustrated in Fig. 18. Alow-profile power module of 10-mm thickness was built withoutusing any PCB cutting method.

IV. CONCLUSION

This paper proposes a profile-optimized converter based onthe series–parallel-connected LLC structure. 10-mm height dualoutput LLC converter is utilized by two transformers withoutusing any PCB cutting method. In particular, realizing a low-profile power module with high power density is very suitablefor applications in LCD/LED TVs. Analysis, simulation, andexperimental results of the LLC converter have been presentedand compared with 13.5 mm before designed LLC converter interms of efficiency. The 200-W LLC series resonant converterprototype achieves 89.4% efficiency at nomimal load condi-tions. The control of output voltage ripple is a challenging workdue to cross regulation of the windings. Ripple levels for bothoutputs are lower than 500 mV for designed LLC converter.In other words, for different conditions (no load and nominalload), the converter proposed for high power density regulatesboth outputs properly with lower than 2% output ripple voltage.The asymmetric connection is realized in order to reduce the cur-rent imbalance of the rectifiers caused by leakage inductances.Maximum 0.09% efficiency difference between with the leak-age reduction method and without the leakage reduction methodis obtained in the range of 30%–100% load ratio. This result im-plies that efficiency is not considerably affected by the leakageinductances. The maximum observed temperature is measuredas 69.2 ◦C under 25 ◦C ambient temperature. EMC emissionsstandard (EN 55013) is satisfied without using any extra filtercomponent for 10-mm LLC converter prototype. This paper canalso be extended by considering other power modules in powercard of LCD/LED TVs.

ACKNOWLEDGMENT

The authors would like to thank Y. Ucar and R. A. Vu-ral for their valuable discussions and collaboration during thisproject.

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Ibrahim Demirel received the B.S. degree in elec-tronics engineering from Uludag University, Bursa,Turkey, in 2010. He is currently working toward theM.S. degree in electronics and communications en-gineering from Yıldız Technical University from Is-tanbul, Turkey.

He is currently with Arcelik Electronics, Istanbul,Turkey as an R&D Engineer. His research inter-ests include design of switching power supplies,resonant converters, high-efficiency and high-power-density dc–dc converters, optoelectronics, and audio

and acoustics signal analysis.

Burcu Erkmen (M’12) received the B.S., M.S., andPh.D. degrees in electronics and communication en-gineering from Yildiz Technical University, Istanbul,Turkey, in 1999, 2001, and 2007, respectively.

From 1999 to 2009, she was a Research Assis-tant in the Department of Electronics and Commu-nications Engineering, Yildiz Technical University,where she has been an Assistant Professor since 2009.She was also engaged in industrial projects involv-ing power electronics. Her current research interestsinclude switching power supplies, resonant convert-

ers, high-efficiency and high-power-density dc–dc converters, and optimizationtechniques in electronic circuits.

Documents

A Very Low-Profile Dual Output LLC Resonant Converter for LCD/LED TV Applications