ACKNOWLEDGMENTS

This work was supported in part by the University Grants Com-

mission (UGC), Government of India and Department of Sci-

ence and Technology (DST), Government of India.

REFERENCES

1. W.C. Liu and W.R. Chen, CPW-fed compact meandered patch

antenna for dual band operation, Electron Lett 40 (2004).

2. W.C. Liu, Broadband dual frequency meandered CPW-fed monopole

antenna, Electron Lett 40 (2004).

3. T.H. Kim and D.C. Park, CPW-fed compact monopole antenna for

dual-band WLAN applications, Electron Lett 41 (2005).

4. C. Wang, Z.H. Yan, P. Xu, J.B. Jiang, and B. Li, Trident-shaped

dual-band CPW-fed monopole antenna for PCS/WLAN applications,

Electron Lett 47 (2011).

5. P. Ashkarali, S. Sreenath, R. Sujith, R. Dinesh, D.D. Krishna, and

C.K. Aanandan, A compact asymmetric coplanar strip fed dual-band

antenna for DCS/WLAN applications, Microwave Opt Technol Lett

54 (2012), 1087–1089.

VC 2013 Wiley Periodicals, Inc.

A NOVEL UWB ANTENNA WITH DUAL-STOPBAND CHARACTERISTICS

Mohammad Akbari, Saman Zarbakhsh, and Marjan MarboutiYoung Researchers and Elite Club, Central Tehran Branch, IslamicAzad University, Tehran, Iran; Corresponding author:akbari.telecom@gmail.com

Received 20 March 2013

ABSTRACT: A novel ultrawideband square antenna with dual band-

notched performances is proposed. To obtain extended impedance band-

width, novel techniques on the ground plane are used. Besides, the stopbands

are realized by a I-shaped defected ground structure as well as a L-shaped

arm ended up shorting pin on the top of the back plane. The experimental

results exhibit that the antenna, with compact size 21 3 17 3 1 mm3, has an

impedance bandwidth of 2.6–11.8 GHz for voltage standing wave ratio less

than 2, except two frequency stopbands of 3.2–3.7, 5.1–5.9 GHz. VC 2013

Wiley Periodicals, Inc. Microwave Opt Technol Lett 55:2741–2745, 2013;

View this article online at wileyonlinelibrary.com. DOI 10.1002/mop.27920

Key words: shorting pin; UWB antenna; notch band

Figure 5 Radiation patterns of the antenna at (a) 2.4 GHz and (b) 5.2 GHz

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 11, November 2013 2741

1. INTRODUCTION

The interest in UWB technology has grown after the US Federal

Communications Commission allocation of the frequency band

on 3.1–10.6 GHz for commercial use [1]. Recently, UWB tech-

nology has been widely used in various radars and has attracted

much attention for communication systems in a way that there

are plenty of UWB antenna which have been published in vari-

ous journals [2–6]. On the other side, the frequency range for

UWB systems will cause interference with existing narrowband

wireless communication systems. As an example, the wireless

local area network for IEEE 802.11a operating at 5.15–5.35 and

5.725–5.825 GHz, the IEEE 802.16 WiMAX system operating

at 3.3–3.6 GHz. Therefore, compact UWB antennas with band-

notched characteristics to avoid the potential interference are

Figure 1 Geometry of proposed antenna

TABLE 1 Optimized Antenna Parameter

Lsub Wsub Lp Wp Lf Wf Lg Wg S d K L1

21 17 12 10 8 1.9 4.5 5.5 0.5 1 0.3 3.5

L2 L3 L4 L5 L6 L7 L8 L9 L10 L11 L12 L13

13 4.5 2 7 3.5 0.8 0.8 2 1.5 4.2 1 1.5

Figure 2 Simulated return loss for different structures showing impedance bandwidth

2742 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 11, November 2013 DOI 10.1002/mop

desirable. Recently, to create the frequency band-notch function,

modified planar monopoles have been recently proposed [7–9].

In Ref. 7, novel shape of the slot (folded trapezoid) is used to

obtain the appropriate band notched characteristics. Single and

multiple [8] half-wavelength U-shaped slots are embedded in

the radiation patch to generate the single and multiple band-

notched functions, respectively. In Ref. 9, band-notch function

is achieved by using a T-shaped coupled-parasitic element in

the ground plane. In this article, a novel ultrawideband square

antenna with two sharp frequency notched bands is presented

and investigated in detail. The antenna consists of a simple

square patch on the front side. By using a I-shaped defected

ground structure (DGS) above the back ground plane, a notched

band of 5.1–5.9 GHz is obtained. To realize another narrow

band of WIMAX centred at 3.5 GHz, a L-shaped arm ended up

shorting pin is used at the upper edge of the back plane. The

measured results of the antenna are presented below.

2. ANTENNA DESIGN

The geometry of the presented antenna is illustrated in Figure 1.

The dimensions of the optimized antenna are depicted in Table 1.

The antenna is fabricated on a 21 3 17 3 1 mm3 FR4 sub-

strate with relative permittivity er 5 4.4. The antenna consists of a

square radiation patch connected to a 50-X microstrip line on the

front side and a partial ground plane on the back side. In order to

achieve a good impedance matching, a couple of inverted L-

shaped slots with length L9 1 L10 1 L11 on the center ground

and an inverted U-shaped arm on the upper edge of the ground is

used. Moreover, to generate double notched bands two different

structures on the back side are used. As mentioned before, by

using a I-shaped DGS above the back ground plane, a notched

band of 5.1–5.9 GHz is achieved. To realize another narrow band

of WiMAX centered at 3.5 GHz, a L-shaped arm ended up short-

ing pin is used at the upper edge of the back plane.

3. RESULTS AND DISCUSSION

The performance of the antenna at parametric studies has been inves-

tigated to find optimized parameters using Ansoft High Frequency

Simulation Structure (ver.13) based on the finite element method

[10]. The measurement of voltage standing wave ratio (VSWR) was

carried out using an Agilent E8362B network analyzer in its full

operational span (10 MHz–20 GHz). As described above, a novel slot

technique on the ground plane and an inverted U-shaped arm at the

upper edge of the ground are used to obtain a good impedance match-

ing which Figure 2 is the best proof for approve of it. According to

Figure 2, the impedance bandwidth increases considerably due to

operating both novel techniques on the ground plane. Conversely,

Figure 3 Simulated VSWR for different structures showing stopbands

Figure 4 Photograph of the fabricated antenna. [Color figure can be

viewed in the online issue, which is available at wileyonlinelibrary.com]

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 11, November 2013 2743

Figure 3 exhibits the effect of I-shaped DGS and L-shaped arm ended

up shorting pin on VSWR curve in a way to create two independent

stopbands from 3.2 to 3.7 and 5.1 to 5.9 GHz. Meanwhile, the simu-

lated result of a reference antenna without notched characteristics is

also depicted for comparison.

The antenna with optimal design, as shown in Figure 1, was

fabricated and tested in the Antenna Measurement Laboratory at

Iran Telecommunication Research Center. Besides, the photo of

the fabricated antenna is shown in Figure 4.

Figure 5 illustrates simulated and measured VSWR versus

frequency for the proposed antenna. It can be observed that the

designed antenna has wideband performance of 2.6–11.8 GHz

for VSWR 5 2, covering the entire UWB frequency band with

dual-notched bands of 3.2–3.7 and 5.1–5.9 GHz. Meanwhile, the

current distribution at two center frequencies of stopbands, 3.5

and 5.5 GHz, are quite apparent. According to the current distri-

bution, it can be concluded that at 3.5 GHz most current is

flowing on L-shaped arm ended up shorting pin, whereas at 5.5

GHz it is flowing on I-shaped DGS. As shown in Figure 5, there

exists a discrepancy between measured data and the simulated

results, and this could be due to the effect of the SMA port. To

confirm the accurate VSWR characteristics for the designed

antenna, it is recommended that the manufacturing and measure-

ment process need to be performed carefully. The simulated

Figure 5 Measured and simulated VSWR characteristics and current distribution at frequencies 3.5 and 5.5 GHz. [Color figure can be viewed in the

online issue, which is available at wileyonlinelibrary.com]

Figure 6 Measured radiation patterns of the proposed antenna at (a) 4 GHz and (b) 7 GHz

2744 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 11, November 2013 DOI 10.1002/mop

radiation patterns at 4, 7 GHz are depicted in Figure 6. It can be

seen that the antenna has a nearly omnidirectional radiation pat-

tern in the H-plane (x-z plane) and a dipole-like radiation pattern

in the E-plane (y-z plane).

4. CONCLUSION

A novel compact UWB planar antenna with dual-notched bands

has been presented and discussed. The band notches are realized

by a I-shaped DGS as well as a L-shaped arm ended up shorting

pin on the back plane. In addition, in order to achieve extended

impedance bandwidth novel techniques on the ground plane are

used. The presented antenna exhibits a broad bandwidth and a

good radiation performance.

REFERENCES

1. H. Schantz, The art and science of ultra wideband antennas, Artech

House, Norwood, MA, 2005.

2. M. Akbari, M. Koohestani, C. Ghobadi, and J. Nourinia, Compact

CPW-fed printed monopole antenna with super wideband perform-

ance, Microwave Opt Technol Lett 53 (2011), 1481–1483.

3. M. Akbari, M. Koohestani, C. Ghobadi, and J. Nourinia, A new

compact planar UWB monopole antenna, Int J RF Microwave Com-

put Aided Eng 21 (2011), 216–220.

4. M. Mighani, M. Akbari, and N. Felegari, Design of a small rhombic

monopole antenna with parasitic rectangle into slot of the feed line for

SWB Application, Appl Comput Electromagn Soc 27 (2012), 74–79.

5. M. Mighani, M. Akbari, and N. Felegari, A CPW dual band notched

UWB antenna, Appl Comput Electromagn Soc (ACES) 27 (2012),

352–359.

6. A.A. Eldek, Numerical analysis of a small ultra wideband

microstrip-fed tap monopole antenna, Prog Electromagn Res PIER

65 (2006), 59–69.

7. M. Ojaroudi, Printed monopole antenna with a novel band-notched

folded trapezoid ultra-Wideband, J Electromagn Waves Appl 23

(2009), 2513–2522.

8. M. Ojaroudi, G. Ghanbari, N. Ojaroudi, and C. Ghobadi, Small

square monopole antenna for UWB applications with variable fre-

quency band-notch function, IEEE Antennas Wireless Propag Lett 8

(2009), 1061–1064.

9. R. Rouhi, C. Ghobadi, J. Nourinia, and M. Ojaroudi, Ultra-wideband

small square monopole antenna with band notched function, Micro-

wave Opt Technol Lett 52 (2010), 2065–2069.

10. Ansoft Corporation, Ansoft high frequency structure simulation

(HFSS), Ver. 13, Ansoft Corporation, Pittsburgh, PA.

VC 2013 Wiley Periodicals, Inc.

RECONFIGURABLE BANDPASS FILTERWITH CENTER FREQUENCY ANDBANDWIDTH CONTROL

Dimitra Psychogiou1 and Dimitrios Peroulis2

1 Department of Information Technology and Electrical Engineering,Laboratory for Electromagnetic Fields and Microwave Electronics,ETH Zurich 8092 Z€urich, Switzerland; Corresponding author:pdimitra@ifh.ee.ethz.ch2 School of Electrical and Computer Engineering, Birck Nanotechnol-ogy Center, Purdue University, West Lafayette, IN 47907

Received 20 March 2013

ABSTRACT: This article presents a bandpass filter with independentcenter frequency and bandwidth control. It is based on four frequency-reconfigurable resonators arranged in a series cascade of a second-

order bandpass filter and a dual-behavior resonator. A measured band-pass filter response with variable bandwidth between 440 and 0 MHz is

demonstrated at 3.6 GHz. The center frequency of the filter can be con-

tinuously varied between 2.76 and 3.79 GHz (1.4:1) with continuousbandwidth variation between 300 and 100 MHz (3:1) for a minimumout-of-band isolation of 215 dB and insertion loss between 0.76 and 2.5

dB. VC 2013 Wiley Periodicals, Inc. Microwave Opt Technol Lett

55:2745–2750, 2013; View this article online at wileyonlinelibrary.com.

DOI 10.1002/mop.27898

Key words: bandwidth tuning; center-frequency tuning; tunable band-pass filter

1. INTRODUCTION

Recent advances in wireless communication systems call for

transceiver/receiver architectures with multiband and multistan-

dard operation which in turn require the use of reconfigurable

components. Bandpass filters (BPFs) with tunable center fre-

quency and bandwidth are in high demand to enable adaptive prese-

lection of the desired frequency band and rejection of the undesired

interference. They facilitate reduced cost and volume transceivers

with limited number of switching elements if compared to tradi-

tional implementation concepts based on filter banks. During the

last three decades, research on reconfigurable filters has mainly

focused on center frequency tuning using ferroelectric materials

[1], semiconductor varactors [2], and radiofrequency micro-

electronic mechanical system [3] as tuning elements.

Recently, BPFs with reconfigurable center frequency and

constant or variable bandwidth have been demonstrated [4–13].

Various methods were used to vary the external and the inter-

resonator coupling coefficients. In Ref. 4, tunable coupling

reducers were utilized for a variable bandwidth ratio up to 5.4:1.

An independent magnetic and electric coupling scheme was

demonstrated in Ref. 5 for the realization of a BPF with con-

stant bandwidth. In a subsequent publication [6], mixed electric

and magnetic coupling were utilized with the aid of two distrib-

uted capacitive loads. In an alternative approach, parallel-

coupled switched delay lines were used [7] for a bandwidth tun-

ing ratio up to 3:1. The majority of presented topologies feature

limited bandwidth control and up to 1.8:1 [5,6,8–11]. Bandpass

filters with large bandwidth control often suffer from high-

insertion loss and can be summarized as follows: In Ref. 4, the

authors demonstrated a BPF with bandwidth tuning ratio of

5.4:1 and insertion loss around 10.5 dB at 550 MHz. A BPF

with 3:1 bandwidth ratio and 1.5:1 frequency tuning range was

introduced in Ref. 7 with an insertion loss around 4 dB. In Ref.

13, Guyette et al. presented a frequency-reconfigurable BPF

with bandwidth variations between 120 and 0 MHz but rela-

tively high-insertion loss (e.g., 15 dB for a bandwidth tuning

ratio of 6:1).

In this article, a reconfigurable BPF with center frequency

and bandwidth control without the need of either external or

inter-resonator coupling tuning is presented. The proposed

implementation concept is based on four frequency-

reconfigurable resonators arranged in a series cascade of a

second-order BPF and a dual-behavior resonator (DBR). Tun-

ing is achieved by means of piezoelectric actuators directly

integrated on top of highly loaded evanescent-mode cavity

resonators with high unloaded quality factor (>600, [3]). A

BPF with quasi-elliptic frequency response and reconfigura-

ble bandwidth between 440 and 0 MHz is demonstrated at

3.6 GHz. The center frequency of the filter can be continu-

ously varied between 2.76 and 3.79 GHz with a bandpass

tuning ratio of more than 3:1 and insertion loss between 0.76

and 2.5 dB.

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 55, No. 11, November 2013 2745