A Novel Single-Phase Soft Switching Microinverter for ...abasghar/papers/power...2013 3rd International Conference on Electric Power and Energy Conversion Systems, Yildiz Technical

2013 3rd International Conference on Electric Power and Energy Conversion Systems, Yildiz Technical University, Istanbul, Turkey, October 2-4, 2013

A Novel Single-Phase Soft Switching Microinverter

for Photovoltaic Applications

Saad Pervaiz, Muneeb Ur Rehman, Ahmed Bilal Asghar, Nauman Zaffar SBA School of Science & Engineering, LUMS

Email: (saad.pervaiz.muneeb.rehman.bilal.asghar.nauman.zaffar)@lums.edu.pk

Abstract-Microinverters provide a direct way to integrate an individual PV panel to the AC grid. We introduce a novel microinverter design comprising a high frequency full-bridge converter, a pseudo dc link filter, and a line commutated inverter. The approach is well matched to the requirements for high efficiency DC to AC grid-tied and stand-alone converters. We present a detailed working analysis of the proposed design in continuous, boundary, and discontinuous conduction mode. We also show that our design enables reduced device losses, flexible control, and low current harmonic distortion. We present a soft switching scheme for the high frequency full bridge converter stage. Results for design simulation in PSIMT M and implementation in hardware are presented.

Index Terms-Module integrated converter, photovoItaic, fullbridge, soft-switching.

I. INTRODUCTION

Photovoltaic (PV) provide a clean source of energy and are

capable of satisfying our growing electricity demand. They are

expected to become major contributor to electricity generation

among all renewable energy candidates by 2040 [1]. Grid

parallel inverters are a key component required to integrate PV

systems in electric grid. Various inverter system architectures

have been studied for PV applications over the course of past

few decades [2], [3], [4]. One approach to PV integration is the

use of module integrated converters or microinverters. These

converters provide a direct way to integrate an individual PV

panel to the AC grid. Such integration allows MPPT of indi

vidual solar panels which yields greater energy extraction than

centralized MPPT of series-connected string of PV modules

[5], [6]. Microinverters also provide system scalability and

better performance in partially shaded regions [7].

Multiple microinverter topologies are possible depending

upon the dc link configuration. These topologies are classified

in three different arrangements [6]:

• Microinverter with a DC link

• Microinverter with a pseudo DC link

• Microinverter without a DC link

A micro inverter interfaced with a single PV panel needs

to provide high voltage transformation under varying input

voltage conditions. A single PV panel provides a low voltage

(20-40 V) at input which needs to be transfonned to AC grid

voltage (220 V, rms). In such applications, efficiency, size and

cost are the driving design considerations for microinverters.

High efficiency can be achieved through soft switching and

reduced reverse recovery losses in rectifier diodes. Cost of in

verter is proportional to the size of energy storage components

Yin ±

Full· bridge Converter Line Com mutated Inverter

Fig. 1. Proposed DC-AC Inverter Design

and transfonner. High switching frequency design is one way

to bring down the size and cost of inverter.

In this paper, we investigate a microinverter based on the

architecture of Figure 1, comprising a high frequency full

bridge converter, a pseudo dc link filter, and a line cOlmnutated

inverter. Similar general architectures ([8], [9]) have long

been known but this specific design has not been explored

in previous work (e.g. see topology review in [3], [4]). We

propose a novel design that reduces device losses, enables

flexible control, and low current harmonic distortion. All

switching devices operate under zero voltage/zero current

switching enabling high efficiencies to be achieved. A trailing

edge pulse width modulation scheme is employed for the high

frequency full bridge converter stage.

Rest of paper is organized as follows: Section II provides

converter design and working. Section III and IV provide

simulation and experimental results to validate the working of

proposed converter. We conclude in Section V by discussing

the benefits of our inverter design.

II . PROPOSED DC-AC INV ERTER

A. Design The proposed inverter topology consists of a full bridge con

verter cascaded with a line commutated inverter, as illustrated

in Figure 1. The design consists of an active H-bridge, S 1-

S4, on the low voltage side of a transformer 'T'. A full-wave

rectifier, D5-D8, on the high voltage side of the transformer

is cascaded with another active H-bridge, S9-S12, which is

connected to the grid supply. The body diodes, D I-D4, of

MOSFETs, S I-S4, are explicitly shown. Cin and C1 are filter

capacitors at the input and pseudo DC link respectively. V in

978-1-4799-0688-8/13/$31.00 ©2013 IEEE


and V 9 mark the input PV voltage and grid supply at output

respectively.

B. Working & Analysis

The proposed converter, shown in Figure 1, consists of two

stages: boost full-bridge stage and line commutated inverter

stage. The full-bridge stage operates at high frequency and pro

duces a rectified sinusoidal PWM waveform which is filtered

by the pseudo-dc link capacitor. This stage is responsible for

providing the required boost in dc voltage for grid integration.

The second stage operates at line frequency (50/60 Hz) and

opens the rectified output waveform of previous stage. Since

this stage operates at line frequency which is much smaller

than operating frequency of first stage (2:: 40 kHz), we can

divide the inverter operation into two steps; positive half cycle

of grid voltage during which S5 and S8 are closed and negative

half cycle during which S6 and S7 are closed. Without any loss

of generality, complete inverter operation can be explained by

just considering the positive half cycle of grid voltage and the

working principle of full-bridge converter during this cycle.

PWM waveforms for the operation of full-bridge converter

are given in figure 12. In this scheme, the lower switches

(S3, S4) are operated at fixed (50%) duty cycle while the top

switches (S I, S2) are trailing edge pulse width modulated.

The converter produces a sinusoidal PWM waveform which

is rectified and filtered at the pseudo-dc link stage. At this

link, a rectified line frequency AC waveform is produced. We

can assume the converter to be operating as a dc-dc converter

for small signal analysis since the operating frequency of full

bridge converter is much higher than line frequency (in other

words change in line frequency waveform is approximately

very small when viewed at 2:: 40 kHz operating frequency of

full-bridge converter).

The full-bridge converter can be operating in Continu

ous Conduction Mode (CCM), Boundary Conduction Mode

(BCM) or Discontinuous Conduction Mode (DCM). The cir

cuit operation in CCM, BCM, and DCM depends on the value

of duty cycle. Duty cycle is a function of output load, however

in our proposed technique, the duty cycle also depends on

grid voltage at output and therefore varies continuously as it

is being modulated by the grid voltage. Using steady state

analysis, it can be shown that converter might be working

in CCM, BCM and DCM during one complete cycle of grid

voltage, even for a fixed amount of output power delivered.

The converter operation can be divided in three operat

ing intervals for positive half cycle of grid voltage. These

intervals are determined by the on/off states of the four

primary switches (S I-S4). During the negative half cycle

of grid voltage, the operation of full-bridge stage remains

unchanged while the line commutated inverter flips the states

of diagonal bridge switches. In the analysis that follows,

MOSFET switches have been modeled with parallel diodes

and parasitic capacitances. The transformer leakage inductance

is shown explicitly, as it is fundamental to circuit operation.

Gl

G4

G2

G3

D

�

D

- Vin

r------I--.......

� l..---I--to t1 t2 t3 t4

(a) Continuous Conduction Mode

Gl

G4

G2

G3

D

f :

I---

V

--- Vin

� to t1

� p==.-

I---

I--

I� V t2 t3 t4

(b) Discontinuous Conduction Mode

tS t6

tS t6

Fig. 2. PWM Gate Signals (GI-G4) of Switches (Sl-S4), Voltage waveform across transformer primary winding, and current through leakage inductance of transformer in CCM & DCM mode

The detailed circuit operation in all three modes is as

follows:

[to-tl] - Energizing Leakage Inductance

to-tl marks the period when S 1 and S4 are closed till the

opening of Sl. In this time period voltage, Yin, appears

across the primary side of the transformer and a current

starts building in the leakage inductance. The corresponding

secondary current completes its path through the grid/load and

thus transfers power to the output. The path of current for this

time period is shown in Figure 3. During this interval, current

builds in the leakage inductance and power is transferred to

output. The circuit operation is similar in CCM, BCM, and

DCM for this interval.

[tl-t2] - De-energizing leakage inductance

During tl-t2, switch S I is opened while S4 remains closed.

Since the current at the primary cannot stop instantly due

to transformer leakage inductance, the inductor current dis

charges switch S3 capacitor until diode D3 is turned on and

makes an alternate freewheeling path. During this time interval


+ DS

'--_---' '--rrrr.----=-l.: N

S3

)7 D8

Fig. 3. [to-tIl - Energizing Leakage Inductance

+ DS

v,

)7 D8

Fig. 4. [tl-t2] - De-energizing leakage inductance

rv Vgrid

the input is disconnected, therefore current dies out as energy

stored in leakage inductance is transferred to the output. The

circuit operation differs depending on the conduction mode

(CCM, BCM, DCM). In CCM and BCM, the current is still

non-zero at the end of this interval; therefore a corresponding

current is flowing through the secondary side as well. In

DCM, the inductor is completely de-energized and the current

becomes zero before the end of this interval. This interval ends

when S4 is opened.

[t2-t3] Dead band mode

This mode begins when S4 is opened at time t2. In DCM, the

current is already zero so this interval is not eventful for DCM.

However, in CCM the current through the leakage inductance

is not zero. So the inductor current discharges capacitor C2

and turns diode D2 on. The current flows through D3 and D2

during this interval. The rectifier current on secondary side is

zero as there is no current through the primary winding of

transformer. Therefore no power is transferred to the output

during this mode. The circuit operation is similar in BCM as

compared to CCM, however the current becomes zero at the

end of this interval in BCM.

[t3-t6] Complementary Intervals

These intervals are similar in operation to those of [to-t3].

The transformer leakage inductance is energized in opposite

direction as compared to the first interval [to-t1]. Complimen

tary diagonal rectifier diodes are turned on during these three

intervals.

The full-bridge converter DC gain in DCM and BCM is

given by equation 1 and 2, respectively. These equations have

been derived in [10]. Here n is the transformer turn ratio, D

is the duty cycle of top switches (S 1, S2), k is the normalized

time constant of the converter, L is the leakage inductance of

)1

Fig. 5. [t2-t3] Dead band mode

+

Dil

transformer, R is the load resistance corresponding to current

delivered to the grid, and T s is switching period.

Vo 2n MDCM = - =

--V-===4=k Yin

1 + 1 + D2

The normalized time constant of converter is given by:

4n2L

(1)

(2)

k = RTs

(3)

C. Device stresses and Switching Losses

All full-bridge converter switches are zero voltage turned

off due to the drain-to-source capacitance of the MOSFET

switches. So turn off losses for all the switches are minimal.

Furthermore, in discontinuous conduction mode, during the

interval tl-t2, the inductor current falls to zero so all the

switches turn on at zero current. The switches S3 and S4 turn

off at zero current in DCM. In CCM, all the switches achieve

zero voltage switching.

The line comrnutated inverter stage has minimal losses due

to following reasons:

• It is being switched at line frequency. So the switching

losses are negligible.

• The switching takes place at zero crossings of the grid

voltage/current. The voltage at dc-link is very small (close

to OV). Therefore the switching takes place at very low

voltage.

At turn on, the only losses are the conduction losses that can

be reduced by using devices with low on resistance or by

paralleling switching devices.

I I I. RESULT S

The proposed design was validated in simulation exper

iments and hardware prototype testing. The converter was

designed to meet the specifications given in Table I. The

components used in making hardware prototype are listed in

Table II.


Pseudo DC link

DC Full bridge Full wave rectifier Unfolding h-bridge Grid "v

Grid Voltage

x)-------'--------'------------' Current demand constant

Fig. 6. Block Diagram of Feedback Control Blocks for Proposed Converter

TABLE I SIMULATION AND HARDWARE CONVERTER DESIGN SPECIFICATIONS

No. Parameter Value

1 Input Voltage (DC) 20-40 (V )

2 Output Voltage (AC) 220 (V rms)

3 Rated Output Current 2.8 A

4 Rated Output Power 600 W

5 Switching Frequency (FuU-bridge) 48 kHz

TABLE II COMPONENTS USED IN HARDWARE PROTOTYPE

Component Part #

MOSFET (Low Voltage side) IRF3710

MOSFET (High Voltage side) IRF740

Power Rectifier Diodes RHR15120

Input Filter Capacitor 10,000 uFIlOO V

Pseudo-dc Link Capacitor 7.5 J.LF

Transformer Ferrite Core (L=2.5 J.LH)

A. Simulation Experiments PSIMT M was used for simulating the proposed converter

design. The model parameters used in simulation were actual

component parameter that are given in table II. The simulation

was done both for stand-alone and grid tied system. Here,

we discuss the results for grid connected circuit simulation.

Feedback control for grid connected operation is shown in

figure 6. A number of test case experiments were done in

simulation experiments.

We varied the input voltage to see the behavior of inverter.

Figure 7 shows the output current waveform of the inverter

at an input voltage of 26 V. The converter was operating in

grid-parallel mode in this simulation experiment. Figure 8 and

9 show similar waveforms for input voltage of 30 V and 36 V

respectively. The output current was fixed at 2 A in all three

cases. % THD of the current fed to the grid was less than 4%

even in all experiments.

We also changed the output current fed to the grid during

3 -------------�-------------�------------, , , ,

-3 - - ------- - ---�--- - ------- - -:_----- - -------:_------- - -----, , , 0.0 1 0.02

T i m e (s) 0.03 0.04

Fig. 7. Test Case 1: Input Voltage 26V; Output Current 2 A; Grid Voltage 220 V

3 -------------�-------------�-------------�-------------

� 1

� 0

8 ·1

, , , , , ,

-3 -------------:--------------:--------------;.. -------------, , , 0 .01 0.02

T i me(s) 0.03 0.04


online operation. Output current could be changed at zero

crossings to minimize current THD. Figure 10 shows the

results for output current change. The smooth transition in

output current shows the agility of feedback loop.

B. Hardware Prototype A 600 Watt prototype, of the proposed converter, was made

to validate the working of proposed converter topology and

control scheme. The components values are listed in Table II.

MOSFETs are used in the line-commutated stage because of

the low power levels, ease of control and low conduction losses

as compared to thyristors. The hardware prototype picture is

shown in figure 11.

The prototype inverter was built and tested for stand

alone systems. The predicted and simulated waveforms were

confirmed. Some of the operating waveforms are given in


3

� 2

'p' C � 0 ... 8 -1

-2 -3

o

I I I I I I - - - - -� - - - - -,- - - - - -.- - - - - -.- - - - -, - - - - - T -----I I I I I I

0.01 0.02 T i me (s )

0.03


, , , 3 ··············;. ·············· T ·------------- ,. --------------

$1 c: 0 � (3 -1

, , , , , , , ,

0.01 0.02 Time(s)

0.03 0.04

Fig. 10. Test Case 4: Output Current changed from 2.8 A to 2 A; Input Voltage 30V; Grid Voltage 220 V

Figure I2(b) and I2(a). The voltage waveform at pseudo dc

link is shown in figure I2(a). The experimental results of

converter are given in stand-alone and open loop configuration

with a first order filter. Because of the absence of the control

loop, the pseudo-dc link voltage is not sinusoidal. The output

current and voltage waveforms are shown in figure I2(b).

IV. CONCLUSION

A new design topology and control scheme is proposed for

a grid-tied photovoltaic micro inverter. The proposed inverter

can be utilized in grid-tied and stand-alone configurations for

PV systems. The topology is based on a high frequency full

bridge converter, a pseudo dc link filter, and a line commutated

inverter. The operational characteristics of the converter are

analyzed in detail for continuous, boundary, and discontinuous

conduction mode. A soft switching technique is presented for

the proposed converter. Simulation and experimental results

are demonstrated for proof-of-concept design specifications.

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Fig. 11. Hardware Prototype

(a) Voltage waveform at pseudo dc-link (100 VI division)

(b) Yellow: Output Voltage (scale 100 V I division); Blue: Output Current (Sensed via 0.1 Resistor) (scale 2 A I division)

Fig. 12. Hardware Prototype Results

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