A multiband hybrid antenna with double layer structures for mobile devices

fiber end under test and the fiber for detection were measured at

1.31- and 1.55-lm wavelengths using laser diodes (LDs), an

optical power meter (OPM), and 3-dB coupler. When both the

measured return losses at those two wavelengths were equal to

�14.7 dB (from 14.7 to 15.7 dB in our experiments), the part of

the fiber end under test could be identified as uneven [shown in

Fig. 1 (a)]. If one of the two losses is not equal to �14.7 dB,

the part can be identified as flat [shown in Fig. 1(b)]. Every part

of the fiber end under test is covered under the scanning inspec-

tion areas and can be determined as uneven or flat. Conse-

quently, the cleaved fiber end under test can be identified as suc-

cessful or failed.

In the experiments, the gap between the fiber for detection

and the fiber under test was set to 40 lm, and each scanning

distance was set to 10 lm. Typical experimental results are

shown in Figure 4. In the figure, (a) and (c) show the flat parts

of the inspected fiber end found using the proposed inspection

method and (b) and (d) shows the SEM image of the flat end.

The fiber ends in Figures 4(a) and 4(c) were found to be cor-

rectly and incorrectly cleaved, respectively. The experimental

image with a correctly cleaved fiber end shows that the flat

parts form a circle of about 140-lm diameter, which is a little

larger than the actual 125-lm diameter fiber end. This is

because a mode field area of light may radiate from the fiber

end for detection. In contrast, the experimental results for the

incorrectly cleaved fiber end show that half the fiber end parts

are flat and the remainder is uneven. The results of the pro-

posed inspection method and those from SEM observation are

in good agreement.

4. CONCLUSION

We have proposed and demonstrated a novel fiber optic sensor

for inspecting cleaved fiber end based on the Fabry–Perot inter-

ferometer. The sensor uses LDs, OPM, 3-dB coupler, and XY

lateral adjustment stage. The images obtained for the inspected

fiber ends were in good agreement with SEM observation

images.

REFERENCES

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market, IEEE Commun Mag 43 (2005), 72–78.

2. D. Gloge, P.W. Smith, D.L. Bisbee, and E.L. Chisnnock, Optical

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(1973), 1579–1587.

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ment of an automatic cutting tool for optical fibers, IEEE/OSA J

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formance of field assembly connectors using incorrectly cleaved

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Winstone, New York, 1985.

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OFC/NFOEC2010, NWE4, 2010.

VC 2012 Wiley Periodicals, Inc.

A MULTIBAND HYBRID ANTENNA WITHDOUBLE LAYER STRUCTURES FORMOBILE DEVICES

Chow-Yen-Desmond Sim, Chun-Heng Chao, and Wen-Bin HeDepartment of Electrical Engineering, Feng Chia University,Taichung, Taiwan 40724, Republic of China; Correspondingauthor: [email protected]

Received 25 August 2011

ABSTRACT: The design of a hybrid antenna with double layer

structures that allow multiband operation in the global system formobile communications/digital communication systems/personal

communication system/universal mobile telecommunications system andwireless local area network band bands is presented. By combining theresonant modes excited separately by the upper layer planar inverted-F

antenna structure (0.88, 0.97, and 1.18 GHz) and lower layer monopoleantenna type structure (1.38, 1.76, and 2.45 GHz), a dual-frequencyoperation with broadband characteristics can be observed, whereby the

lower and upper frequency bands are operating between 0.86 and 1.00GHz and between 1.15 and 2.63 GHz, respectively. From the

experimental results, stable gain and good radiation efficiency are alsoexhibited within the bands of interest. VC 2012 Wiley Periodicals, Inc.

Microwave Opt Technol Lett 54:1543–1548, 2012; View this article

online at wileyonlinelibrary.com. DOI 10.1002/mop.26863

Key words: hybrid antenna; double layer; multiband; planar inverted-Fantenna; monopole

1. INTRODUCTION

Owing to the rapid development of wireless communication sys-

tem, to provide a wide variety of services so as to meet the

exponentially growing user needs, designing a mobile device (or

mobile phone) with multiband operation is currently a challeng-

ing topic for the antenna engineers. Over the past decade, multi-

band antenna designs operating within the third generation (3G)

mobile services that includes the global system for mobile com-

munications (GSM, 890–960 MHz), digital communication sys-

tems (DCS, 1710–1880 MHz), personal communication system

(PCS, 1850–1990 MHz), and universal mobile telecommunica-

tions system (UMTS, 1920–2170 MHz) bands have been

reported [1–5]. As most modern mobile devices are now able to

access into the internet; hence, integrating the wireless local

area network band (WLAN, 2400–2480 MHz) for IEEE 802.11

b/g into the aforementioned operating bands are now a prerequi-

site requirement for future antenna design [6–9].

Recently, the techniques of applying parasitic element [10,

11] and integrating two different antenna structures (also refer

to as hybrid antenna) [12–14] have been proposed to achieve

multiband operation for mobile handset applications. For the

parasitic element technique, single [10] or multiple [11] parasitic

elements are coupled to the main radiating element so that band-

width enhancement can be achieved. In comparison, the hybrid

antenna technique can also enhance the bandwidth by combining

different resonant modes excited from two entirely different

antenna types. Notably, one of the main advantages of such

design is the ability to control the two antenna types separately,

so that easy adjustment of all the excited modes can be achieved

[14]. From Refs. 12 and 13, both hybrid antennas are mainly a

combination of ‘‘monopole slot antenna’’ and other antenna

types are such as monopole type [12] and loop type [13] with

bending structures. Although Ref. 14 is a low-profile planar type

that allows GSM/DCS/PCS/UMTS/WLAN operations, it is also

a combination of monopole slot antenna and a T-monopole

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 54, No. 6, June 2012 1543

antenna coupled to an open-loop back coupling strip. Note that

this monopole slot antenna type requires a certain region to be

reserved from the system ground plane (so that the open-end

slot can be loaded), thus, such antenna type might not be feasi-

ble if only a small ground plane is available, or the entire sys-

tem ground plane is to be used by the RF circuit.

In this article, two promising antenna types that altogether

excited six usable resonant modes are integrated to form two

operating bands fL and fH, which allows GSM and DCS/PCS/

UMTS/WLAN operation, respectively. The first antenna is a pla-

nar inverted-F antenna (PIFA) type, and its top radiating ele-

ment is comprised of three narrow printed folding strips. As for

the second antenna, it is a simple monopole type with two

printed folding arms of dissimilar length. Details of the pro-

posed hybrid antenna designs are described and typical experi-

mental results are presented and discussed.

2. ANTENNA CONFIGURATION

The proposed hybrid antenna is a two layer structure as depicted

in Figure 1. The size of the ground plane is 70 � 40 mm2, and

a 50 ohms SMA with radius 0.65 mm (located exactly along the

mid point of the x-axis) is used to feed both the upper and lower

antennas as presented in Figure 2. Notably, both antenna types

are printed on a 0.8 mm FR4 substrate with er ¼ 4.4 and loss

tangent ¼ 0.02. Here, locating 5 mm above the bottom layer

antenna is the PIFA type with dimension 40 � 17 mm2 as pre-

sented in Figure 2(a). In this figure, two narrow strips (strip 1

and 2) with dissimilar length and folding structure are extended

from the shorting pin located at the bottom right corner of the

PIFA while a third strip (strip 3) with wider width is also

extended with a folding structure. Note that the open-ended hor-

izontal section of strip 3 is with dimension L ¼ 35 mm and W¼ 4 mm. From this PIFA structure, the two narrow strips will

excite two separate resonant modes at 0.88 and 0.97 GHz, which

in turn forms the lower frequency band fL and satisfy the GSM

operating band. As for strip 3, it will excite a resonant mode at

1.18 GHz that will combine with the other three resonant modes

excited by the bottom layer monopole antenna type so as to

form the upper frequency band fH. To induce these three reso-

nant modes at 1.38, 1.76, and 2.45 GHz from the bottom

antenna type, the technique of extending two folding monopole

arms with dissimilar length is used as depicted in Figure 2(b).

Here, the total dimension of this monopole antenna type is 40 �15 mm2, which includes a vertical feeding line with 1.5 mm

width connecting the antenna to the feed point.

3. RESULTS AND DISCUSSION

The measured and simulated return losses of the proposed

hybrid antenna are depicted in Figure 3. In this figure, good

agreements are exhibited between both measured and simulated

results, and the slight discrepancies are mainly attributed by the

SMA connector and the unexpected tolerance during the process

of fabricating the proposed antenna. Here, the impedance band-

width of both fL and fH measured along VSWR 3:1 is 15 and

78.3%, respectively, centered at 0.93 and 1.89 GHz. To further

discern the development of this proposed hybrid antenna, simu-

lated return losses for each antenna type (when the other is

absent) are presented in Figure 4. Note that all simulations

described here are performed by using the commercially avail-

able software high frequency structure simulator. In Figure 4, it

is obvious that if the bottom layer monopole antenna is removed

from the proposed antenna, the stand-alone top layer PIFA type

(Ref. 1) will excite three resonant modes f1, f2, and f3 at around

0.9, 1.0, and 1.3 GHz, respectively, whereby f1 and f2 will

Figure 1 Geometry of the proposed hybrid antenna (unit: mm)

Figure 2 Detail dimensions of (a) PIFA type (upper layer) and (b) monopole type (lower layer)

Figure 3 Measured and simulated return losses of proposed antenna.

[Color figure can be viewed in the online issue, which is available at

wileyonlinelibrary.com]

1544 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 54, No. 6, June 2012 DOI 10.1002/mop

combine together to form the GSM 900 operating band. If the

top layer PIFA is removed from the proposed antenna, the

stand-alone bottom layer monopole antenna (Ref. 2) will induce

another three resonant modes f4, f5, and f6 at around 1.4, 1.95,

and 2.4 GHz, respectively. Although the impedance matching of

these resonant modes (f4–f6) excited by Ref. 2 are not as desira-

ble as f1–f3 as exhibited in Figure 4, however, when the two ref-

erence antenna types are combined as a hybrid type, the imped-

ance matching of these three modes are greatly improved.

Furthermore, in Figure 4, it is obvious that f4 and f5 are respon-

sible for the DCS/PCS operation while f6 is mainly designated

for WLAN 2.4 GHz operation. Note that the current distribution

paths for f5 and f6 are due to the quarter-wavelength distribution

approximately along paths AC and AB, respectively, as shown

in Figure 2(b) while the current distribution path for f4 is due to

the half-wavelength distribution approximately along the path

BC. As the structure for bottom monopole antenna type are sim-

ple and easy to comprehend, hence, for brevity, the parametric

studies on the various vital parameters that will influence the

performances (f4–f6) of this antenna type will not be further

discussed.

To further comprehend the influence of various vital parame-

ters that will affect f1–f3, parametric studies via simulation are

carried out to provide further details and guideline for antenna

optimization. To discern the individual effect of both strips 1

and 2, the simulated return losses of the proposed antenna, and

the proposed antenna when strip 1 or 2 is removed one at a time

are illustrated in Figure 5. Here, when strip 2 is removed (only

strip 1), it is obvious that a resonant mode at around 0.91 GHz

is excited alone at around the 900 MHz band. If strip 1 is

removed instead (only strip 2), a resonant mode is induced at

Figure 4 Simulated return losses of Refs. 1 and 2 and proposed

antenna. [Color figure can be viewed in the online issue, which is avail-

able at wileyonlinelibrary.com]

Figure 5 Simulated return losses of proposed antenna when either

strips 1 or 2 is present. [Color figure can be viewed in the online issue,

which is available at wileyonlinelibrary.com]

Figure 6 Simulated return losses when strip 3 is removed. [Color

figure can be viewed in the online issue, which is available at


Figure 7 Simulated return losses when tuning parameter L. [Color



Figure 8 Simulated return losses when tuning parameter W. [Color




around 0.99 GHz. Thus, when both strips are loaded into the

PIFA type, the two resonant modes as aforementioned will

somehow shift slightly to the lower frequency band and inte-

grated to become a wide operating band fL that allow GSM

operation. Note that this phenomenon may be due to the cou-

pling effects between the two strips that allow slow wave effects

to take place for both resonant modes.

Figure 6 shows the effects of removing strip 3 from the PIFA

type, which demonstrated a missing f3 when compared to the pro-

posed antenna. Thus, this result has further proven that f3 is

highly dependent on strip 3. However, even though f4–f6 remains

fairly constant when strip 3 is absent, both f1 and f2 are deviated

to the higher frequency band while undesirable impedance match-

ing is also observed for f1. Therefore, the vital parameters of strip

3, L and W are further investigated as shown in Figures 7 and 8,

respectively. In Figure 7, a decrease in parameter L from 35 to

33 mm will result in shifting both f1 and f2 to the higher fre-

quency band whereas an increase in L from 35 to 37 will only

Figure 9 Normalized radiation patterns centered at GSM band

Figure 10 Normalized radiation patterns centered at DCS band

Figure 11 Normalized radiation patterns centered at PCS band

Figure 12 Normalized radiation patterns centered at UMTS band

Figure 13 Normalized radiation patterns centered at WLAN band


shift f2 to the lower band, which led to overlapping of both f1 andf2. As depicted in Figure 8, a slight decrease in parameter Wfrom 4 to 3 mm will have an adverse effect on the impedance

matching of both f1 and f2. In contrast, a slight increase in Wfrom 4 to 5 mm will aid in improving the impedance matching

for both f1 and f2, while both frequencies are also being shifted to

the higher frequency band. Thus, it is obvious that the impedance

matching of both f1 and f2 are also highly dependent on strip 3,

and it is highly suspected that besides inducing f3, strip 3 is also

acting as a tuning stub for both f1 and f2.The measured radiation patterns in three principal planes,

namely, the x–z, y–z, and x–y planes, for GSM, DCS, PCS,

UMTS, and WLAN operating bands are plotted, respectively,

from Figures 9 to 13, whereby all radiation patterns are meas-

ured at their respective center frequency. Here, except for the

GSM band that demonstrated near omnidirectional patterns in

the x–z plane, multiple nulls (or dips) and variation in the pat-

terns are observed for the rest of the operating bands. Figure

14 presents the measured peak gain and efficiency variation

diagram of the proposed antenna for all bands of interest. As

the excitation of GSM band is mainly attributed by the two

coupled narrow strips (strip 1 and 2) with small aperture size,

hence, a low gain variation between �2 and 0 dBi is observed

in the GSM band with efficiency that varies between 33 and

42%. As for the DCS/PCS/UMTS bands, a gain variation

between 3.4 and 5.4 dBi is measured while exhibiting an effi-

ciency that varies between 81 and 87%. Note that similar effi-

ciency between 80 and 84% is also measured across the

WLAN band, which demonstrated a stable gain variation

between 4.5 and 5.4 dBi.

4. CONCLUSIONS

A hybrid antenna with double layer structures is proposed and

successfully implemented. Parametric studies of this proposed

design were performed via simulation, and the influences on all

the resonant modes that form the lower and upper operating

bands, which cover the GSM and DCS/PCS/UMTS/WLAN

bands, respectively, have been investigated in detail. The meas-

ured results show satisfying impedance bandwidth, stable gain,

and efficiency across the bands of interest.

ACKNOWLEDGMENTS

This work is partially supported by the National Science Council of

Taiwan, ROC, project number NSC 99-2221-E-035-023. The

authors would like to thank Mr. Po-Chun Cheng for his advice in

designing this antenna.

REFERENCES

1. K.L. Wong and Y.C. Lin, Thin internal planar antenna for GSM/

DCS/PCS/UMTS operation in a PDA phone, Microwave Opt Tech-

nol Lett 47 (2005), 423–426.

2. K.L. Wong and C.H. Huang, Printed loop antenna with a perpendic-

ular feed for penta-band mobile phone application, IEEE Trans

Antennas Propag 56 (2008), 2138–2141.

3. C.I. Lin and K.L. Wong, Internal multiband loop antenna for GSM/

DCS/PCS/UMTS operation in the small-size mobile device, Micro-

wave Opt Technol Lett 50 (2008), 1279–1285.

4. Y.W. Chi and K.L. Wong, Very small-size printed loop antenna for

GSM/DCS/PCS/UMTS operation in the mobile phone, Microwave

Opt Technol Lett 51 (2009), 184–192.

5. Y.W. Chi and K.L. Wong, Quarter-wavelength printed loop antenna

with an internal printed matching circuit for GSM/DCS/PCS/UMTS

operation in the mobile phone, IEEE Trans Antennas Propag 57

(2009), 2541–2547.

6. H. Rhyu, J.H. Jo, F.J. Harackiewicz, and B. Lee, Multiband internal

antenna using two layer shorted patches for mobile handsets, Micro-


Figure 14 Measured radiation efficiency and peak gain variation of all the operating bands


7. R.A. Bhatti, Y.T. Im, N.N. Anh, and S.O. Park, Multiband internal

monopole antenna for mobile phones, Microwave Opt Technol Lett

51 (2009), 739–742.

8. C.T. Lee and K.L. Wong, Uniplanar coupled-fed printed PIFA for

WWAN/WLAN operation in the mobile phone, Microwave Opt

Technol Lett 51 (2009), 1250–1257.

9. C.C. Chen, C.Y.D. Sim, and F.S. Chen, Novel compact quad-band

narrow strip-loaded printed monopole antenna, IEEE Antennas

Wireless Propag Lett 8 (2009), 974–976.

10. K.S. Yoon, S. B. Park, S. Min Kim, and W.G. Yang, Penta-band

internal antenna for mobile handset applications using

parasitic element, Microwave Opt Technol Lett 50 (2008), 3045–3048.

11. P. Ciais, R. Staraj, G. Kossiavas, and C. Luxey, Design of an inter-

nal quad-band antenna for mobile phones, IEEE Microwave Wire-

less Compon Lett 14 (2004), 148–150.

12. C.I. Lin and K.L. Wong, Internal hybrid antenna for multiband operation

in the mobile phone, Microwave Opt Technol Lett 50 (2008), 38–42.

13. C.H. Wu and K.L. Wong, Internal hybrid loop/monopole slot

antenna for quad-band operation in the mobile phone, Microwave

Opt Technol Lett 50 (2008), 795–801.

14. C.Y.D. Sim, P.C. Cheng, and C.H. Lee, Low-profile multiband pla-

nar hybrid antenna design for mobile handset applications, Micro-



COMMENTS ON ‘‘A TRIANGULARDIELECTRIC RESONATOR ANTENNAEXCITED BY A COAXIAL PROBE’’

Sudipta Maity and Bhaskar GuptaDepartment of Electronics and Tele-Communication Engineering,Jadavpur University, Kolkata 700032, India

Received 19 September 2011

ABSTRACT: Original article published Microwave Opt Technol Lett

30:340–341, 2001. VC 2012 Wiley Periodicals, Inc. Microwave Opt

Technol Lett 54:1548, 2012; View this article online at

wileyonlinelibrary.com. DOI 10.1002/mop.26808

Key words: dielectric resonator antenna; transcendental equation

In the above communication [1], Ahmed A. Kishk has given

an approximate expression to compute the resonant frequency of

an equilateral triangular dielectric resonator antenna (DRA) with

height h and an equi-triangular uniform cross section with a side

length Ld, placed on a ground plane. He has given a transcen-

dental equation for computing the wave number kz in the zdirection as given below:

kz tan kzh� p=2ð Þ ¼ er � 1ð Þ � kz½ �1=2 (1)

where er is the relative permittivity of DRA. Hence, the reso-

nance frequency can be found by using the following expres-

sion:

fmn ¼ c

2pffiffiffiffier

p 4p3Ld

� �2

m2 þ mnþ n2� �þ k2z

" #1=2

(2)

where c is the velocity of light in free space. He has used the

resonance frequency index mn instead of mnl because

l ¼ mþ nð Þ (3)

But by applying image theory to remove the ground plane at

z ¼ 0 and enforcing continuity of tangential field to the surface

of dielectric resonator at z ¼ 6h, for TM to z mode, the follow-

ing transcendental equation will result

kz tan kzh� p=2ð Þ ¼ er � 1ð Þk20 � k2z� �1=2

(4)

where k0 denotes the free-space wave number corresponding to

the resonant frequency. Equation (4) satisfies all the data of col-

umn 2 of Table 1 in that communication [1]. It might have been

a typographical error. Hence, Eq. (1) of the above-mentioned ar-

ticle [1] must be corrected or replaced by Eq. (4).

Now, the DRA has triangular cross section. Hence, we need

triangular co-ordinate of a point which can be found by trilinear

transformation. For equilateral triangular cross section, we have

to use homogeneous trilinear coordinates [2] which satisfy the

relation:

lþ mþ n ¼ 0 (5)

Hence,

l ¼ � mþ nð Þ (6)

Similarly, the expression for third index (l) in that article [1]

must be corrected or replaced by Eq. (6).

REFERENCES

1. A.A. Kishk, Microwave Opt Technol Lett 30 (2001), 340–341.

2. S.A. Schelkunoff, Electromagnetic waves, Van Nostrand, New

York, 1943, pp. 393–396.



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A multiband hybrid antenna with double layer structures for mobile devices