6952 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 65, NO. 12, DECEMBER 2017

A 24 GHz Dual-Polarized and RobustDielectric Rod Antenna

Michael Sporer , Member, IEEE, Robert Weigel, Fellow, IEEE, and Alexander Koelpin, Senior Member, IEEE

Abstract— This paper presents a robust and easy manufac-turable dual linear-polarized dielectric waveguide antenna (DRA)for the 24 GHz industrial, scientific, and medical band. Dual-polarized operation is achieved by utilizing a square metalwaveguide and a conical horn for exciting the surface waveon a tapered dielectric rod. In order to couple a signal intothe square waveguide, an innovative dual-polarized, aperture-coupled microstrip-to-waveguide transition is shown. Addition-ally, two kinds of radomes, i.e., a foam radome and a half-wavewall monolithic radome are presented to protect the DRA.The antenna achieves a 10 dB bandwidth of 1.7 GHz and again of 11.5 dB including losses caused by the transitions andconnectors. The sidelobe level and the cross-polarization levelare lower than −25 and −26 dB, respectively. Designs andmeasurement results of all components are described in detailin this paper.

Index Terms— Antenna feeds, dielectric antennas, dielectricwaveguides, directional antennas, polarization.

I. INTRODUCTION

D IELECTRIC rod antennas (DRAs) are of notable inter-est especially in millimeter-wave applications where

mechanical robustness and focused radiation in endfire direc-tion is required, e.g., in antenna arrays due to space lim-itations [1]. Further advantages are low losses, ease offabrication, sealing of the feed system, and the possibility forthe antenna to be easily and stably mounted onto a panel.

The research on dielectric waveguides and rod antennasdates back to 1910 when Hondros and Debey [2] theoreticallyshowed that it is possible for electromagnetic waves to beguided by dielectric wires. It was determined that outsidethe wire the field decreases exponentially with increasingdistance and the thinner the wire the faster it declines. Theyalso realized that the phase velocity of the guided wavec depends on the relative permittivity εr and the diameterof the wire. For thin wires, c is almost equal to the freespace velocity c0 since the field expands far into air. Forthick wires, the electromagnetic energy is concentrated largelyinside. Hence, c is mostly determined by εr and approaches

Manuscript received April 3, 2017; revised September 20, 2017; acceptedOctober 1, 2017. Date of publication October 20, 2017; date of current versionNovember 30, 2017. This work was supported by the Bavarian Ministry ofEconomic Affairs and Media, Energy and Technology (StMWi), Munich,Germany, under Project MST-1405-0004//BAY195/003. (Correspondingauthor: Michael Sporer.)

The authors are with the Friedrich-Alexander University ofErlangen–Nuremberg, 91054 Erlangen, Germany (e-mail: michael.sporer@fau.de).

Color versions of one or more of the figures in this paper are availableonline at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TAP.2017.2764530

Fig. 1. Presented DRA with dual-polarized microstrip-to-waveguidetransition, horn-launcher, and tapered dielectric rod.

the characteristic velocity of the material c0/√

εr . For thesereasons, the diameter of a DRA is usually large at the feedpoint because of the higher excitation efficiency and decreasestoward the end in order to smoothly convert the bound fieldinto an unguided plane wave (see Fig. 1). This theoretical workwas expanded in [3] and experimentally confirmed in [4].

Pioneering theoretical and experimental work in the field ofDRAs was done by Mallach [5] and Halliday and Kiely [6].Mallach [5] showed that for good directivity the guided wave-length λg should be larger than 0.8λ0, with λ0 being thevacuum wavelength at the same frequency, and he presentsformulas for an optimal rod diameter and cross-sectional area,respectively. Furthermore, he noticed that both the directivityand the sidelobe level increase with increasing length, but thelatter can be reduced by decreasing the diameter of the rod.For this reason, he proposed and experimentally confirmedthat a slight tapering of the rod toward the end considerablyimproves the radiation pattern.

Radiation occurs at each discontinuity, i.e., the feed end,the end of the rod, and possible tapering along the rod.Since the superposition of these sources must be consid-ered, predicting the radiation pattern is complex and paperand pencil methods are approximations only. Work regardingthe radiation mechanism and optimization of the taper pro-file has been done by Mueller and Tyrrell [7], Neumann [8],and James [9], [10]. Experimental results regarding differenttapers are shown in [11]. Formulas, feeding techniques, anddesign guidelines for maximum gain and minimum sidelobelevel are given (see [12]–[14]).

0018-926X © 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.

SPORER et al.: 24 GHz DUAL-POLARIZED AND ROBUST DRA 6953

Fig. 2. Waveguide-to-DR transition. Dimensions: a = 5, b = 11, D = 18,D f = 25, w = 8, Lwg = 35, L1 = 15, L2 = 13, and LDRA = 69(dimensions in mm). Square waveguide made of copper, conical horn madeof brass, and dielectric rod made of PTFE.

Today, the radiation pattern can be computed numericallywith high accuracy by means of powerful field simulationsoftware and computing hardware. Nevertheless, the radiationmechanism of dielectric rods, optimum taper profiles, andfeeding structures are still current research topics [15]–[18].

This paper presents a dual linear-polarized DRA for appli-cations in the 24 GHz industrial, scientific, and medical (ISM)band fed by a metallic waveguide of quadratic cross section(see Fig. 1). Although a dual-polarized operation of DRAs isgenerally possible if axial-symmetric waveguides are used, lit-tle investigations have been done hereto [19], mostly focusingon circular polarized operation [20], [21]. Therefore, in thispaper, we mainly focus on the design and the evaluation ofappropriate feed and excitation methods for dual-polarizedoperation. Care has been taken for the designs to be robust,easy machinable, and cost-efficient.

Section II describes the design of the antenna consisting of:1) a transition from square waveguide to dielectric rod using acircular horn in order to excite the surface wave; 2) a new kindof dual-polarized, aperture-coupled, and shielded microstripto square waveguide transition; and 3) two kinds of radomeshave been designed in order to protect the antenna from theenvironment. Section III discusses measurement results of thefabricated antenna and its components.

II. DESIGN

A. Transition From Square Waveguide to Dielectric Rod

Unguided radiation by the feed end usually increases thesidelobe level of a DRA and therefore should be minimized byan appropriate feeding method. A surface wave can be excitedin different ways [14]. Here, a square metallic waveguideterminated by a horn is used since it provides dual polarizationcapability (Fig. 2).

The inner side length w is chosen so that only the twofundamental modes can propagate at the design frequency,i.e., the two degenerate modes TE10 and TE01 which onlydiffer in polarization. A portion of a tapered dielectric rodis inserted into the waveguide and the horn, respectively.This allows a smooth conversion between the TE10 (orTE01) mode and the corresponding HEx

11 (or HEy11) mode

of the dielectric waveguide. The taper performs broadbandimpedance matching between the hollow and the dielectric-filled waveguide sections. Regarding the rod diameter at thefeed end dmax, we follow the proposal of Mallach [5] who

Fig. 3. Guidance characteristics of the HE11 mode on a cylindrical rodmade of PTFE (εr = 2.1). For electrical thick wires (d � λ0), c convergesthe characteristic velocity of the material c0/

√εr thus λg → λ0/

√εr .

states that for optimal radiation characteristics, λg/λ0 alongthe rod should be within approximately 0.8 . . . 1.0. This rangecan be mapped to a physical diameter by examining theguidance characteristics of the HE11 mode for a cylindricalrod made of Polytetrafluoroethylene (PTFE) as shown inFig. 3. The curve was calculated by applying the guidancecondition for a dielectric rod formulated by Carson et al. [3]and from [22]. PTFE has been chosen because it turned outthat its permittivity εr = 2.1 allows dmax to be equal tothe inner side length of the waveguide (dmax = w) whileobtaining λg = 0.84λ0 at the feed end. In this way therod is mechanically supported and a stable mounting of theantenna is simplified (see Fig. 2). Further advantages are goodavailability and low dielectric losses.

A slight exponential taper guarantees good impedancematching to free-space and reduces the antenna sidelobe lev-els [16]. Therefore, our dielectric rod is tapered exponentiallyfrom dmax = w = 8 mm (λg = 0.84λ0) at the feed todmin = 2.5 mm (λg ≈ λ0) at the end.

For manufacturing reasons, the horn used to excite thesurface wave is not quadratically but conically shaped andattached to the square waveguide as illustrated in Fig. 2. In thisway the horn furthermore seals and protects the hollow squarewaveguide.

In order to minimize unguided radiation from the feed,the flare angle and the length of the horn were optimizedby means of field simulation software similar to [16]. Carehas been taken to limit the size of the feed, i.e., the apertureof the horn while still maintaining good excitation efficiency.Fig. 4 shows a snapshot of the electrical field distribution ofthe optimized transition. The good suppression of unguidedradiation at the feed can clearly be observed as well as thewave propagation along the structure.

To further reduce spurious radiation and reflections due todiscontinuities, the feeding mechanism is divided into twoparts with tapered transitions in between. At the feed of thedielectric rod, the HEx/y

11 mode should be bound inside thedielectric waveguide. Therefore, the TE10/01 waveguide modefirst enters a dielectric-filled waveguide section. The field isnow mostly concentrated inside the dielectric and can easilybe decoupled from the metallic waveguide by flaring thewaveguide walls. In this way, the waveguide mode is converted

6954 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 65, NO. 12, DECEMBER 2017

Fig. 4. Simulated instantaneous field distribution (amplitude of electricalfield). Transition from hollow into dielectric-filled metallic waveguide andinto dielectric waveguide.

Fig. 5. Photograph of the fabricated tapered dielectric rod inside the conicalhorn launcher.

into the HEx/y11 mode. By a slight tapering of the dielectric

rod, the field can in turn smoothly decouple from the rod andradiate into air. Fig. 5 shows a photograph of the fabricateddielectric rod. Tapped holes on the backside of the conicalsection facilitate the antenna to be mounted on, e.g., a panel.

As stated above, we have used a square metallic waveguideand a conical horn to feed the dielectric rod (Fig. 2). However,it is also possible to use a circular metallic waveguide insteadof a square one due to the similar field configuration insideboth kinds of waveguides. This solution would avoid reflec-tions and losses at the otherwise existing junction betweenthe square and the circular section. Key requirements of thisantenna are mechanical robustness and ease of fabrication.This is most easily achieved if the inner diameter of the usedwaveguide is identical to the diameter of the dielectric rod atthe feed end (w = 8 mm) so that the waveguide mechanicallysupports the dielectric rod. Regarding a hypothetical circularwaveguide, this would lead to cutoff frequencies of the fun-damental TE11 mode and the next higher order TM01 modeof 22.0 and 28.7 GHz, respectively [14]. Considering a safetymargin, especially for the TE11 mode to be well above cutoffof usually 10–20%, the usable frequency range is outside the24 GHz ISM band which makes this solution unsuitable.

In contrast, the frequency range of the used square wave-guide limited by the cutoff frequencies is between 18.8 GHz(TE10,01) and 26.5 GHz (TE11, TM11) [14]. The 24 GHzISM band is well inside this range. Furthermore, possibleslight irregularities and eccentricities in the waveguide causedby fabrication tolerances lead to crosstalk between the twopolarizations [23]. We believe that this effect is lower insquare waveguides due to less axial symmetry and moredefined polarization planes. Apart from this, the influenceof the coupling joint between the square and the circularsection has been shown to be low since most of the fieldis concentrated inside the dielectric rod, i.e., the axis of thewaveguide (see Section III-B).

Fig. 6. Exploded view of microstrip-to-waveguide transition. Vias are shownas black circles, relative layer thicknesses due to visualization not in scale.Dimensions: square waveguide inner/outer side length: 8.0/10 mm, for otherdimensions see Fig. 7.

For this reason, the mode patterns within the transitionregion from waveguide to dielectric rod are not plain TE10,01modes but distorted, typically hybrid versions thereof. Due tothe increased permittivity the corresponding cutoff frequen-cies are reduced and strongly depend on the shape and thepermittivity of the dielectric [24]. A computer-aided modalanalysis showed that the cutoff frequency of both fundamentaldegenerate modes is 13.2 GHz. The next two higher modescorrespond to distorted versions of the TM11 and the TE11mode and are also capable of propagation. However, due to theshort and axial-symmetrical dielectric-filled section, excitationof these modes is assumed to be low [24]. Furthermore,the dielectric rod only supports the fundamental HE11 mode(cutoff frequency of next higher mode: 27.4 GHz [25]) whichprevents higher order modes from radiating and thus fromaffecting the radiation characteristic. The measured radiationpattern confirms this (see Section III-D).

B. Dual–Polarized Microstrip–to–Waveguide Transition

The square waveguide is fed by a planar, dual linearpolarized microstrip-line to waveguide transition. This allowsintegration of further circuit components if required and can befabricated in a conventional multilayer PCB process withoutextra machining steps. For stability reasons a symmetric layerstack is used.

As shown in Figs. 6 and 7, the transition consists of asquare patch on the top layer, being aperture-coupled throughtwo orthogonal H-shaped slots on the third layer which areexcited each by a 50 � open-ended microstrip line (MSL) onthe bottom layer. In order to increase the isolation betweenthe feeds the open end of each MSL is bent. The secondlayer is free from metal in the transition area. The centeredpatch and the slots are embedded in a cavity surrounded byvias. The square waveguide is directly soldered to the PCBso that the vias virtually lengthen the waveguide into thePCB and suppress the propagation of surface waves. Theground plane containing the slots terminates the waveguideand acts as a reflector for the patch thus eliminating the

SPORER et al.: 24 GHz DUAL-POLARIZED AND ROBUST DRA 6955

Fig. 7. Bottom view of microstrip-to-waveguide PCB. Vias are shownas black circles. Dimensions: w = 8.0, lp = 2.51, l1 = 1.5, l2 = 1.0,l3 = 1.4, l4 = 0.7, l5 = l6 = 0.8, d = 1.8, and w f = 0.5, slot width is 0.2(all dimensions in mm).

need for a separate back-short block. Additionally, it achievesisolation of the radiator and its feed. This allows both to beoptimized independently and furthermore provides room forother components to be placed beneath the transition.

Since a waveguide wall feed-through is not necessarythe whole transition is electrically shielded and stray fieldsare inherently suppressed. This is especially useful inhigh dynamic range transmit/receive systems where TX/RXcrosstalk must be minimized. The relative large distancebetween patch and slots due to the inner FR4 core increasesthe bandwidth [26].

An advantage of aperture coupling is the freedom to adjustthe shape, size, and position of the slots. These parametersmust be carefully tuned to ensure good impedance matchingand coupling efficiency. Here, H-shaped apertures are used,i.e., two rectangular apertures each loaded by four shortslots. Due to the loading, the transverse field becomes nearlyuniform along the aperture and thus enhances the couplingbetween MSL and patch [26]. This in turn allows the sizeof the apertures to be reduced and both to be placed underthe patch for good coupling efficiency while still maintaininghigh isolation between them. By shortening the rectangularapertures the resonance frequency of the patch increases [26].This effect facilitates the reduction of the patch size. Addition-ally, the cavity below, and the waveguide walls above the patchlead to an increased capacitive loading. Therefore, the size ofthe patch can be further reduced to less than half a guidedwavelength which allows the patch to fit into the cross sectionof the waveguide. A photograph of the fabricated transition isshown in Fig. 8.

C. Radome

Since the antenna should be operated outdoors, a radome isrequired in order to protect the antenna from the environment.The radome must be electrical transparent at the design fre-quency and exhibit low dielectric losses for not degrading the

Fig. 8. Photograph of the dual–polarized and planar microstrip–to–waveguidetransition with attached connectors.

Fig. 9. Cross section of foam radome A (left) and half-wave wallradome B (right). Dimensions: a1 = 50 mm, b1 = 18 mm, h1 = 85 mm,a2 = 55 mm, b2 = 48.4 mm, h2 = 75 mm, and d1 = 3.3 mm.

performance of the antenna. Additionally, mechanical stabilityand easy fabrication are demanded.

Two kinds of radomes have been designed (see Fig. 9).Radome A is made of a relatively thick closed-cell rigid foambased on polymethacrylimide. The material was chosen mainlybecause of its low weight, low cost, and easy machinability.The dielectric properties are given to be εr,A = 1.05 andtan(δ)A = 0.01 at 26.5 GHz [27]. For the design, a cylindricalbody with hemispheric cover was chosen since it guaranteessmall incident angles and thus good transmission through thewalls. It furthermore shows good aerodynamic properties andinherently minimizes boresight error due to its axial symmetry.It was originally thought that the influence of the materialcan be neglected due to the low permittivity. Therefore,the shape of radome A was not further optimized. However,measurements showed unexpected high transmission losses(see Fig. 15). For this reason, a second radome (radome B) wasfabricated based on a half-wave wall monolithic design madeof epoxy resin [εr,B ≈ 3.5 and tan(δ)B ≈ 0.04 (empiricalvalues)]. The wall thickness can be computed as

d1 = c0

2√

εr,B · 24 G H z= 3.3 mm (1)

which also ensures sufficient mechanical stability. The radomewas fabricated in a stereolithography prototyping process.Again, a cylindrical body with hemispheric cover was cho-sen. The height of the cylindrical section and the radius ofthe hemispheric section were optimized for minimal side-lobe degradation with help of field simulation software.Fig. 10 shows a photograph of the radomes.

6956 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 65, NO. 12, DECEMBER 2017

Fig. 10. Photograph of fabricated radomes. Half-wave wall radome (B) madeof epoxy resin (left). Foam radome (A) made of closed-cell rigid foam basedon polymethacrylimide (right).

Fig. 11. Measured and simulated S–parameters of the dual–polarizedmicrostrip-to-waveguide transition (open–ended waveguide).

III. RESULTS

A. Dual–Polarized Microstrip–to–Waveguide Transition

Fig. 11 shows the simulated and measured S-parameters ofthe dual-polarized microstrip-to-waveguide transition in caseof an open-ended waveguide. The RF-connectors are calibratedout by means of a preceding thru, reflect, line calibration.

Within our frequency range of interest, i.e., the 24 GHzISM band, the measured return loss at both ports is higherthan 15 dB. The 10 dB bandwidth is approximately 1.5 GHz.The isolation between the ports is better than 32 dB indicatinga low cross-polarization. The different return losses and theslight shift between the resonant frequencies of ports 1 and 2are caused by the unequal coupling between the respective slotand the patch (see Fig. 6).

In order to evaluate the insertion loss of the transition andthe polarization characteristics, two identical transitions havebeen placed back-to-back in the same orientation. Fig. 12shows the measured S-parameters and Fig. 13 the correspond-ing measurement setup.

As for the open-ended configuration, the return losses arehigher than 15 dB in the 24 GHz ISM band. For the co-polarized ports the insertion loss is 2.7 dB, i.e., 1.35 dB for asingle transition, respectively. The phase difference betweenthe two co-polarized signals is lower than 4°. The lossesare mainly due to dielectric losses (especially the FR4 coreof the PCB) and the inherently nonideal electrical contactbetween the two open-ended square waveguides during the

Fig. 12. Measured S–parameters of two identical dual–polarized microstrip-to-waveguide transitions in back-to-back configuration. Ports 1 and 2 referto the first, and ports 3 and 4 to the second transition. Phase difference� S31 − � S42 is plotted on the right axis.

Fig. 13. Measurement setup of two equal microstrip–to-waveguide transitionsplaced back–to–back. The coupling joint between the waveguides is coveredby copper tape for mechanical stability and to ensure electrical contact duringthe measurements.

measurement (see Fig. 13). The cross–polarization level islower than −27 dB.

B. Transition From Square Waveguide to Dielectric Rod

In contrast to the previous section and Fig. 11, Fig. 14 showsthe measured S-parameters of the transition with mountedconical horn and dielectric rod. Additionally, a measurementwith conical horn only is plotted for comparison.

Obviously, the frequency response is strongly distorted ifthe conical horn is attached to the waveguide. This is causedby reflections at the coupling joint due to the different crosssection shapes of horn and waveguide. With inserted dielectricrod, these distortions vanish since the field is now morefocused inside the rod. As a result, the return loss increases to18 dB in comparison to the open-ended case due to the bettermatching. The isolation remains unchanged.

C. Radomes

Fig. 15 shows the simulated and measured radiation patternsof both radomes for scans in E- and H-planes. Additionally,

SPORER et al.: 24 GHz DUAL-POLARIZED AND ROBUST DRA 6957

Fig. 14. Measured S-parameters of the transition with horn only (horn) andboth, horn and rod (horn and rod).

Fig. 15. Radiation patterns of rod only (w/o radome) and its simulation (sim),with foam radome (A), and half-wave wall radome (B). H-plane (left) andE-plane (right).

the measured and simulated patterns of the rod without radomeare shown for comparison. The measured patterns are normal-ized to the peak gain of the antenna without radome. Thesimulated and measured patterns without radome are in goodagreement. Regarding the radomes, due to the symmetry of thedesign the boresight error is negligible for both radomes. Theradiation characteristics mainly differ with regard to insertionloss and sidelobe degradation. The insertion loss of the half-wave wall epoxy radome (B) is 0.8 dB and thereby 1.0 dBlower compared to the foam radome (A). This is due to therelative thick foam and thus the higher dielectric losses in mainradiation direction. The sidelobes increase by approximately3 dB in case of radome A, and by approximately 2 dB forradome B.

D. Radiation Pattern and Gain

The pattern and gain measurements have been conductedin an anechoic chamber while applying the gain-comparisonmethod [28]. A fixed transmitting rectangular horn antennaexcites a planer linear polarized field at the position of theantenna under test (AUT). By measuring the received powerPAUT of the AUT and PS of a linear polarized standard gainhorn placed at the same position as the AUT, the gain of the

Fig. 16. Co- and cross-polarization (co-pol, cross-pol) radiation patterns ofthe rod (without radome) for 23.5, 24.0, and 25.5 GHz (23.5, 24.0, 25.5).H-plane (left) and E-plane (right).

AUT GAUT can be computed as

GAUT = PAUT

PS· GS (2)

with GS being the gain of the standard gain horn. Sincethe antenna is fed at the coax connector, the measured gainsinclude the losses caused by the connector and the tran-sitions. Fig. 16 shows the resulting gain patterns of scansin E- and H-planes and for co- and cross-polarization atdifferent frequencies without radome.

The patterns show only minimal variations within themeasured frequency range of 23.5–24.5 GHz confirming thebroadband characteristics of DRAs and indicating usabilityfar beyond the intended 24 GHz ISM band. No sidelobesother than the simulated ones are visible which confirms thathigher modes do not appear or at least do not radiate. Thesidelobe level is approximately −25 dB for E- and H-planesand the cross-polarization level is lower than approximately−26 dB. It is similar to the measured cross-polarization levelof the microstrip–to–waveguide transition (see Fig. 12) statingthat the isotropic dielectric rod has no, or only minimaldepolarization effect.

The measured gain of the antenna (Fig. 1) is 11.5 dB.This includes all losses in the structure, especially connectorinsertion losses and losses caused by the transitions.

IV. CONCLUSION AND DISCUSSION

The intention of this paper was to develop a robust duallinear-polarized directive antenna for harsh environments inthe 24 GHz ISM band. A design based on a waveguide-fedDRA was chosen since it inherently provides hermetic sealingand thus protection from water and dirt. Furthermore, due tothe wide bandwidth of a DRA, this kind of antenna is robustagainst fabrication tolerances and temperature, long-term andaging effects. While most DRAs found in the literature are fedby a standard rectangular waveguide little work can be foundon dual-polarized DRAs. Therefore, this paper is focusedon the development of appropriate feeding and excitationtechniques for dual-polarized operation.

6958 IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 65, NO. 12, DECEMBER 2017

Most of the components presented here are based on alongitudinal symmetric design, which naturally ensures dual-polarization capability. These are a conical horn for excitinga surface wave on a conical dielectric waveguide, a squaremetallic waveguide, and two kinds of radomes. Additionally,an aperture-coupled microstrip-to-square waveguide transitionwas developed to excite a horizontal, a vertical, or bothpolarizations. The microstrip-to-square waveguide transitionis designed to cover and shield the waveguide end and inthis way provide further environmental and electrical isolationfrom the outside. The performance was measured in a back-to-back configuration showing low insertion losses of 1.35 dBand a low cross-polarization.

Since in our application the polarization is switched,the phase characteristic of the complete antenna has notbeen measured. However, due to symmetry reasons it canbe assumed that the phase shift in the metal and dielec-tric waveguide is identical for both polarizations. The phaseimbalance of the antenna thus depends almost solely on thephase imbalance of the microstrip-to-waveguide transition.It is lower than 4° and constant within approximately 0.5°in the 24 GHz ISM band. Therefore, also left- or right-handedcircular polarization is basically possible if both ports aresimultaneously fed with appropriate phase difference.

The 10 dB bandwidth of the whole antenna is 1.7 GHz.The radiation characteristic and the gain of the DRA weremeasured in an anechoic chamber with and without radomes.It was shown that the half-wave wall radome performs betterthan the radome made of thick closed-cell rigid foam and onlyminimally affects the radiation characteristic. The insertionloss is 0.8 dB. In general, the simulated and measured radiationpatterns are in good agreement. The antenna shows lowsidelobe levels of −25 dB as well as low cross-polarizationlevels of approximately −26 dB making it a good candidatefor, e.g., radar polarimetry in harsh environments.

ACKNOWLEDGMENT

The authors would like to thank the reviewers for theirvaluable comments and suggestions to improve the quality ofthis paper.

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[27] Evonik Industries AG. Technical Information Rohacell HF. Accessed:Apr. 3, 2017. [Online]. Available: http://www.rohacell.com

[28] W. L. Stutzman and G. A. Thiele, Antenna Theory and Design. Hoboken,NJ, USA: Wiley, 2012.

Michael Sporer (S’13–M’16) received the B.Sc.degree in electrical, electronics, and communicationengineering and the M.Sc. degree (Hons.) in systemsof information and multimedia technology from theFriedrich-Alexander University Erlangen–Nürnberg,Erlangen, Germany, in 2010 and 2012, respectively.

In 2013, he joined the Institute for Electron-ics Engineering, Friedrich-Alexander University ofErlangen–Nuremberg, as a Research Assistant.His current research interests include radar sig-nal processing, millimeter-wave antenna, and circuitdesign.

SPORER et al.: 24 GHz DUAL-POLARIZED AND ROBUST DRA 6959

Robert Weigel (S’88–M’89–SM’95–F’02) was bornin Ebermannstadt, Germany, in 1956. He receivedthe Dr.-Ing. and Dr.-Ing.habil. degrees in electricalengineering and computer science from the MunichUniversity of Technology, Munich, Germany, in1989 and 1992, respectively.

From 1982 to 1988, he was a Research Engineer,from 1988 to 1994, a Senior Research Engineer, andfrom 1994 to 1996, a Professor of RF circuits andsystems with the Munich University of Technology.From 1994 to 1995, he was a Guest Professor of

SAW technology with the Vienna University of Technology, Vienna, Austria.From 1996 to 2002, he was the Director with the Institute for Communicationsand Information Engineering, University of Linz, Linz, Austria. In 2000,he was appointed a Professor of RF Engineering with Tongji University,Shanghai, China. Since 2002, he has been the Head of the Institute for Elec-tronics Engineering, Friedrich-Alexander University of Erlangen–Nuremberg,Erlangen, Germany.

Alexander Koelpin (S’04–M’10–SM’16) receivedthe Diploma in electrical engineering, the Ph.D.degree, and the "venia legendi" from the Universityof Erlangen–Nuremberg (FAU), Erlangen, Germany,in 2005, 2010, and 2014, respectively.

From 2007 to 2010, he was a Team Leader, from2010 to 2015, a Group Leader of Circuits, Systems,and Hardware Test, and since 2015 he has been aLeader of the Electronic System Group with FAU.He is currently a Lecturing Professor and a ResearchGroup Leader with FAU, where he has been with the

Institute for Electronics Engineering since 2005. His current research interestsinclude microwave circuits and systems, wireless communication systems,local positioning, and six-port technology. He has authored or co-authoredmore than 180 publications in his areas of interest.

Dr. Koelpin serves as a reviewer for several journals and conferences.He is the Co-Chair of the IEEE MTT-S Technical Committee MTT-16,the Commission A: Electromagnetic Metrology of U.R.S.I., and since 2012 hehas been the Conference Co-Chair of the IEEE Topical Conference onWireless Sensors and Sensor Networks. In 2016, he was a recipient of theIEEE MTT-S Outstanding Young Engineer Award.