270 IEEE TRANSACTIONS ON MICROWAVE THEORY AND …mingyu/publications/p201102.pdf · 2013-11-23 · 270 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 59, NO. 2, FEBRUARY

270 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 59, NO. 2, FEBRUARY 2011

Enhanced Microwave Multiplexing NetworkMing Yu, Fellow, IEEE, and Ying Wang, Member, IEEE

Abstract—A new approach for designing a microwave mul-tiplexing network is presented in order to improve channelperformance. Through design and optimization of the connectingtransmission line structure, the new approach ensures that anextra pole is formed in the passband of each channel. The channelfilter order is therefore increased by one without additionalresonators. Significant improvement in the performance of themultiplexer is achieved without the penalty of increased size orweight of the hardware, which is extremely important for appli-cations in communication satellite. Simulation and measurementresults are used to demonstrate the feasibility of such enhancedmicrowave multiplexing networks.

Index Terms—Filter design, multiplexer design, multiplexingnetwork, optimization.

I. INTRODUCTION

M ICROWAVE multiplexing networks are widely used incommunication satellites to combine several frequency

bands for transmission via a common antenna [1], [2]. Such amultiplexer typically consists of a number of channel filters cou-pled to a common output via waveguide or coaxial transmissionlines.

Multiplexing network filter performance is particularly im-portant in satellite applications since an increase in the inser-tion loss of the channel filters in the microwave multiplexingnetwork results in a reduction of Effective Isotropic RadiatedPower (EIRP) emitted by the satellite and accordingly a re-duction in the amount of radio frequency (RF) transmissionpower. Rejection between channels also limits the transmissionof spectral re-growth from the power amplifiers that are con-nected at input of the filters. For space applications, a multi-plexer must comply with stringent RF in-band and out-of-bandcharacteristics, such as passband gain and group delay flatnessand stopband rejection, over a wide environmental temperaturerange while combining high power input signals [3]–[6]. Con-ventional design techniques achieve increased filter rejection byincreasing the order of a cross-coupled filter. Accordingly, theuse of a high order filter allows for the bandwidth of the channelfilter to be expanded since the extra poles provide extra rejec-tion. Overall this results in increased filter bandwidth. At the

Manuscript received July 05, 2010; revised September 24, 2010; acceptedSeptember 29, 2010. Date of publication November 01, 2010; date of currentversion February 16, 2011. This work was supported in part by Natural Sciencesand Engineering Research Council of Canada and in part by COM DEV Ltd.

M. Yu is with the COM DEV Ltd., Cambridge, ON, Canada N1R 7H6 (e-mail:[email protected]).

Y. Wang is with Faculty of Engineering and Applied Sciences, Universityof Ontario Institute of Technology, Oshawa, ON, Canada L1H 7K4 (e-mail:[email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TMTT.2010.2086471

Fig. 1. N-channel manifold-coupled multiplexer.

same time, satisfactory filter rejection is maintained. However,in order to do this, extra resonators are typically added to realizeadditional poles. This approach usually increases the weight andsize of the multiplexer, which is a significant drawback for ex-tremely weight sensitive satellite applications.

The virtually lossless direct connection of channels throughthe common transmission line and the large number of designvariables involved make the design of multiplexing networksdifficult. Interactions among channel filters need to be takeninto account in order to achieve channel performance close toa stand-alone filter [1], [2], [7]–[9]. In conventional design, themanifold serves as part of a matching network for each indi-vidual channel. Resonances generated by the interconnectingtransmission line are in general to be avoided during the designsince they cause unwanted peaks to appear in or near channelfilter passband.

In this paper, a new approach that can effectively harness themanifold resonances is presented and an enhanced microwavemultiplexing network results. It takes advantages of the con-necting transmission line structure to form additional resonatorsfor each channel filter [10]. The distance between channel filtersand the distance between each channel filter and the manifoldare determined to ensure that a pole is formed causing an ad-ditional real reflection zero to be brought into the passband ofthe microwave multiplexing network. The channel filter order istherefore increased by one without additional resonators. Sig-nificant improvement in multiplexer performance is achieved,as will be demonstrated using Ku band and C band multiplexerdesigns. Such improvement is crucial in view of the increas-ingly stringent requirement in the channel filter response andminimal available design margins. Moreover, the enhancementin performance is achieved without increase of size or weightof the hardware. The proposed method is applicable to all mul-tiplexers using transmission lines to connect channel filters.

II. ENHANCED MULTIPLEXING NETWORK

A conventional multiplexer configuration intended for spaceapplication is shown in Fig. 1. Channel filters are connectedto the manifold through waveguide E-plane or H-plane T-junc-tions. The manifold waveguide is short-circuited in one end andthe other end is the output common port.

0018-9480/$26.00 © 2010 IEEE

YU AND WANG: ENHANCED MICROWAVE MULTIPLEXING NETWORK 271

Fig. 2. (a) Series connection of two short-circuited waveguides and (b) itsequivalent circuit.

A. Waveguide T-Junction Equivalent Circuit

The proposed design approach is presented starting from asingle channel filter connected to a manifold, i.e., inFig. 1. An th order channel filter is attached to a waveguideT-junction with one port short-circuited. In order for the struc-ture to behave as an order filter, the T-junction will serveas the first resonator. The distance between channel filter 1 to themanifold and the distance between the channel filter to the shortcircuit are designed to obtain the desired resonance frequencyand the required coupling value. The channel filter is replacedwith a short circuit in the following analysis. A simplified equiv-alent circuit in Fig. 2(a) showing two short-circuited waveguidesections in series connection is used for demonstration, whichcan be realized using E-plane waveguide T-junction as will beshown later. For H-plane T-junction, the equivalent circuit be-comes a parallel connection of the two waveguide sections.

The impedance at the common port in Fig. 2(a) is

(1)

where is the characteristic impedance, is the propagationconstant, ignoring loss of the waveguide, and and are thelengths of the two waveguides, respectively. The short-circuitedwaveguide section with length of can be viewed as a couplingelement with susceptance . Resonance occurs in the vicinitywhere or , where and

is the guided wavelength at the resonance frequency. For thisstructure it is more convenient to find the coupling value and theresonance frequency using group delay [11]. This is acceptablesince only a narrow band filter is used for each channel. Thereflection coefficient at the common port can be found using (1)as

(2)

Therefore

(3)

The group delay is given by

(4)

Let and

(5)

Fig. 3. (a) Waveguide E-plane T-junction with two ports short-circuited. (b)The equivalent circuit of the E-plane T-junction with the reference planes de-fined in (a) [13].

Assume at resonance, which coincides with the centerfrequency of the channel filter. The group delay at resonancecan be derived as

(6)

When or is odd, let . Then. When or is even, let .

. Equation (6) is thus reduced to

(7)

The coupling value is related to group delay as [12]

(8)

where BW is the bandwidth of the channel filter. For requiredcoupling value , length of the short-circuited waveguide can becalculated using (7) and (8). Note that when , theelectrical length of the short-circuited waveguide section withlength of is . However when , the electricallength is . is subsequently calculated ensuring the correctresonance frequency, i.e.,

(9)

The structure in Fig. 2 can be realized using waveguideE-plane junction. The lengths of each waveguide section needto be corrected taking into account the junction effect. Theequivalent circuit of the T-junction [13] is used in Fig. 3, inwhich a waveguide E-plane junction with cross section area

is assumed.The input impedance at the common port and reference planeis changed to

(10)

in which

(11)


Equations for calculation of and are given in[13]. The reflection coefficient at the common port in (2) and(3) is modified accordingly as

(12)

(13)

It can be shown that the group delay at resonance can be esti-mated using

(14)

Let . For design purpose,can be calculated using (7) and (8) for required coupling

value . is then determined ensuring the correct resonancefrequency, i.e., . An accurate estimate of group delaycan then be found using (14). The waveguide lengths and

in Fig. 3 are subsequently obtained by correcting andusing (11).

Once the desired coupling value and resonance frequency aredetermined, the waveguide junction can be used as the first res-onator. The rest of the filter is designed following conventionalfilter design methodology [12].

B. Example of Single Channel Design

The waveguide with mm and mm(WR75) is used in the following example. As shown in Table I,lengths and are obtained using (7), (9) and (11) to realizethe desired group delay , in which the resonance frequency

GHz. In Table I(a), , i.e., , while inTable I(b), , i.e., . The group delay is then re-calculated using (14) to obtain in Table I. The equivalentcircuit of the E-plane T-junction in Fig. 3 is used for verifica-tion. The responses are plotted in Fig. 4 and the group delaysand resonance frequencies obtained from the data in Fig. 4 arelisted in the last two columns in Table I for comparison.

As can be seen, the group delay, hence coupling coefficient,can be readily adjusted varying lengths of the waveguides. Fur-thermore, (14) gives more accurate prediction of group delay.The difference in group delays and deviation of the resonancefrequencies from 12 GHz are due to the approximation in theassumption of at resonance. For the structure in Fig. 2(a)to resonate at

(15)

Solutions to (15) are complex and do not coincide with thosefrom (7) and (9) precisely. However such inaccuracy can beeasily corrected in practical use. The calculated waveguide sec-tion lengths serve as good initial values.

Using the initial values given in Table I(b), single channel fil-ters connected to waveguide E-plane T-junctions are designed.The T-junctions are first simulated using mode-matching basedfull electromagnetic (EM) simulator. The S parameters of the3-port waveguide T-junction networks are cascaded with circuit

Fig. 4. Group delay responses of the equivalent circuit of the E-plane T-junc-tion in Fig. 3. (a) the waveguide lengths are as case 1 to case 4 in Table I(a).(b) the waveguide lengths are as case 5 to case 8 in Table I(b).

TABLE IGROUP DELAY RESPONSE AS WAVEGUIDE LENGTHS CHANGE

model of channel filters using coupling matrices [14]. The wave-guide lengths as well as coupling coefficients of the channelfilter are optimized to ensure good impedance matching in thefilter passband [15].

Taking the case 7 in Table I(b) as the initial value, a four-pole filter centered at 12 GHz with 36 MHz bandwidth is at-tached to the waveguide T-junction. After optimization,


Fig. 5. Comparison of the near-band responses of the four-pole filter withwaveguide T-junction and a five-pole stand-alone filter.

TABLE IIOPTIMIZED WAVEGUIDE LENGTHS WITH CORRESPONDING GROUP DELAY

AND COUPLING VALUES FOR CHANNEL FILTERS CENTERED AT 12 GHZ AND

WITH 36 MHZ BANDWIDTH

mm and mm to achieve 15 ns group delay at12 GHz, which is equivalent to coupling value from(8). The difference between the optimized waveguide lengthsand the ones in Table I is largely caused by the difference inthe model based on EM simulated S parameters of the wave-guide junction and the model based on the equivalent circuit in[13]. Fig. 5 shows the common port return loss, i.e., , andthe transmission coefficient between the input of the channelfilter and the common port, i.e., , which clearly demon-strates that an extra pole is formed by the waveguide T-junctioncausing an additional real reflection zero to be brought into thepassband. Therefore, a five-pole response results although thechannel filter order is only four.

For comparison, the response of a five-pole stand-alone filtercentered at 12 GHz with 36 MHz bandwidth is also plotted inFig. 5. The coupling values of the five-pole filter are

, and input/output coupling. The near-band responses of the four-pole

filter with waveguide T-junction and the five-pole stand-alonefilter are identical.

In the following, all cases in Table I(b) are used for designof enhanced channel filters. The waveguide lengths after opti-mization, group delay at center frequency of 12 GHz and cor-responding coupling value for 36 MHz bandwidth are listedin Table II. The near-band responses of are compared inFig. 6(a). It is seen that as the coupling value increases, the im-provement in out-of-band rejection performance decreases. Al-though it is thus desirable to have small coupling value or largegroup delay, it becomes difficult to achieve good return loss as

Fig. 6. Comparison of responses of four-pole filters connected with waveguideT-junctions with different waveguide lengths. (a) Near-band responses of thetransmission coefficient �� . (b) Common port return loss �� . (c) out-of-band �� responses.

shown in Fig. 6(b) with the design case 8 demonstrating be-havior of a single-ended filter. Furthermore, it can be seen fromFig. 6 that improvement in the out-of-band rejection can still beachieved even though the return loss does not show five distinctpeaks in the design cases 5 and 6. The out-of-band rejection re-sponses in Fig. 6(c) show that the waveguide section with thelength of is causing an extracted pole at the frequency whereit becomes a short-circuited quarter-wavelength resonator.

C. Enhanced Multiplexer Design Process

The proposed method is extended to multi-channel design.Connecting transmission lines are designed so that proper


group delays, or coupling values, are achieved in the passbandof each channel resulting in extra reflection zeros. Takingadvantages of the existing waveguide manifold structure, thechannel filter order is therefore effectively increased withoutadditional resonators.

The general design steps include the following.1) The manifold is first designed with the channels filters re-

placed with short circuits. As will be shown, lengths ob-tained following these steps give good initial values whensingle-ended channel filters are later attached to the mani-fold. It is also due to the fact that each channel filter behavesas a short circuit in its out-of-band. The distance betweenchannel filters and the distance between each channel filterand the manifold are optimized to the desired group delayvalues in the passband of channel filters.

2) Check if the manifold dimensions are physically realizablein order to accommodate each channel filter. Repeat Step1 when necessary.

3) Attach single-ended channel filters to the manifold.4) All dimensions including channel to channel spacing,

channel filter to manifold spacing, and internal dimensionsof the channel filters are optimized to meet performancerequirement [1].

As demonstrated in the previous section, small coupling valueor large group delay results in higher level of performance en-hancement. However large group delays are in general difficultto obtain. Besides the fact that it is difficult to achieve good re-turn loss as shown in Fig. 6(b), the response becomes highlysensitive to changes in the manifold lengths for large group de-lays. It is found that as the number of channels increases thegroup delay should be designed to have a relatively low valueover the passband of the multiplexer for practical realization.The amount of realizable delay may be limited by factors suchas overall size constraint, the number of channels, center fre-quency of each channel and bandwidth of each channel. In mostpractical applications, a suitable solution can be found by op-timization. Moreover, selected channels may be designed withhigher level of performance enhancement.

D. Example of Non-Contiguous Multiplexer

To illustrate the effectiveness of the design process, a two-channel noncontiguous multiplexer is designed with the firstchannel filter centered at 12.75 GHz and the second channelfilter centered at 12.53 GHz, each with 72 MHz bandwidth.

Manifold waveguide lengths and are as shown inFig. 7, where and 2 for channel filter 1 and 2, respectively.Waveguide WR75 E-plane T-junctions are used. Comparison ofconventional design techniques [1], [7] with the proposed de-sign process has been conducted. Table III compares the valuesfor manifold waveguide lengths obtained using both methods.Fig. 8 shows the group delay response of the common port re-turn loss as the result of the optimization in step 1 of the de-sign process with the channel filters replaced with short circuits.Two resonances at the corresponding center frequencies of thechannel filters are clearly shown. The group delay response ofthe conventional design is also plotted in Fig. 8 for comparison.The common port return loss, i.e., , the transmission coef-ficients between port and the common port, i.e., , where

Fig. 7. Two-channel multiplexer.

TABLE IIIMANIFOLD LENGTHS FOR A TWO-CHANNEL NON-CONTIGUOUS MULTIPLEXER

OBTAINED FROM CONVENTIONAL DESIGN APPROACH AND THE

PROPOSED METHOD

Fig. 8. Group delay response of the common port return loss with thechannel filters replaced with short circuits for the two-channel noncontiguousmultiplexer.

is the port number and , are shown in Fig. 9. Com-paring the curve of Fig. 9(a) of the conventional design,to the curve of Fig. 9(b) of the enhanced multiplexing net-work, it can be seen that Fig. 9(b) is essentially a combinationof two five-pole filters whereas the conventional design is es-sentially a combination of two four-pole filters.

To clearly evidence the improvement in the performance ofthe each individual channel filter the transmission coefficientsof Fig. 9 have been isolated and are comparatively shown inFig. 10. For channel 1, in comparing the two curves in Fig. 10(a)it is obvious that the curve of the new design is flat over awider frequency range than the curve of the conventionaldesign, thus the channel filter 1 of the enhanced multiplexer hasa greater bandwidth. It also can be seen that this increased band-width is not achieved at the expense of rejection in the out ofband region as it can be seen that the curve of the enhanceddesign has a much steeper rejection rate. The transmission coef-ficients for channel 2 experience similar improvement as shownin Fig. 10(b). Table IV summarizes the flatness of the transmis-sion coefficients in the passband of each channel filter. The spec-


Fig. 9. Common port return loss, i.e., �� , the transmission coefficient be-tween port 2 and the common port, i.e., �� , and the transmission coefficientbetween port 3 and the common port, i.e., �� , of (a) the conventional designand (b) the enhanced multiplexing network.

ifications are typically given based on normalized values ofthat are normalized to the minimum value of in the pass-band. As can be seen from Table IV, for different flatness spec-ification, the improvements range from 5.2 MHz to 6.9 MHz,which represent 7% to 9% improvement of the useable band-width. Such improvements are significant for satellite commu-nication system, where less than 1 ppm/ C frequency stabilityis required [3].

The extracted coupling matrices of the two five-pole (equiv-alent) channel filters as shown in Fig. 9(b) are

for channel 1 and

for channel 2.

Fig. 10. Comparison of conventional and the enhanced multiplexing networks:(a) the transmission coefficient for channel 1 and (b) the transmission coefficientfor channel 2.

TABLE IVFLATNESS OF THE TRANSMISSION COEFFICIENTS IN THE PASSBAND OF EACH

CHANNEL FILTER

TABLE VCHANNEL FILTER LENGTHS OBTAINED FROM CONVENTIONAL DESIGN

APPROACH USING FIVE-POLE FILTERS AND THE PROPOSED METHOD

To further demonstrate the advantage of the proposed method,the multiplexer is designed with two five-pole channel filtersusing conventional method. The lengths of channel filters arecompared in Table V. The channel filters are classical dual-


Fig. 11. Simulated results of the enhanced design of the four-channel con-tiguous multiplexer. (a) Transmission coefficient between the input port of eachchannel filter and the common port, i.e., �� , where � is the port number.(b) Input return loss of each channel filter, i.e., �� .

mode waveguide filters with the cavity diameter of 26.162 mm.The filter lengths of the enhanced multiplexer are about 2/3 ofthe five-pole channel filters since a five-pole filter requires threecavities whereas a four-pole file only requires two cavities.

E. Example of Contiguous Multiplexer With Measured Data

The enhanced multiplexing network design approach hasbeen applied to a four-channel contiguous multiplexer. Thechannel filters are all four-pole filters centered at 3.902 GHz,3.822 GHz, 3.742 GHz and 3.662 GHz, respectively, each with72 MHz bandwidth. Half height WR229 waveguide E-planeT-junctions with mm and mm areused in this example.

Fig. 11(a) shows the simulated transmission coefficient be-tween the input port of each channel filter and the common port,i.e., , where is the port number, of the enhanced designof the four-channel contiguous multiplexer. It is more conve-nient to view the additional resonance created by the manifoldfrom the input return loss of channel filters, i.e., . As shownin Fig. 11(b), all channel filters behave like five-pole filters al-though only four-pole filters are attached to the manifold.

The four-channel contiguous enhanced multiplexer has beenbuilt and tested. Fig. 12 shows the isometric view (ProE) of thefabricated multiplexer. As shown in Fig. 13, measurement re-sults and simulation results agree very well for all channel fil-

Fig. 12. Isometric view (ProE) of the fabricated multiplexer.

Fig. 13. Measurement results of (a) the transmission coefficient and (b) theinput return loss of each channel filter of the enhanced multiplexing network.

ters, validating the design methodology presented herein. Notethat the resonances have shifted slightly during the tuning of themultiplexer, which is required to compensate manufacture tol-erances. As a result, for channel 2, centered at 3.822 GHz, andchannel 4, centered at 3.662 GHz, the additional resonances arenot shown as clearly as in the simulation. For channel 4, a com-parison between the measurement in Fig. 13(b) and the simula-tion in Fig. 11(b) shows that there is an extra resonance locatedat 3.69 GHz. For channel 2, the input return loss does not showfive distinct peaks. However the improvement in the channelfilter performance is preserved as the same design specificationsare met using both simulation and measurement results.

III. CONCLUSION

A method for systematical design of enhanced microwavemultiplexing network is presented in order to improve channel


performance. It is shown that interconnecting network such asmanifold can be utilized to form an additional real reflectionzero for channel filters through design and optimization ofthe distance between channel filters and the distance betweeneach channel filter and the manifold. The channel filter order istherefore increased without additional resonators. Significantimprovement in multiplexer performance is achieved withoutincrease of size or weight. Waveguide E-plane T-junctionsare used for demonstration of the design approach. However,the proposed method is not limited to waveguide manifold.It is applicable to all multiplexers using transmission lines toconnect channel filters. Simulation and measurement resultsdemonstrate the feasibility of such multiplexing networks.

REFERENCES

[1] R. Cameron and M. Yu, “Design of manifold coupled multiplexers,”IEEE Microw. Mag., pp. 46–59, Oct. 2007.

[2] C. Kudsia, R. Cameron, and W.-C. Tang, “Innovations in microwavefilters and multiplexing networks for communications satellitesystems,” IEEE Trans. Microw. Theory Tech., vol. 40, no. 6, pp.1133–1149, Jun. 1992.

[3] S. Lundquist, M. Yu, D. Smith, and W. Fitzpatrick, “Ku- band tem-perature compensated high power multiplexers,” presented at the Proc.20th AIAA Int. Commun. Satellite Syst. Conf. Exhibit, Montreal, QC,Canada, May 2002.

[4] S. Lundquist, M. Mississian, M. Yu, and D. Smith, “Application of highpower output multiplexers for communications satellites,” presentedat the Proc. 18th AIAA Int. Commun. Satellite Syst. Conf. Exhibit,Oakland, CA, Apr. 2000.

[5] M. Yu, D. J. Smith, A. Sivadas, and W. Fitzpatrick, “A dual mode filterwith trifurcated iris and reduced footprint,” in Proc. IEEE MTT-S Int.Microw. Symp. Dig., Seattle, WA, Jun. 2002, pp. 1457–1460.

[6] J. Zheng and M. Yu, “Rigorous mode-matching method of circular tooff-center rectangular side-coupled waveguide junctions for filter ap-plications,” IEEE Trans. Microw. Theory Tech., vol. 55, no. 11, pp.2365–2373, Nov. 2007.

[7] A. E. Atia, “Computer-aided design of waveguide multiplexers,” IEEETrans. Microw. Theory Tech., vol. 22, pp. 332–336, Mar. 1974.

[8] J. D. Rhodes and R. Levy, “A generalized multiplexer theory,” IEEETrans. Microw. Theory Tech., vol. 27, no. 2, pp. 99–111, Feb. 1979.

[9] J. D. Rhodes and R. Levy, “Design of general manifold multiplexers,”IEEE Trans. Microw. Theory Tech., vol. 27, no. 2, pp. 111–123, Feb.1979.

[10] “Enhanced Microwave Multiplexing Network,” U.S. Patent 7397325B2, Jul. 8, 2008.

[11] J. B. Ness, “A unified approach to the design, measurement, and tuningof coupled-resonator filters,” IEEE Trans. Microw. Theory Tech., vol.46, no. 4, pp. 343–351, Apr. 1998.

[12] R. J. Cameron, C. M. Kudsia, and R. R. Mansour, Microwave Filters forCommunication Systems: Fundamentals, Design and Applications.New York: Wiley, 2007.

[13] N. Marcuvitz, Waveguide Handbook The Institution of Electrical En-gineers. Piscataway, NJ, 1986.

[14] A. E. Atia and A. E. Williams, “Narrow-bandpass waveguide filters,”IEEE Trans. Microw. Theory Tech., vol. 20, no. 4, pp. 258–265, Apr.1972.

[15] M. A. Ismail, D. Smith, A. Panariello, Y. Wang, and M. Yu, “EM-based design of large-scale dielectric-resonator filters and multiplexersby space mapping,” IEEE Trans. Microw. Theory Tech., vol. 52, no. 1,pt. 2, pp. 386–392, Jan. 2004.

Ming Yu (S’90–M’93–SM’01–F’09) received thePh.D. degree in electrical engineering from theUniversity of Victoria, Victoria, BC, Canada, in1995.

In 1993, while working on his doctoral disserta-tion part time, he joined COM DEV, Cambridge, ON,Canada, as a Member of Technical Staff. He was in-volved in designing passive microwave/RF hardwarefrom 100 MHz to 90 GHz for both space and groundbased applications. He was also a principal developerof a variety of COM DEV’s core design and tuning

software for microwave filters and multiplexers, including computer aide tuningsoftware in 1994 and fully automated robotic diplexer tuning system in 1999.His varied experience also includes being the Manager of Filter/MultiplexerTechnology (Space Group) and Staff Scientist of Corporate Research and De-velopment (R&D). He is currently the Chief Scientist and Director of R&D.He is responsible for overseeing the development of company R&D Roadmapand next generation products and technologies, including high frequency andhigh power engineering, electromagnetic based CAD and tuning for complexand large problems, novel miniaturization techniques for microwave networks.He is also an Adjunct Professor with the University of Waterloo, ON, Canada.He holds NSERC Discovery Grant from 2004–2013 with Waterloo. He has au-thored or coauthored over 90 publications and numerous proprietary reports. Heholds 8 patents with 6 more pending.

Dr. Yu is an IEEE Distinguished Microwave Lecturer from 2010 to 2012. Heis MTT Filter committee Chair (MTT-8) since 2010 and also served as Chairof TPC-11. He is an Associate Editor of IEEE TRANSACTIONS ON MICROWAVE

THEORY AND TECHNIQUE. He was the recipient of the 1995 and 2006 COM DEVAchievement Award for the development a computer-aided tuning algorithmsand systems for microwave filters and multiplexers.

Ying Wang (M’05) received the B.Eng. and theM.S. degrees in electronic engineering from NanjingUniversity of Science and Technology, Nanjing,China, and the Ph.D. degree in electrical engineeringfrom University of Waterloo, Waterloo, ON, Canada,in 1993, 1996, and 2000, respectively.

From 2000 to 2007, she was with COM DEV,Cambridge, Ontario, Canada. As a Senior Member ofTechnical Staff, she was involved in development ofCAD software for design, simulation and optimiza-tion of microwave circuits for space application. In

2007, she joined the Faculty of Engineering and Applied Science, University ofOntario Institute of Technology, Oshawa, ON, Canada, where she is currentlyan Assistant Professor. Her research interests include RF/microwave computeraided design, microwave circuits design and radio wave propagation modeling.

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270 IEEE TRANSACTIONS ON MICROWAVE THEORY AND …mingyu/publications/p201102.pdf · 2013-11-23 · 270 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 59, NO. 2, FEBRUARY