1974 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, … · 2020-04-20 · 1974 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 4, APRIL 2012 High Step-up Boost Converter Integrated

1974 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 4, APRIL 2012

High Step-up Boost Converter Integrated Witha Transformer-Assisted Auxiliary Circuit Employing

Quasi-Resonant OperationKi-Bum Park, Member, IEEE, Gun-Woo Moon, Member, IEEE, and Myung-Joong Youn, Senior Member, IEEE

Abstract—Stacking an auxiliary step-up circuit on top of a boostconverter is one of the most attractive structures for nonisolatedhigh step-up applications. In this paper, in order to avoid the largeinput current ripple of coupled-inductor-based circuits, an auxil-iary step-up circuit is integrated via an additional transformer andits balancing capacitor. A voltage-doubler is adopted as an auxiliarystep-up circuit, which is inherently suitable for high-voltage appli-cations due to its simple structure and low-voltage stress. More-over, the transformer leakage inductor and the balancing capaci-tor constitute a resonant tank so that the quasi-resonant operationmakes the current sinusoidal. As a result, a reduced switch turn-OFF loss and reverse recovery of the diode can be expected. The pro-posed converter is verified with a 24 V input, 160 W – 200 V outputprototype.

Index Terms—Coupled-inductor, high step-up converter, quasi-resonant voltage-doubler.

I. INTRODUCTION

FOR battery-powered systems, electric vehicles, fuel cellsystems, and photovoltaic systems, where low-voltage

sources need to be converted into high voltages, the demand fornonisolated high step-up dc–dc conversion techniques are grad-ually increasing [1]–[6]. A classic boost converter is widelyused due to its simple structure and its continuous input cur-rent. However, it is hard to achieve a high-voltage conversionratio with just a plain boost converter, since the parasitic re-sistance of the circuit causes a severe loss as the duty cycle isincreased which limits the step-up gain [7]. Especially in highoutput voltage applications, high-voltage stress on switches anddiodes degrades the performance of devices, causing a severehard switching loss, a conduction loss, and a reverse recoveryproblem [8]–[10]. Moreover, an increased duty cycle to obtaina high step-up gain has a detrimental effect on the dynamicperformance of a boost converter [7]. Therefore, to relieve the

Manuscript received June 23, 2011; accepted September 14, 2011. Dateof current version February 20, 2012. This paper was presented in part atthe IEEE ECCE, 2010, under the title “High step-up boost converter inte-grated with voltage-doubler.” Recommended for publication by Associate EditorM. Vitelli.

K.-B. Park is with the Power Electronic Systems Group, ABB Corpo-rate Research, Segelhofstrasse 1K 5405 Baden-Daettwil, Switzerland (e-mail:[email protected]).

G.-W. Moon and M.-J. Youn are with the KAIST, Electrical Engineer-ing, 373-1, Kuseong-dong, Yuseong-gu, Daejeon 305-701, Republic of Korea(e-mail: [email protected], [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TPEL.2011.2170223

abovementioned limitations on boost converters in high step-upapplications, it is necessary to reduce the operating duty cycleand distribute the voltage stresses across the devices allowingfor the use of low-voltage high-performance devices.

Till now, various types of step-up techniques based on a boostconverter have been developed [11]–[29]. Cascading a boostconverter is a simple way to achieve a high step-up gain byemploying more components. However, an increased numberof powering processes degrades the efficiency and the burden ofa high-voltage stress still remains [11], [12]. Coupled-inductorboost converters are favorable candidates due to their simplestructure. However, their input current ripple is large due to acoupled–inductor effect and an additional voltage clamp cir-cuit for the switch and diode is required [13]–[20]. A voltage-multiplier cell or a switch-capacitor circuit can also be usefulto raise the step-up gain in collaboration with basic topolo-gies [21]–[24]. However, when a higher output voltage is re-quired, the number of step-up stages is increased, which re-quires more components. In addition, a current snubber is alsorequired to suppress the excessive peak current for chargingthe capacitors. In terms of isolated type converters, current-fedconverters easily offer a high step-up gain through the turn ratioof the transformer, which makes them inherently suitable forhigh step-up applications [25], [26]. As a price for utilizing atransformer, an additional voltage snubber is needed to limit theswitch voltage spikes caused by the transformer leakage induc-tance. In addition, an auxiliary circuit for operation below 0.5duty cycles is required.

Among the abovementioned high step-up circuits, a coupled-inductor-based circuit seems to be one of the most suitable can-didates in low-to-medium power applications due to its simplestructure. In order to remove the large input current ripple andimprove the performance of this circuit further, an alternativestructure, which is based on a boost converter integrated with atransformer assisted auxiliary step-up circuit, is investigated inthis paper. Furthermore, a quasi-resonant operation is adoptedto reduce the switch turn-OFF loss caused by the large switchcurrent in a high step-up converter, without additional circuitry.

II. PROPOSED HIGH STEP-UP CONVERTER

In order to further raise the step-up gain of a boost con-verter, alternative structures which combines a boost converterwith an auxiliary step-up circuit in series have been devel-oped [15], [17], [27]–[29]. Proper selection of an auxiliary mod-ule can offer advantages such as high step-up capability, design

0885-8993/$26.00 © 2011 IEEE

PARK et al.: HIGH STEP-UP BOOST CONVERTER INTEGRATED WITH A TRANSFORMER-ASSISTED AUXILIARY CIRCUIT 1975

Fig. 1. High step-up converter employing auxiliary step-up circuit on top ofboost converter. (a) Conventional coupled-inductor-assisted auxiliary circuit.(b) Proposed transformer-assisted auxiliary circuit.

flexibility, and distributed voltage stress across devices whichallows for low voltage high performance devices. Among them,the coupled-inductor-assisted auxiliary step-up circuit shown inFig. 1(a) is promising due to its simple structure, having onlyone switch and an easy high step-up capability utilizing theturn ratio of the coupled-inductor [15], [17]. However, thesestructures also suffer from some side effects that come from thecoupled-inductor as follows.

As the output voltage of the auxiliary circuit is increased,more power is driven by the coupled-inductor to the secondaryside. Since the input current consists of the boost inductor cur-rent, ILb , and the reflected auxiliary circuit current, nIaux , theinput current ripple can get considerably larger (‘n’ is the turnratio of the coupled-inductor). That is, as n increases, the in-put current ripple increases, and more input filtering might berequired, which has a detrimental effect on the total efficiency.Moreover, since this input current flows through the switch, alarge input current ripple could cause a large switch turn-OFF

loss.In terms of the size of the coupled-inductor in high step-

up applications, it can be rather large when compared witha plain inductor, since it requires many secondary turns anda large ac-current being superimposed to a dc-current flowsthrough the windings. That is, the coupled-inductor needs to bedesigned like a flyback transformer [30]. So that the coupled-inductor size is not increased too much, a smaller inductancedesign is one option. However, this increases the peak inductorcurrent in return, which results in an even higher switch turn-OFF loss. As the input current increases, these drawbacks couldimpose a greater burden on the magnetic component design andefficiency.

Fig. 2. Proposed high step-up boost converter integrated with voltage-doubleras auxiliary step-up circuit, employing quasi-resonant operation.

In order to relieve aforementioned drawbacks of coupled-inductor-based circuits, a transformer-assisted auxiliary highstep-up circuit, which has a continuous input current, is in-troduced in this paper as shown in Fig. 1(b). In addition, aquasi-resonant operation utilizing the transformer leakage in-ductance is adopted in part [31]–[34], which provides partialsoft-switching characteristics for both the switches and thediodes. Fig. 2 shows the proposed high step-up converter, andits main features are as follows.

A. Voltage-Doubler as an Auxiliary Step-up Circuit

Various types of rectifiers can be adopted as an auxiliary step-up circuit. Among them, a voltage-doubler is inherently suitablefor high-voltage applications due to its simple structure, whichconsists of two diodes and two capacitors, and due to its low-voltage stress on devices, which is clamped to the output voltageof the voltage-doubler. Therefore, they are widely adopted inmany topologies as a part of the circuit [17], [19], [27], [29],[31], [35], [37]. For the proposed converter, a voltage-doubler isalso employed as an auxiliary step-up circuit, but it is integratedwith a boost converter in a different manner that allows for thefollowing distinctive properties.

B. Separated Boost-Inductor and Transformer

In the proposed circuit, unlike a coupled-inductor-based cir-cuit, the interface between the boost converter and the voltage-doubler is accomplished by an additional transformer, whichalso contributes to the step-up gain by means of the turn ratio n.Since a square voltage waveform, i.e., an ac voltage, is appliedacross the switch Q, the transformer can be inserted in parallelwith Q. Then, the balancing capacitor, CR , is inserted into theprimary side of the transformer to make up for the flux-balanceof the transformer. Thereby, the voltage-doubler is coupled withthe boost converter by sharing a common switch. Therefore,by the switching action of Q, both the boost converter and thevoltage-doubler are operated at the same time.

Compared with the large input current ripple of the coupled-inductor assisted boost converters in [13]–[17], the proposedconverter maintains a continuous input current of ILb , which


could require less input filtering. In other words, the proposedcircuit decouples the large ac-current from the input side tothe transformer at the price of additional components, i.e., thetransformer and the balancing capacitor. As a result, the inputcurrent ripple becomes continuous, and each inductor and trans-former can be designed optimally compared with the coupled-inductor [30].

C. Quasi-Resonant Operation

Although the large input current ripple is removed, it is trans-ferred to the transformer and this large current ripple still flowsthrough the switch in addition to the boost inductor currentILb , which could result in a high turn-OFF loss. To relievethis, a quasi-resonant operation, which considerably reducesthe switch turn-OFF current, is adopted as follows.

In a separated transformer structure followed by the voltage-doubler, the leakage inductance of the transformer, Llkg , andthe balancing capacitor, CR , can constitute a resonant tank sim-ilar to that of the series resonant type converters in [31]–[34].The quasi-resonant operation between Llkg and CR makes thecurrent on the transformer primary side and the voltage-doublersinusoidal during the switch-ON state. Owing to this sinusoidalcurrent, the switch turn-OFF current can be considerably reducedresulting in less turn-OFF loss [36]. Moreover, since the sinu-soidal current guarantees slow di/dt, the reverse recovery on thediode of the voltage-doubler can also be alleviated.

III. OPERATION PRINCIPLES

The proposed converter combines the operations of a boostconverter and a voltage-doubler, with the common switch-ing function of Q, employing pulse-width modulation (PWM).Since the voltage-doubler utilizes the quasi-resonant operationbetween Llkg and CR , its operation can be divided into two re-gions according to the relationship between the resonant period,TR , in (1) and the duty cycle, D. That is, the above-resonant(AR) region, where TR /2 > DTS , and the below-resonant (BR)region, where TR /2 < DTS , which is similar to conventional res-onant converters [31]–[34]. The detailed operation is presentedas follows.

TR = 2π√

LlkgCR (1)

.

A. Below-Resonant Region (TR /2 < DTS )

The key waveform and the topological states in the BR regionare shown in Figs. 3 and 4, respectively.

Mode 1 [t0 ∼ t1]: Q is in the ON-state and VS is appliedto the boost inductor LB . The boost inductor current, ILb ,flows through Q and is increased linearly. At the same time,the voltage-doubler is operated with the common switching ac-tion of Q. The powering path from CR to the lower output ofthe voltage-doubler, Vo2 , is formed through the transformer,Q, and Do2 , as represented by the dotted line. The Llkg andCR constitute a resonant tank and derive a powering currentwith a sinusoidal shape. The resonant capacitor voltage, VCr , is

Fig. 3. Key waveforms in below-resonant region (TR /2 < DTS ).

decreased. The switch current, IQ , comprises ILb and the trans-former primary current of the voltage-doubler, Ilkg . Since Do 2 isturned-OFF with a very slow slope of IDo2 , the reverse recoverycan be minimized. Do3 is blocked by Vo 2 + Vo 3 .

Mode 2 [t1 ∼ t2]: Since TR /2 is shorter than the switch on-interval DTS , the resonant operation is finished at t1 before Qis turned-OFF. Only ILb flows through Q. Therefore, the switchturn-OFF loss is only affected by ILb . Since no current flowsthrough CR , VCr keeps its value during this interval.emphasis

Mode 3 [t2 ∼ t3] : Q is turned-OFF at t2 , then ILb flows throughDo1 . Meanwhile, the voltage-doubler starts to conduct in theopposite direction. That is, the resonant powering path from theoutput of the boost converter, Vo1 , to the upper output of thevoltage-doubler, Vo 3 , is formed through Do 1 , the transformerand Do3 , as represented by the dotted line. Therefore, by theresonant operation between Llkg and CR , IDo3 is increased.Since the resonant current flows through Do1 in the oppositedirection of ILb , IDo1 is decreased accordingly. Do 2 is blockedby Vo 2 + Vo 3 .

Mode 4 [t3 ∼ t4]: IDo1 reaches zero at t3 . Then all of ILbflows through the transformer and Do3 of the voltage-doubler.The ILb charges CR , increasing VCr linearly. The Do 1 is blockedby Vo 1-VCr-Vo 3 /n, which is slowly decreased. The same amountof change can be observed in VQ .

Mode 5 [t4 ∼ t5]: At t4 , VCr is increased enough to conductDo1 again. In this mode, unlike mode 3, the powering path fromVo3 to Vo1 , is formed through Do3 , the transformer and Do 1 by


Fig. 4. Topological states of below-resonant region. (a) Mode 1 [t0 ∼ t1 ]. (b) Mode 2 [t1 ∼ t2 ]. (c) Mode 3 [t2 ∼ t3 ]. (d) Mode 4 [t3 ∼ t4 ]. (e) Mode 5 [t4 ∼t5 ]. (f) Mode 6 [t5 ∼ t6 ].

the resonant operation between Llkg and CR . In other words, theflow of ILb is shifted slowly in the resonant way from IDo3 toIDo1 . Here, the reverser recovery on Do3 can be reduced by theslow slope of IDo3 .

Mode 6 [t5 ∼ t6]: IDo3 reaches zero at t5 . Then all of ILb flowthrough Do 1 . Since there is no current flowing through CR , VCrkeeps its value during this mode. As D is increased, modes 5and 6 gradually fade and disappear.

B. Above-Resonant Region (TR /2 > DTS )

Operation in the AR region is similar to that of the BR regionexcept for mode 2 of the BR region, where ILb flowing solelythrough Q, is ignored since TR /2 is longer than DTS . The keywaveform and the topological states in the AR region are shownin Figs. 5 and 6, respectively. Since some of the topologicalstates are the same as those of the BR region, only the differenttopological states, the intervals t0 ∼ t1 and t2 ∼ t3 , are presentedin Fig. 6. The topological states of the intervals t1 ∼ t2 , t3 ∼ t4 ,and t4 ∼ t5 in the AR region correspond to Fig. 4(a), 4(c), and4(d), respectively.

It is noted that, in the AR region, since Q is turned-OFF whilestill powering through the transformer, the switch current at theturn-OFF instant of t2 comprises ILb and Ilkg . Therefore, theturn-OFF loss can be increased compared with that in the BRregion where only ILb flows at the switch turn-OFF instant.

Fig. 5. Key waveforms in above-resonanat region (Tr /2 > DTs ).


Fig. 6. Topological states of above-resonant region. (a) t0 ∼ t1 . (b) t2 ∼ t3 .

Fig. 7. Transformer turn ratio n according to a variation of M.

IV. ANALYSIS AND CHARACTERISTICS

A. Input-Output Voltage Gain

For the sake of analysis, assuming the ripple of VCr is ig-nored and using a flux-balance on the boost inductor and thetransformer, the following voltage equations are obtained:

Vo1 =1

1 − DVS (2)

Vo2 = nVS (3)

Vo3 =nD

1 − DVS (4)

VO =1 + n

1 − DVS (5)

VCr avg = VS . (6)

The Vo1 is the same as the output voltage of a classical boostconverter and the voltage-doubler provides a voltage that isn times higher, nVo 1 ( = Vo 2 + Vo3). Fig. 7 shows the re-quired turn ratio n according to the variation of D and the input-output voltage conversion ratio M. When D becomes zero, thevoltage-doubler does not operated and ILb flows through all ofthe series-connected diodes, Do1 , Do 2 , and Do3 . That is, Vo 2and Vo 3 become zero and VO follows VS in the same manner asa conventional boost converter.

B. Voltage and Current Stress on the Device

In the boost converter, the voltage stresses on Q and Do 1 areVo 1 , i.e., VS /(1-D). In the voltage-doubler, the voltage stresses

on Do 2 and Do 3 are Vo2+Vo 3 , i.e., nVS /(1-D). That is, thevoltage stress on the voltage-doubler is n times higher than thatof the boost converter. Since the voltage-doubler provides n/(1+ n) of the total output voltage, the transformer handles n/(1+n)of the total power accordingly. The voltage ripple and the peakvoltage stress of VCr are expressed as (7) and (8), respectively.

VCr rp =nIO TS

CR(7)

VCr peak = VCr avg +VCr rp

2= VS +

nIO TS

2CR. (8)

Since Co 1 , Co 2 , and Co 3 are connected in series, the averagecurrent of IDo1 , IDo2 , and IDo3 is the same as IO . The peakcurrent of IDo1 is the same as the turn-OFF current of IQ , whichis especially high in the AR region due to the remaining resonantcurrent at the turn-OFF instant. The peak current of IDo3 is similarto that of ILb reflected to the transformer secondary. Assumingthat DTS ≈ TR /2, the peak current stresses on Do 2 and IQ canbe expressed as in (9) and (10), respectively.

IDo2 peak ≈ πTS IO

TR≈ πIO

2D(9)

IQ peak ≈ Iin avg +nπTS IO

TR

≈{

M (π + 2D − πD) − π

2D

}IO (10)

IQ rms ≈

√1TS

∫ T R2

0

(nπTS IO

TRsin (ωRt) + Iin avg

)2

dt

≈ IO

√1

8D{M 2−2M + 1+(16M 2 − 14M) D−9M 2D2}.

(11)

C. ZCS on the Diodes

The diode currents in the voltage-doubler always flow throughLlkg that provides a current snubbing effect. Therefore, the re-verse recovery on Do 2 and Do3 can always be reduced by aslow di/dt. The steepest slopes of IDo2 and IDo3 can occur at theswitching transition in the AR region as shown in Fig. 5. Equa-tions (12) and (13) represent the decreasing di/dt of IDo2 andIDo3 , respectively, which are the same. In order to sufficiently


Fig. 8. (a) Switch peak current stress and (b) switch rms current according to a function of M.

Fig. 9. Operating duty cycle D according to variations of VS and n.

reduce the reverse recovery, di/dt should be at least less than100 A/μs [9].

dIDo2

dt=

VCR(t2) + Vo3/n

nLlkg=

1nLlkg

(1

1 − DVS − IO

CRFS

)

(12)

dIDo3

dt=

Vo1 + Vo2/n − VCR(t5)nLlkg

=1

nLlkg

(1

1 − DVS − IO

CRFS

). (13)

On the other hand, in the BR region, where the half-period reso-nant operation between Llkg and CR provides an extremely slowslope on IDo2 , zero-current-switching (ZCS) can be achieved onDo2 , minimizing the reverse recovery.

In the case of Do 1 , during the switch-OFF state, IDo1 is de-creased to zero and then it is increased again as described in themode analysis in the BR region of Section II. This operationis also caused by the resonant operation between Llkg and CR ,therefore, it depends on TR and DTS . In both the BR and ARregions, as TR and DTS get smaller, there is a greater chance ofreincreasing IDo1 . In this case, when Q is turned-ON, an abruptchange in IDo1 occurs causing a reverse recovery. Unless IDo1is increased again, a reverse recovery on Do 1 will not occur.

Fig. 10. Area-product, AP , of transformer according to variation of VS .

Fig. 11. Resonant current waveforms according to a variation of TR .

V. DESIGN CONSIDERATION

In general, the power level of a certain topology is mainlydetermined by the component count and rating. That is, the moreswitches are employed, the higher the power level. The proposedconverter utilizes only one switch, therefore, it is expected to besuitable for low-to-medium power applications.

To illustrate the design procedure for the proposed circuit, an18 ∼ 30 V input, 200 V output, 160 W prototype converter ispresented. The required input-output voltage gain M is variedfrom 6.7 ( = 200/30), for a maximum input of 30 V, to 11.1 ( =200/18) for a minimum input of 18 V. The nominal input voltageis 24 V, for which the required gain M is 8.3 ( = 200/24).


TABLE IEXPERIMENTAL PARAMETERS

Fig. 12. Experimental waveforms at VS = 24 V with full load condition.

A. Transformer Turn Ratio and Duty Cycle

In the proposed converter design, the selection of a switch,which is burdened by the sum of the boost inductor current andthe resonant current, is primarily considered in terms of cost andefficiency. As presented in (11) and Fig. 8(b), for the same M,the rms value of the switch current is slowly decreased as theduty cycle is increased. On the other hand, the switch voltagestress, Vo1 = VS /(1- D), is decreased with a decrease in the duty

cycle, which leads to the use of a lower-voltage switch having asmaller on-resistance. However, a smaller duty cycle results in alarger turn ratio n as show in Fig. 7, which increases the voltagestress of the voltage-doubler. Therefore, a duty cycle should beselected to accommodate as low a voltage stress on the switch aspossible while not increasing the burden of the voltage-doublertoo much.

By selecting the transformer turn ratio n = 3.5, the duty cyclevaries from 0.4 ∼ 0.6 in response to the VS change from 30 V



to 18 V, as can be seen in Fig. 9. The ratio of the boost converteroutput, Vo 1 , to the voltage-doubler output, Vo 2 + Vo3 , is always1 : n + 1 regardless of the input variation, as can be seenin (2)–(5). When n = 3.5, Vo1 and Vo 2 + Vo 3 become about45 V and 155 V, respectively. Therefore, 100 V devices for theboost converter and 200 V devices for the voltage-doubler areavailable.

B. Inductor and Transformer

The design of the boost inductor is the same as those ofconventional ones. Assuming the current ripple to be 15% ofthe input current 6.7 A, LB is designed for 120 μH [30].

Since the balancing capacitor, CR , is inserted into the pri-mary side and the voltage-doubler capacitors, Co 2 and Co 3 , arelocated in the secondary side, no dc-current can flow throughthe transformer by the charge-balance of the capacitors, eventhough the voltage applied to the transformer is asymmetrical.Therefore, the magnetizing current has no dc-offset, which isbeneficial for the transformer design [30].

Normally, the area-product AP method can be used to pre-dict the size of the magnetic core [30]. The AP represents theproduct between a cross-section area and the window area ofthe magnetic core. In the case of the proposed converter, the

AP of one transformer can be obtained as in (14), where Ku

is the window utilization factor, J is the current density, andBmax is the maximum flux density. Assuming Ku to be 0.3, J tobe 300 A/cm2 , Bmax to be 0.1 T, IO to be 0.8 A, and FS to be100 kHz, the AP of the transformer according to the function ofVS is illustrated in Fig. 10, where the dot represents the case ofn = 3.5. The AP is varied according to a change in VS and themaximum AP is 0.73 cm4 in the case of VS = 24 V.

AP =D (VO (1 − D) − VS ) IO

BmaxFS KuJ

√π2

8D+

11 − D

. (14)

C. Resonant Tank

Fig. 11 shows the current waveform according to TR . In theAR region, the switch turn-OFF loss is increased. On the otherhand, in the BR region, the switch turn-OFF loss is reducedand Do2 achieves a zero-current-switching (ZCS) turn-OFF thatminimizes the reverse recovery of the diode. However, the cur-rent stress and the conduction loss of the devices are increased.Therefore, TR should be designed around the midpoint, TR /2 =DTS , to achieve ZCS of the diode while minimizing the switchturn-OFF loss and the conduction loss. To be designed at thispoint, once Llkg is obtained from the fabricated transformer,



Llkg can be set as it is and CR can then be selected as in (15).

CR =D2T 2

S

π2Llkg. (15)

With respect to the reverse recovery of the diode, Llkg shouldalso be considered as a current snubber. The minimum Llkgwhich guarantees di/dt below 100 A/μs is approximately ob-tained at 150 nH from (10). If the obtained Llkg from the fabri-cated transformer has a larger inductance than 150 nH, a severereverse recovery will be relieved. As Llkg is increased, it is morebeneficial to reverse recovery. However, a larger Llkg reducesCR to maintain the same TR as presented in (15), which lead toa larger voltage ripple in VCr in return, as noted in (7).

VI. EXPERIMENTAL RESULTS

To verify the proposed converter, a 160 W prototype is im-plemented. The specifications and design parameters obtainedfrom the design example are presented in Table I.

Fig. 12 shows the experimental waveforms at a nominal inputof 24 V under the full load condition. The duty cycle is about0.5 and it is similar to TR /2. The resonant operation betweenLlkg and CR makes Ilkg sinusoidal and only the boost inductorcurrent ILb flows through the switch Q at the turn-OFF instant,resulting in a reduced switch turn-OFF loss. That is, althoughthe peak switch current exceeds 16 A, the turn-OFF current isunder 9 A. Moreover, both Do2 and Do3 achieve ZCS turn-OFF, which alleviates the reverser recovery. The boost converteroutput Vo 1 is about 50 V. Therefore, the voltage stresses on Q

and Do1 are under 100 V, including the voltage spikes causedby parasitic inductances, which allows for the use of a Schottkydiode for Do1 . The voltage stresses on Do 2 and Do3 are clampedto Vo2+Vo 3 , at about 150 V. The input current, i.e., the boostinductor current ILb , is continuous.

Figs. 13 and 14 show the experimental waveforms at VS =18 V and VS = 30 V, respectively. In the case of VS = 18 V, theduty cycle is increased to regulate VO and the circuit is operatedin the BR region of DTS > TR /2. Here, the switch turn-OFF

current is still the same as ILb . In the case of VS = 30 V, thecircuit is operated in the AR region, i.e., DTS < TR /2, and theswitch turn-OFF current is about 12 A despite the fact that ILbis only about 7 A. In both cases, the reverse recoveries on Do 2and Do3 are sufficiently suppressed by the current snubbingeffect of Llkg . The voltage stress on the switch in the steady-state is still about 50 V regardless of the input variation. Thisimplies that the switch turn-ON loss is rarely affected by an inputchange.

Fig. 15 shows the efficiency curves with respect to the vari-ation of VS . Since the resonant tank is designed to satisfy thecondition DTS = TR /2 at VS = 24 V, the proposed circuit showshigh efficiency, over 93%, at this point along a wide load range.In the case of VS = 18 V, where the converter is operated inthe BR region, the increased conduction loss caused by the in-creased input current would degrade the efficiency. On the otherhand, in the case of VS = 30 V, the operation in the AR regionincreases the switch turn-OFF loss even though the conductionloss is decreased by a low input current. Consequently, it is notedthat the efficiency of the proposed converter is affected by the


Fig. 15. Measured efficiency.

resonant tank design. That is, the proper selection of a balancingcapacitor, CR , can improve the circuit performance at a certainoperating point, which in this experiment is the nominal 24 Vinput, as expected.

VII. CONCLUSION

A coupled-inductor-assisted auxiliary step-up circuit is an at-tractive candidate in low-to-medium power applications due toits simple structure and it can easily achieve a high step-up gainby increasing the turn ratio of the coupled-inductor. However, ithas large input current ripple, which may require more input fil-tering, and large switch current could cause a high turn-OFF loss.In order to remove the large input current ripple, an alternativestructure, where the auxiliary step-up circuit is integrated viaan additional transformer and its balancing capacitor, is intro-duced in this paper. A voltage-doubler is adopted as an auxiliarystep-up circuit, which provides a simple structure and low volt-age stress. In addition, the transformer leakage inductor and thebalancing capacitor, followed by the voltage-doubler, constitutea series resonant tank, and thereby the sinusoidal current canconsiderably reduce the switch turn-OFF loss and the reverserecovery on the diode. It is noted that other types of rectifierscan also be integrated with a boost converter, when interfacedby a transformer and a balancing capacitor.

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Ki-Bum Park (S’07–M’10) was born in Korea, in1981. He received the B.S., M.S., and Ph.D. degreesin electrical engineering from the Korea AdvancedInstitute of Science and Technology (KAIST), Dae-jeon, Korea, in 2003, 2005, and 2010, respectively.

He is currently a Scientist with ABB CorporateResearch Center, Baden-Dattwil, Switzerland. Hisresearch interests include power converters, multi-level inverter, server power supply, high power den-sity adapter, battery management system, and displaydriver circuit.

Dr. Park received the Second Prize Paper Award from the InternationalTelecommunications Energy Conference (INTELEC) 2009 and the Third PrizePaper Award from the Energy Conversion Congress and Exposition (ECCE)-Asia 2011.

Gun-Woo Moon (S’92–M’00) received the M.S. andPh.D. degrees in electrical engineering from the Ko-rea Advanced Institute of Science and Technology(KAIST), Daejeon, in 1992 and 1996, respectively.

He is currently a Professor in the Department ofElectrical Engineering, Korea Advanced Institute ofScience and Technology (KAIST), Daejeon, Korea.His research interests include modeling, design andcontrol of power converters, soft-switching powerconverters, resonant inverters, distributed power sys-tems, power-factor correction, electric drive systems,

driver circuits of plasma display panels, and flexible ac transmission systems.Dr. Moon is a member of the Korean Institute of Power Electronics (KIPE),

Korean Institute of Electrical Engineers (KIEE), Korea Institute of Telematicsand Electronics (KITE), Korea Institute of Illumination Electronics and Indus-trial Equipment (KIIEIE), and Society for Information Display (SID).

Myung-Joong Youn (S’74–M’78–SM’98) was bornin Seoul, Korea, in 1946. He received the B.S. degreein electrical engineering from Seoul National Univer-sity, Seoul, in 1970, and the M.S. and Ph.D. degreesin electrical engineering from the University ofMissouri, Columbia, in 1974 and 1978, respectively.

In 1978, he joined the Air-Craft Equipment Divi-sion, General Electric Company, Erie, PA, where hewas an Individual Contributor on Aerospace Electri-cal System Engineering. Since 1983, he has been aProfessor at the Korea Advanced Institute of Science

and Technology (KAIST), Daejeon, Korea. His research activities are in the ar-eas of power electronics and control, which include the drive systems, rotatingelectrical machine design, and high-performance switching regulators.

Dr. Youn is a member of the Institution of Electrical Engnieers, U.K., theKorean Institute of Power Electronics (KIPE), the Korean Institute of Electri-cal Engineers (KIEE), and the Korea Institute of Telematics and Electronics(KITE).

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1974 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, … · 2020-04-20 · 1974 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 27, NO. 4, APRIL 2012 High Step-up Boost Converter Integrated