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LTC1435A

High Efficiency Low NoiseSynchronous Step-Down

Switching Regulator

Figure 1. High Efficiency Step-Down Converter

FEATURES DESCRIPTIONU

Dual N-Channel MOSFET Synchronous Drive Programmable Fixed Frequency Wide VINRange: 3.5V to 36V Operation Low Minimum On-Time (300ns) for High

Frequency, Low Duty Cycle Applications Very Low Dropout Operation: 99% Duty Cycle Low Standby Current Secondary Feedback Control Programmable Soft Start

Remote Output Voltage Sense Logic Controlled Micropower Shutdown: IQ < 25A Foldback Current Limiting (Optional) Current Mode Operation for Excellent Line and Load

Transient Response Output Voltages from 1.19V to 9V Available in 16-Lead Narrow SO and SSOP Packages

The LTC1435A is a synchronous step-down switchingregulator controller that drives external N-channel powerMOSFETs using a fixed frequency architecture. A wideduty cycle range of 5% to 99% allows high VINto low VOUTDC/DC conversion, as well as low dropout operation thatextends operating time in battery-operated systems. BurstModeTM operation provides high efficiency at low loadcurrents.

The operating frequency is set by an external capacitor

allowing maximum flexibility in optimizing efficiency. Asecondary winding feedback control pin, SFB, guaranteesregulation regardless of load on the main output byforcing continuous operation. Burst Mode operation isinhibited when the SFB pin is pulled low, which reducesnoise and RF interference.

Soft start is provided by an external capacitor that can beused to properly sequence supplies. The operating cur-rent level is user-programmable via an external currentsense resistor. Wide input supply range allows operation

from 3.5V to 30V (36V maximum).

Notebook and Palmtop Computers, PDAs Cellular Telephones and Wireless Modems Portable Instruments Battery-Operated Devices DC Power Distribution Systems

APPLICATIONSU

TYPICAL APPLICATIONU

Burst Mode is a trademark of Linear Technology Corporation., LTC and LT are registered trademarks of Linear Technology Corporation.

ITH

TGCOSC43pF

CC

330pF

100pF

RC10k

CSS0.1F

SENSE SENSE+

LTC1435A

1000pF

+4.7F

DB

CMDSH-3CB

0.1F

M2Si4412DY

D1MBRS140T3

L14.7H

M1Si4412DY RSENSE

0.033

+ CIN22F35V2

VIN4.5V TO 22V

+COUT100F6.3V2

R135.7k

R2102k

VOUT1.6V/3A

PGND

BG

COSC VIN

VOSENSE

BOOST

1435A F01

SGND

RUN/SS

SW

INTVCC

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LTC1435A

ABSOLUTE MAXIMUMRATINGSWWW U

PACKAGE/ORDER INFORMATIONW UU

ORDER PARTNUMBER

Input Supply Voltage (VIN) .........................36V to 0.3VTopside Driver Supply Voltage (BOOST)....42V to 0.3VSwitch Voltage (SW)............................. VIN+ 5V to 5V

EXTVCCVoltage ........................................ 10V to 0.3VSENSE +, SENSE Voltages ...... INTVCC+ 0.3V to 0.3VITH, VOSENSEVoltages .............................. 2.7V to 0.3VSFB, Run/SS Voltages .............................. 10V to 0.3VPeak Driver Output Current < 10s (TG, BG) ............. 2AINTVCCOutput Current ........................................ 50mAOperating Ambient Temperature Range

LTC1435AC ............................................ 0C to 70CLTC1435AI ......................................... 40C to 85C

Junction Temperature (Note 1) ............................. 125CStorage Temperature Range ................. 65C to 150CLead Temperature (Soldering, 10 sec).................. 300C

TJMAX= 125C, JA= 130C/W (G)TJMAX= 125C, JA= 110C/W (S)

Consult factory for Military grade parts.

LTC1435ACGLTC1435ACSLTC1435AIGLTC1435AIS

TOP VIEW

S PACKAGE16-LEAD PLASTIC SO

G PACKAGE16-LEAD PLASTIC SSOP

1

23

4

5

6

7

8

16

1514

13

12

11

10

9

COSC

RUN/SSITH

SFB

SGND

VOSENSE

SENSE

SENSE+

TG

BOOSTSW

VIN

INTVCC

BG

PGND

EXTVCC

TA = 25C, VIN = 15V, VRUN/SS= 5V unless otherwise noted.ELECTRICAL CHARACTERISTICS

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

Main Control Loop

IIN VOSENSE Feedback Current (Note 2) 10 50 nA

VOSENSE Feedback Voltage (Note 2) 1.178 1.19 1.202 V

VLINEREG Reference Voltage Line Regulation VIN= 3.6V to 20V (Note 2) 0.002 0.01 %/VVLOADREG Output Voltage Load Regulation ITHSinking 5A (Note 2) 0.5 0.8 %

ITHSourcing 5A 0.5 0.8 %VSFB Secondary Feedback Threshold VSFBRamping Negative 1.16 1.19 1.22 V

ISFB Secondary Feedback Current VSFB= 1.5V 1 2 AVOVL Output Overvoltage Lockout 1.24 1.28 1.32 V

IQ Input DC Supply Current EXTVCC = 5V (Note 3) Normal Mode 3.6V < VIN < 30V 280 A Shutdown VRUN/SS = 0V, 3.6V < VIN < 15V 16 25 A

VRUN/SS Run Pin Threshold 0.8 1.3 2 V

IRUN/SS Soft Start Current Source VRUN/SS = 0V 1.5 3 4.5 AVSENSE(MAX) Maximum Current Sense Threshold VOSENSE= 0V, 5V 130 150 180 mVtON(MIN) Minimum On-Time Tested with Square Wave, SENSE

= 1.6V, 250 300 nsVSENSE= 20mV (Note 5 )

TG Transition Time

TG tr Rise Time CLOAD= 3000pF 50 150 nsTG tf Fall Time CLOAD= 3000pF 50 150 ns

BG Transition TimeBG tr Rise Time CLOAD= 3000pF 50 150 nsBG tf Fall Time CLOAD= 3000pF 40 150 ns

Internal VCCRegulator

VINTVCC Internal VCCVoltage 6V < VIN< 30V, VEXTVCC= 4V 4.8 5.0 5.2 V

VLDO INT INTVCCLoad Regulation IINTVCC= 15mA, VEXTVCC= 4V 0.2 1 %

VLDO EXT EXTVCCVoltage Drop IINTVCC= 15mA, VEXTVCC= 5V 130 230 mV

VEXTVCC EXTVCCSwitchover Voltage IINTVCC= 15mA, VEXTVCCRamping Positive 4.5 4.7 V

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LTC1435A

ELECTRICAL CHARACTERISTICS TA = 25C, VIN = 15V, VRUN/SS= 5V unless otherwise noted.

The denotes specifications which apply over the full operatingtemperature range.

LTC1435ACG/LTC1435ACS: 0C TA70CLTC1435AIG/LTC1435AIS: 40C TA85C

Note 1: TJis calculated from the ambient temperature TAand powerdissipation PDaccording to the following formula:

LTC1435ACG/LTC1435AIG: TJ= TA+ (PD)(130C/W)LTC1435ACS/LTC1435AIS: TJ= TA+ (PD)(110C/W)

Note 2:The LTC1435A is tested in a feedback loop which servos VOSENSEto the balance point for the error amplifier (V ITH = 1.19V).

Note 3:Dynamic supply current is higher due to the gate charge beingdelivered at the switching frequency. See Applications Information.

Note 4:Oscillator frequency is tested by measuring the COSCcharge anddischarge currents and applying the formula:

fOSC (kHz) = + 18.4(108)

COSC (pF) + 11( ) 1ICHG( )

1IDIS

Note 5:The minimum on-time test condition corresponds to an inductorpeak-to-peak ripple current 40% of IMAX(see Minimum On-TimeConsiderations in the Applications Information section).

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

Oscillator

fOSC Oscillator Frequency COSC = 100pF (Note 4) 112 125 138 kHz

TYPICAL PERFORMANCE CHARACTERISTICSUW

INPUT VOLTAGE (V)

070

EFFICIENCY(%)

75

80

85

90

100

5 10 15 20

1435A G01

25 30

95

ILOAD= 1A

ILOAD= 100mA

VOUT= 3.3V

Efficiency vs Input VoltageVOUT= 3.3V

VIN VOUTDropout Voltagevs Load Current

LOAD CURRENT (A)

00

VINVOUT(V)

0.2

0.1

0.3

0.4

0.5

0.5 1.0 1.5 2.0

1435A G04

2.5 3.0

RSENSE= 0.033VOUTDROP OF 5%

LOAD CURRENT (A)

0.00150

EFFICIENCY(%)

55

65

70

75

100

85

0.01 0.1 1

1435A G03

60

90

95

80

10

CONTINUOUSMODE

VIN= 10VVOUT= 5VRSENSE= 0.033

Burst ModeOPERATION

Efficiency vs Load Current

VITHPin Voltage vs Output Current

OUTPUT CURRENT (%)

0

VITH(V)

1.0

2.0

3.0

0.5

1.5

2.5

20 40 60 80

1435A G06

100100 30 50 70 90

Burst ModeOPERATION CONTINUOUS

MODE

INPUT VOLTAGE (V)

070

EFFICIENCY(%)

75

80

85

90

100

5 10 15 20

1435A G02

25 30

95ILOAD= 1A

ILOAD= 100mA

VOUT= 5V

Efficiency vs Input VoltageVOUT= 5V

Load Regulation

LOAD CURRENT (A)

0

VOUT(%)

0

0.5 1.0 1.5 2.0

1435A G05

2.5 3.0

0.25

0.50

0.75

1.00

1.25

1.50

RSENSE= 0.033

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LTC1435A


Input Supply and ShutdownCurrent vs Input Voltage

INPUT VOLTAGE (V)

00

SUPPLYCURRENT(mA)

SHUTDOWNCURRENT(A)

0.5

1.0

1.5

2.0

2.5

0

20

40

60

80

100

5 10 15 20

1435A G07

25 30

VOUT= 3.3VEXTVCC= OPEN

VOUT= 5VEXTVCC= VOUT

SHUTDOWN

EXTVCCSwitch Dropvs INTVCCLoad Current

INTVCCLOAD CURRENT (mA)

0

EXTVCCINTVCC(mV)

120

160

200

16

1435A G09

80

40

100

140

180

60

20

04 8 122 186 10 14 20

55C

25C

70C

INTVCCLOAD CURRENT (mA)

0

INTVCC(%)

0

0.3

20

1435A G08

0.3

0.55 10

70C

25C

15

0.5VEXTVCC= 0V

INTVCCRegulationvs INTVCC Load Current

Normalized Oscillator Frequencyvs Temperature

TEMPERATURE (C)

40

FREQUENCY(%)

5

10

35 85

1435A G10

fO

15 10 60 110 135

5

10

RUN/SS Pin Currentvs Temperature

TEMPERATURE (C)

400

RUN/SSCURRENT(A)

1

2

3

4

15 10 35 60

1435A G11

85 110 135

SFB Pin Current vs Temperature

TEMPERATURE (C)

40

SFBCURRENT(A)

0.50

0.25

0

35 85

1435A G12

0.75

1.00

15 10 60 110 135

1.25

1.50

TEMPERATURE (C)

40146

CURRENTSENSETHRESHOLD(mV)

148

150

152

154

15 10 35 60

1435A G13

85 110 135

Transient ResponseTransient ResponseMaximum Current SenseThreshold Voltage vs Temperature

VOUT50mV/DIV

ILOAD = 50mA to 1A 1435A G14 ILOAD = 1A to 3A 1435A G15

VOUT50mV/DIV

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LTC1435A


Soft Start: Load Current vs TimeBurst Mode Operation

PINFUNCTIONSUUU

ever EXTVCCis higher than 4.7V. See EXTVCCconnectionin Applications Information section. Do not exceed 10V onthis pin. Connect to VOUTif VOUT5V.

PGND (Pin 10):Driver Power Ground. Connects to sourceof bottom N-channel MOSFET and the () terminal of CIN.

BG (Pin 11): High Current Gate Drive for BottomN-Channel MOSFET. Voltage swing at this pin is fromground to INTVCC.

INTVCC (Pin 12):Output of the Internal 5V Regulator and

EXTVCCSwitch. The driver and control circuits are pow-ered from this voltage. Must be closely decoupled to powerground with a minimum of 2.2F tantalum or electrolyticcapacitor.

VIN (Pin 13):Main Supply Pin. Must be closely decoupledto the ICs signal ground pin.

SW (Pin 14):Switch Node Connection to Inductor. Volt-age swing at this pin is from a Schottky diode (external)voltage drop below ground to VIN.

BOOST (Pin 15):Supply to Topside Floating Driver. Thebootstrap capacitor is returned to this pin. Voltage swingat this pin is from INTVCC to VIN+ INTVCC.

TG (Pin 16):High Current Gate Drive for Top N-ChannelMOSFET. This is the output of a floating driver with avoltage swing equal to INTVCC superimposed on theswitch node voltage SW.

COSC (Pin 1): External capacitor COSC from this pin toground sets the operating frequency.

RUN/SS (Pin 2): Combination of Soft Start and RunControl Inputs. A capacitor to ground at this pin sets theramp time to full current output. The time is approximately0.5s/F. Forcing this pin below 1.3V causes the device tobe shut down. In shutdown all functions are disabled.

ITH (Pin 3): Error Amplifier Compensation Point. Thecurrent comparator threshold increases with this control

voltage. Nominal voltage range for this pin is 0V to 2.5V.SFB (Pin 4):Secondary Winding Feedback Input. Nor-mally connected to a feedback resistive divider from thesecondary winding. This pin should be tied to: ground toforce continuous operation; INTVCC in applications thatdont use a secondary winding; and a resistive divider fromthe output in applications using a secondary winding.

SGND (Pin 5): Small-Signal Ground. Must be routedseparately from other grounds to the () terminal of COUT.

VOSENSE(Pin 6):Receives the feedback voltage from an

external resistive divider across the output.

SENSE(Pin 7):The () Input to the Current Comparator.

SENSE+(Pin 8):The (+) Input to the Current Comparator.Built-in offsets between SENSE and SENSE+ pins inconjunction with RSENSEset the current trip thresholds.

EXTVCC (Pin 9):Input to the Internal Switch Connected toINTVCC. This switch closes and supplies VCC power when-

ILOAD = 50mA 1435A G16

VOUT20mV/DIV

VITH200mV/DIV

1435A G17

RUN/SS5V/DIV

INDUCTORCURRENT

1A/DIV

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LTC1435A

FUNCTIONAL DIAGRAUU W

+

+

+

+

SWITCHLOGIC

DROPOUTDET

0.6V

1.19V

QS

R

I1

INTVCC

4k

+

8k

180k

30k

+

+

EA

SHUTDOWN

6V

3A

1.19V

1.28V

+4.8V

VIN

SHUTDOWN

1.19VREF

CSS CC

RCRUN/SS2 ITH3

DFB*

SENSE+ 78 SENSE EXTVCC9

+

R2

VOSENSE

OV

1A

OSC

PGND

10

BG

INTVCC

12

SW

14

TG

16

BOOST

SGNDVIN13SFB4COSC1

RSENSE

+

+

VSEC

CSEC

VOUT

COUT

D1

CB

DB

INTVCC

+CIN

VIN

COSC

5VLDOREG

RUNSOFT

START

1435A FD

VFB

R1

I2

gm= 1m

* FOLDBACK CURRENT LIMITING OPTION

5

6

11

15

(Refer to Functional Diagram)OPERATIONU

Main Control Loop

The LTC1435A uses a constant frequency, current modestep-down architecture. During normal operation, the top

MOSFET is turned on each cycle when the oscillator setsthe RS latch, and turned off when the main current com-parator I1resets the RS latch. The peak inductor current atwhich I1resets the RS latch is controlled by the voltage onthe ITH pin , which is the output of error amplifier EA. TheVOSENSEpin, described in the Pin Functions section, allowsEA to receive an output feedback voltage VFBfrom an ex-ternal resistive divider. When the load current increases,it causes a slight decrease in VFBrelative to the 1.19V ref-

erence, which in turn causes the ITHvoltage to increase untilthe average inductor current matches the new load current.While the top MOSFET is off, the bottom MOSFET is turned

on until either the inductor current starts to reverse, asindicated by current comparator I2, or the beginning of thenext cycle.

The top MOSFET driver is biased from floating bootstrapcapacitor CB, which normally is recharged during each offcycle. However, when VINdecreases to a voltage close toVOUT, the loop may enter dropout and attempt to turn onthe top MOSFET continuously. The dropout detector countsthe number of oscillator cycles that the top MOSFET remains

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LTC1435A

(Refer to Functional Diagram)OPERATIONU

on and periodically forces a brief off period to allow CBtorecharge.

The main control loop is shut down by pulling the RUN/SSpin low. Releasing RUN/SS allows an internal 3A currentsource to charge soft start capacitor CSS. When CSSreaches1.3V, the main control loop is enabled with the ITHvoltageclamped at approximately 30% of its maximum value. AsCSScontinues to charge, ITHis gradually released allowingnormal operation to resume.

Comparator OV guards against transient overshoots> 7.5% by turning off the top MOSFET and keeping it offuntil the fault is removed.

Low Current OperationThe LTC1435A is capable of Burst Mode operation in whichthe external MOSFETs operate intermittently based on loaddemand. The transition to low current operation beginswhen comparator I2detects current reversal and turns offthe bottom MOSFET. If the voltage across RSENSEdoes notexceed the hysteresis of I2(approximately 20mV) for onefull cycle, then on following cycles the top and bottom drivesare disabled. This continues until an inductor current peakexceeds 20mV/RSENSEor the ITHvoltage exceeds 0.6V,

either of which causes drive to be returned to the TG pinon the next cycle.

Two conditions can force continuous synchronous opera-tion, even when the load current would otherwise dictatelow current operation. One is when the common modevoltage of the SENSE+and SENSE pins is below 1.4V andthe other is when the SFB pin is below 1.19V. The lattercondition is used to assist in secondary winding regulationas described in the Applications Information section.

INTVCC/EXTVCCPower

Power for the top and bottom MOSFET drivers and mostof the other LTC1435A circuitry is derived from the INTVCC

pin. The bottom MOSFET driver supply pin is internallyconnected to INTVCCin the LTC1435A. When the EXTVCCpin is left open, an internal 5V low dropout regulatorsupplies INTVCCpower. If EXTVCCis taken above 4.8V, the5V regulator is turned off and an internal switch is turnedon to connect EXTVCCto INTVCC. This allows the INTVCCpower to be derived from a high efficiency external sourcesuch as the output of the regulator itself or a secondarywinding, as described in the Applications Informationsection.

APPLICATIONS INFORMATIONWU UU

The basic LTC1435A application circuit is shown in Figure1, High Efficiency Step-Down Converter. External compo-nent selection is driven by the load requirement and beginswith the selection of RSENSE. Once RSENSEis known, COSCand L can be chosen. Next, the power MOSFETs and D1 areselected. Finally, CINand COUTare selected. The circuitshown in Figure 1 can be configured for operation up to aninput voltage of 28V (limited by the external MOSFETs).

RSENSESelection for Output Current

RSENSEis chosen based on the required output current. TheLTC1435A current comparator has a maximum thresholdof 150mV/RSENSEand an input common mode range ofSGND to INTVCC. The current comparator threshold setsthe peak of the inductor current, yielding a maximum av-erage output current IMAXequal to the peak value less halfthe peak-to-peak ripple current IL.

Allowing a margin for variations in the LTC1435A andexternal component values yields:

R mV

ISENSE

MAX

=100

The LTC1435A works well with RSENSEvalues 0.005.

COSCSelection for Operating Frequency

The LTC1435A uses a constant frequency architecture withthe frequency determined by an external oscillator capaci-tor COSC. Each time the topside MOSFET turns on, thevoltage COSCis reset to ground. During the on-time, COSCis charged by a fixed current. When the voltage on the ca-pacitor reaches 1.19V, COSCis reset to ground. The processthen repeats.

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LTC1435A


The value of COSCis calculated from the desired operatingfrequency:

C pFOSC( ) =

1.37(10 )Frequency (kHz)

4 11

A graph for selecting COSCvs frequency is given in Figure2. As the operating frequency is increased the gate chargelosses will be higher, reducing efficiency (see EfficiencyConsiderations). The maximum recommended switchingfrequency is 400kHz.

greater core losses. A reasonable starting point for settingripple current is IL= 0.4(IMAX). Remember, the maximumILoccurs at the maximum input voltage.

The inductor value also has an effect on low current opera-tion. The transition to low current operation begins whenthe inductor current reaches zero while the bottom MOSFETis on. Lower inductor values (higher IL) will cause this tooccur at higher load currents, which can cause a dip inefficiency in the upper range of low current operation. InBurst Mode operation, lower inductance values will causethe burst frequency to decrease.

The Figure 3 graph gives a range of recommended induc-tor values vs operating frequency and VOUT.

OPERATING FREQUENCY (kHz)

COSCVALUE(pF)

300

250

200

150

100

50

0100 200 300 400

1435A F02

5000

Figure 2. Timing Capacitor Value

Inductor Value CalculationThe operating frequency and inductor selection are inter-related in that higher operating frequencies allow the useof smaller inductor and capacitor values. So why wouldanyone ever choose to operate at lower frequencies withlarger components? The answer is efficiency. A higherfrequency generally results in lower efficiency because ofMOSFET gate charge losses. In addition to this basic trade-off, the effect of inductor value on ripple current and lowcurrent operation must also be considered.

The inductor value has a direct effect on ripple current. Theinductor ripple current ILdecreases with higher induc-tance or frequency and increases with higher VINor VOUT:

If L

V V

VL OUT

OUT

IN=

( )( )

11

Accepting larger values of ILallows the use of low induc-tances, but results in higher output voltage ripple and Kool Mis a registered trademark of Magnetics, Inc.

For low duty cycle, high frequency applications where therequired minimum on-time,

t V

V fON MIN

OUT

IN MAX( )

( )

,=( )( )

is less than 350ns, there may be further restrictions on theinductance to ensure proper operation. See Minimum On-Time Considerations section for more details.

Inductor Core Selection

Once the value for L is known, the type of inductor must beselected. High efficiency converters generally cannot affordthe core loss found in low cost powdered iron cores, forc-ing the use of more expensive ferrite, molypermalloy or KoolMcores. Actual core loss is independent of core size for

OPERATING FREQUENCY (kHz)

00

INDUCTORVALUE(H)

10

20

30

40

60

50 100 150 200

1435A F03

250 300

50

VOUT= 5.0VVOUT= 3.3VVOUT2.5V

Figure 3. Recommended Inductor Values

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LTC1435A


a fixed inductor value, but it is very dependent on induc-tance selected. As inductance increases, core losses godown. Unfortunately, increased inductance requires more

turns of wire and therefore copper losses will increase.

Ferrite designs have very low core loss and are preferredat high switching frequencies, so design goals canconcentrate on copper loss and preventing saturation.Ferrite core material saturates hard, which means that in-ductance collapses abruptly when the peak design currentis exceeded. This results in an abrupt increase in inductorripple current and consequent output voltage ripple.Do notallow the core to saturate!

Molypermalloy (from Magnetics, Inc.) is a very good, low

loss core material for toroids, but it is more expensive thanferrite. A reasonable compromise from the same manufac-turer is Kool M. Toroids are very space efficient, especiallywhen you can use several layers of wire. Because theygenerally lack a bobbin, mounting is more difficult. How-ever, designs for surface mount are available which do notincrease the height significantly.

Power MOSFET and D1 Selection

Two external power MOSFETs must be selected for use withthe LTC1435A: an N-channel MOSFET for the top (main)switch and an N-channel MOSFET for the bottom (synchro-nous) switch.

The peak-to-peak gate drive levels are set by the INTVCCvoltage. This voltage is typically 5V during start-up (seeEXTVCCPin Connection). Consequently, logic level thresh-old MOSFETs must be used in most LTC1435A applications.The only exception is applications in which EXTVCCispowered from an external supply greater than 8V (must beless than 10V), in which standard threshold MOSFETs(VGS(TH)< 4V) may be used. Pay close attention to the BVDSS

specification for the MOSFETs as well; many of the logic levelMOSFETs are limited to 30V or less.

Selection criteria for the power MOSFETs include the ONresistance RDS(ON), reverse transfer capacitance CRSS, in-put voltage and maximum output current. When theLTC1435A is operating in continuous mode the duty cyclesfor the top and bottom MOSFETs are given by:

Main Switch Duty Cycle =V

V

Synchronous Switch Duty Cycle = V

OUT

IN

IN( )VVOUT

IN

The MOSFET power dissipations at maximum output cur-rent are given by:

P V

V I R

I C f

P V VV

I R

MAIN OUT

INMAX DS ON

MAX RSS

SYNC IN OUT

INMAX DS ON

= ( ) +( ) +

( ) ( )( )( )

= ( ) +( )

( )

( )

2

1 85

2

1

1

k VIN.

where is the temperature dependency of RDS(ON)and kis a constant inversely related to the gate drive current.

Both MOSFETs have I2R losses while the topsideN-channel equation includes an additional term for tran-sition losses, which are highest at high input voltages.For VIN < 20V the high current efficiency generally im-proves with larger MOSFETs, while for VIN > 20V thetransition losses rapidly increase to the point that the use

of a higher RDS(ON) device with lower CRSSactual pro-vides higher efficiency. The synchronous MOSFET lossesare greatest at high input voltage or during a short circuitwhen the duty cycle in this switch is nearly 100%. Referto the Foldback Current Limiting section for further appli-cations information.

The term (1 + ) is generally given for a MOSFET in the formof a normalized RDS(ON)vs Temperature curve, but= 0.005/C can be used as an approximation for lowvoltage MOSFETs. CRSSis usually specified in the MOSFET

characteristics. The constant k = 2.5 can be used to esti-mate the contributions of the two terms in the main switchdissipation equation.

The Schottky diode D1 shown in Figure 1 conducts duringthe dead-time between the conduction of the two largepower MOSFETs. This prevents the body diode of the bot-tom MOSFET from turning on and storing charge during thedead-time, which could cost as much as 1% in efficiency.A 1A Schottky is generally a good size for 3A regulators.

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LTC1435A


CINand COUTSelection

In continuous mode, the source current of the top

N-channel MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR inputcapacitor sized for the maximum RMS current must beused. The maximum RMS capacitor current is given by:

C requiredIN I IV V V

VRMS MAX

OUT IN OUT

IN

( )[ ]

1 2/

This formula has a maximum at V IN = 2VOUT , whereIRMS = IOUT/2. This simple worst-case condition is com-monly used for design because even significant deviations

do not offer much relief. Note that capacitor manufacturersripple current ratings are often based on only 2000 hoursof life. This makes it advisable to further derate the capaci-tor or to choose a capacitor rated at a higher temperaturethan required. Several capacitors may also be paralleled tomeet size or height requirements in the design. Alwaysconsult the manufacturer if there is any question.

The selection of COUTis driven by the required effectiveseries resistance (ESR). Typically, once the ESR require-ment is satisfied the capacitance is adequate for filtering.

The output ripple (VOUT) is approximated by:

V I ESRfC

OUT LOUT

+

1

4

where f = operating frequency, COUT= output capacitanceand IL= ripple current in the inductor. The output rippleis highest at maximum input voltage since ILincreaseswith input voltage. With IL= 0.4IOUT(MAX)the output ripplewill be less than 100mV at max VINassuming:

COUTrequired ESR < 2RSENSE

Manufacturers such as Nichicon, United Chemicon andSanyo should be considered for high performance through-hole capacitors. The OS-CON semiconductor dielectriccapacitor available from Sanyo has the lowest ESR(size)product of any aluminum electrolytic at a somewhathigher price. Once the ESR requirement for COUThas beenmet, the RMS current rating generally far exceeds theIRIPPLE(P-P)requirement.

In surface mount applications multiple capacitors may haveto be paralleled to meet the ESR or RMS current handlingrequirements of the application. Aluminum electrolytic and

dry tantalum capacitors are both available in surface mountconfigurations. In the case of tantalum, it is critical that thecapacitors are surge tested for use in switching powersupplies. An excellent choice is the AVX TPS series ofsurface mount tantalum, available in case heights rangingfrom 2mm to 4mm. Other capacitor types include SanyoOS-CON, Nichicon PL series and Sprague 593D and 595Dseries. Consult the manufacturer for other specific recom-mendations.

INTVCCRegulator

An internal P-channel low dropout regulator produces the5V supply that powers the drivers and internal circuitrywithin the LTC1435A. The INTVCCpin can supply up to15mA and must be bypassed to ground with a minimumof 2.2F tantalum or low ESR electrolytic. Good bypassingis necessary to supply the high transient currents requiredby the MOSFET gate drivers.

High input voltage applications, in which large MOSFETsare being driven at high frequencies, may cause the maxi-mum junction temperature rating for the LTC1435A to be

exceeded. The IC supply current is dominated by the gatecharge supply current when not using an output derivedEXTVCCsource. The gate charge is dependent on operat-ing frequency as discussed in the Efficiency Considerationssection. The junction temperature can be estimated by usingthe equations given in Note 1 of the Electrical Character-istics. For example, the LTC1435A is limited to less than17mA from a 30V supply:

TJ= 70C + (17mA)(30V)(100C/W) = 126C

To prevent maximum junction temperature from being

exceeded, the input supply current must be checked whenoperating in continuous mode at maximum VIN.

EXTVCCConnection

The LTC1435A contains an internal P-channel MOSFETswitch connected between the EXTVCCand INTVCCpins. Theswitch closes and supplies the INTVCCpower whenever theEXTVCCpin is above 4.8V, and remains closed until EXTVCCdrops below 4.5V. This allows the MOSFET driver and

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control power to be derived from the output during normaloperation (4.8V < VOUT< 9V) and from the internal regu-lator when the output is out of regulation (start-up, short

circuit). Do not apply greater than 10V to the EXTVCCpinand ensure that EXTVCC< VIN.

Significant efficiency gains can be realized by poweringINTVCCfrom the output, since the VINcurrent resulting fromthe driver and control currents will be scaled by a factor ofDuty Cycle/Efficiency. For 5V regulators this supply meansconnecting the EXTVCCpin directly to VOUT. However, for3.3V and other lower voltage regulators, additional circuitryis required to derive INTVCCpower from the output.

The following list summarizes the four possible connections

for EXTVCC:

1. EXTVCCleft open (or grounded). This will cause INTVCCto be powered from the internal 5V regulator resultingin an efficiency penalty of up to 10% at high input volt-ages.

2. EXTVCCconnected directly to VOUT. This is the normalconnection for a 5V regulator and provides the highestefficiency.

3. EXTVCCconnected to an output-derived boost network.For 3.3V and other low voltage regulators, efficiencygains can still be realized by connecting EXTVCCto anoutput-derived voltage which has been boosted togreater than 4.8V. This can be done with either the in-ductive boost winding as shown in Figure 4a or thecapacitive charge pump shown in Figure 4b. The chargepump has the advantage of simple magnetics.

4. EXTVCCconnected to an external supply. If an externalsupply is available in the 5V to 10V range (EXTVCCVIN),it may be used to power EXTVCCproviding it is compat-ible with the MOSFET gate drive requirements. When

driving standard threshold MOSFETs, the external sup-ply must always be present during operation to preventMOSFET failure due to insufficient gate drive.

Topside MOSFET Driver Supply (CB, DB)

An external bootstrap capacitor CBconnected to the Boostpin supplies the gate drive voltage for the topside MOSFET.Capacitor CBin the Functional Diagram is charged throughdiode DBfrom INTVCCwhen the SW pin is low. When the

Figure 4a. Secondary Output Loop and EXTVCCConnection

EXTVCC

VIN

TG

BG

PGND

LTC1435A

N-CH

N-CH

+CIN

VIN

+COUT

L1 RSENSEVOUT

+1F

1435A F04b

VN2222LL

BAT85 BAT85

BAT85

0.22F

SW

Figure 4b. Capacitive Charge Pump for EXTVCC

topside MOSFET is to be turned on, the driver places theCBvoltage across the gate source of the MOSFET. This en-hances the MOSFET and turns on the topside switch. Theswitch node voltage SW rises to VINand the Boost pin risesto VIN+ INTVCC. The value of the boost capacitor CBneedsto be 100 times greater than the total input capacitance ofthe topside MOSFET. In most applications 0.1F is ad-equate. The reverse breakdown on DBmust be greater than

VIN(MAX).

Output Voltage Programming

The output voltage is set by a resistive divider accordingto the following formula:

V VR

RV VOUT OUT= +

1 19 1

2

1119. , .

R6

R5

EXTVCC

SFB

SGND

VIN

TG

BG

PGND

LTC1435A

N-CH

N-CH

+CIN

VIN

1N4148

+1F

+COUT

VSEC

L11:N

RSENSEVOUT

OPTIONALEXT VCC

CONNECTION5V VSEC9V

1435A F04a

SW

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The external resistive divider is connected to the output asshown in Figure 5 allowing remote voltage sensing.

VOSENSE

LTC1435A R1

R2

1435A F05

100pF

1.19V VOUT 9V

SGND

Figure 5. Setting the LTC1435A Output Voltage

Run/ Soft Start Function

The RUN/SS pin is a dual purpose pin that provides the softstart function and a means to shut down the LTC1435A. Soft

start reduces surge currents from VINby gradually increas-ing the internal current limit.Power supply sequencingcanalso be accomplished using this pin.

An internal 3A current source charges up an externalcapacitor CSS. When the voltage on RUN/SS reaches 1.3Vthe LTC1435A begins operating. As the voltage on RUN/SScontinues to ramp from 1.3V to 2.4V, the internal currentlimit is also ramped at a proportional linear rate. The cur-rent limit begins at approximately 50mV/RSENSE(at VRUN/SS= 1.3V) and ends at 150mV/RSENSE(VRUN/SS> 2.7V). The

output current thus ramps up slowly, charging the outputcapacitor. If RUN/SS has been pulled all the way to groundthere is a delay before starting of approximately 500ms/F,followed by an additional 500ms/F to reach full current.

tDELAY= 5(105)CSSSeconds

Pulling the RUN/SS pin below 1.3V puts the LTC1435A intoa low quiescent current shutdown (IQ< 25A). This pin canbe driven directly from logic as shown in Figure 6. DiodeD1 in Figure 6 reduces the start delay but allows CSSto rampup slowly for the soft start function; this diode and CSScan

be deleted if soft start is not needed. The RUN/SS pin hasan internal 6V Zener clamp (See Functional Diagram).

Foldback Current Limiting

As described in Power MOSFET and D1 Selection, the worst-case dissipation for either MOSFET occurs with a short-circuited output, when the synchronous MOSFET conductsthe current limit value almost continuously. In most appli-cations this will not cause excessive heating, even forextended fault intervals. However, when heat sinking is ata premium or higher RDS(ON) MOSFETs are being used,foldback current limiting should be added to reduce thecurrent in proportion to the severity of the fault.

Foldback current limiting is implemented by adding diodeDFBbetween the output and the ITHpin as shown in theFunctional Diagram. In a hard short (VOUT= 0V) the cur-

rent will be reduced to approximately 25% of the maximumoutput current. This technique may be used for all applica-tions with regulated output voltages of 1.8V or greater.

SFB Pin Operation

When the SFB pin drops below its ground referenced 1.19Vthreshold, continuous mode operation is forced. In continu-ous mode, the large N-channel main and synchronousswitches are used regardless of the load on the main output.

In addition to providing a logic input to force continuous

synchronous operation, the SFB pin provides a means toregulate a flyback winding output. Continuous synchronousoperation allows power to be drawn from the auxiliarywindings without regard to the primary output load. The SFBpin provides a way to force continuous synchronous op-eration as needed by the flyback winding.

The secondary output voltage is set by the turns ratio of thetransformer in conjunction with a pair of external resistorsreturned to the SFB pin as shown in Figure 4a. The second-ary regulated voltage, VSEC, in Figure 4a is given by:

V N V RR

SEC OUT +( ) > +

1 1 19 1 6

5.

where N is the turns ratio of the transformer and VOUTisthe main output voltage sensed by VOSENSE.

Minimum On-Time Considerations

Minimum on-time, tON(MIN), is the smallest amount of timethat the LTC1435A is capable of turning the top MOSFET

1435 F06

CSS

D1

3.3V OR 5V RUN/SS

CSS

RUN/SS

Figure 6. RUN/SS Pin Interfacing

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on and off again. It is determined by internal timing delaysand the gate charge required to turn on the top MOSFET.Low duty cycle applications may approach this minimum

on-time limit. If the duty cycle falls below what can beaccommodated by the minimum on-time, the LTC1435Awill begin to skip cycles. The output voltage will continueto be regulated, but the ripple current and ripple voltage willincrease. Therefore this limit should be avoided.

The minimum on-time for the LTC1435A in a properlyconfigured application is less than 300ns but increases atlow ripple current amplitudes (see Figure 7). If an appli-cation is expected to operate close to the minimum on-timelimit, an inductor value must be chosen that is low enough

to provide sufficient ripple amplitude to meet the minimumon-time requirement. To determine the proper value, usethe following procedure:

1. Calculate on-time at maximum supply, tON(MIN)=(1/f)(VOUT/VIN(MAX)).

2. Use Figure 7 to obtain the peak-to-peak inductor ripplecurrent as a percentage of IMAXnecessary to achieve thecalculated tON(MIN).

3. Ripple amplitude IL(MIN)= (% from Figure 7)(IMAX)where IMAX= 0.1/RSENSE.

4. LMAX= t V V

ION MIN

IN MAX OUT

L MIN( )

( )

( )

Choose an inductor less than or equal to the calculated LMAXto ensure proper operation.

Because of the sensitivity of the LTC1435A current com-parator when operating close to the minimum on-time limit,it is important to prevent stray magnetic flux generated by

the inductor from inducing noise on the current sense re-sistor, which may occur when axial type cores are used. Byorienting the sense resistor on the radial axis of the induc-tor (see Figure 8), this noise will be minimized.

Figure 7. Minimum On-Time vs Inductor Ripple Current

INDUCTOR RIPPLE CURRENT (% OF IMAX)

0200

MINIMUM

ON-T

IME(ns)

250

300

350

400

RECOMMENDEDREGION FOR MIN

ON-TIME ANDMAX EFFICIENCY

10 20 30 40

1435A F07

50 60 70

L

INDUCTOR

1435A F08

Figure 8. Allowable Inductor/RSENSELayout Orientations

Efficiency Considerations

The efficiency of a switching regulator is equal to the out-put power divided by the input power times 100%. It is oftenuseful to analyze individual losses to determine what islimiting the efficiency and which change would produce themost improvement. Efficiency can be expressed as:

Efficiency = 100% (L1 + L2 + L3 + ...)

where L1, L2, etc. are the individual losses as a percentage

of input power.Although all dissipative elements in the circuit producelosses, four main sources usually account for most of thelosses in LTC1435A circuits. LTC1435A VINcurrent, INTVCCcurrent, I2R losses, and topside MOSFET transition losses.

1. The VINcurrent is the DC supply current given in theelectrical characteristics which excludes MOSFET driverand control currents. VINcurrent results in a small(< 1%) loss which increases with VIN.

2. INTVCCcurrent is the sum of the MOSFET driver and

control currents. The MOSFET driver current results fromswitching the gate capacitance of the power MOSFETs.Each time a MOSFET gate is switched from low to highto low again, a packet of charge dQ moves from INTVCCto ground. The resulting dQ/dt is a current out of INTVCCthat is typically much larger than the control circuit cur-rent. In continuous mode, IGATECHG= f(QT + QB), whereQTand QBare the gate charges of the topside and bot-tom side MOSFETs.

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By powering EXTVCCfrom an output-derived source, theadditional VIN current resulting from the driver andcontrol currents will be scaled by a factor of

Duty Cycle/Efficiency. For example, in a 20V to 5V ap-plication, 10mA of INTVCCcurrent results in approxi-mately 3mA of VINcurrent. This reduces the midcurrentloss from 10% or more (if the driver was powered di-rectly from VIN) to only a few percent.

3. I2R losses are predicted from the DC resistances of theMOSFET, inductor and current shunt. In continuousmode the average output current flows through L andRSENSE, but is chopped between the topside mainMOSFET and the synchronous MOSFET. If the two

MOSFETs have approximately the same RDS(ON), thenthe resistance of one MOSFET can simply be summedwith the resistances of L and RSENSEto obtain I

2Rlosses. For example, if each RDS(ON)= 0.05,RL= 0.15, and RSENSE= 0.05, then the total resis-tance is 0.25. This results in losses ranging from 3%to 10% as the output current increases from 0.5A to2A. I2R losses cause the efficiency to drop at high outputcurrents.

4. Transition losses apply only to the topside MOSFET(s),and only when operating at high input voltages (typically

20V or greater). Transition losses can be estimated from:

Transition Loss = 2.5 (VIN)1.85(IMAX)(CRSS)(f)

Other losses, including CINand COUTESR dissipativelosses, Schottky conduction losses during dead-time,and inductor core losses, generally account for less than2% total additional loss.

Checking Transient Response

The regulator loop response can be checked by looking atthe load transient response. Switching regulators take sev-eral cycles to respond to a step in DC (resistive) load cur-rent. When a load step occurs, VOUTimmediately shifts byan amount equal to (ILOAD)(ESR), where ESR is the ef-fective series resistance of COUT. ILOADalso begins tocharge or discharge COUTwhich generates a feedback errorsignal. The regulator loop then acts to return VOUTto itssteady-state value. During this recovery time VOUTcan bemonitored for overshoot or ringing, which would indicatea stability problem. The ITHexternal components shown in

the Figure 1 circuit will provide adequate compensation formost applications.

A second, more severe transient is caused by switching inloads with large (>1F) supply bypass capacitors. Thedischarged bypass capacitors are effectively put in parallelwith COUT, causing a rapid drop in VOUT. No regulator candeliver enough current to prevent this problem if the loadswitch resistance is low and it is driven quickly. The onlysolution is to limit the rise time of the switch drive so thatthe load rise time is limited to approximately (25)(CLOAD).Thus a 10F capacitor would require a 250s rise time,limiting the charging current to about 200mA.

Automotive Considerations:Plugging into the Cigarette Lighter

As battery-powered devices go mobile, there is a naturalinterest in plugging into the cigarette lighter in order toconserve or even recharge battery packs during operation.But before you connect, be advised: you are plugging intothe supply from hell. The main battery line in an automo-bile is the source of a number of nasty potential transients,including load dump, reverse battery and double battery.

Load dump is the result of a loose battery cable. When thecable breaks connection, the field collapse in the alternatorcan cause a positive spike as high as 60V which takes severalhundred milliseconds to decay. Reverse battery is just whatit says, while double battery is a consequence of tow truckoperators finding that a 24V jump start cranks cold enginesfaster than 12V.

The network shown in Figure 9 is the most straightforwardapproach to protect a DC/DC converter from the ravagesof an automotive battery line. The series diode preventscurrent from flowing during reverse battery, while thetransient suppressor clamps the input voltage during load

dump. Note that the transient suppressor should not

Figure 9. Automotive Application Protection

1435A F09

50A IPKRATING

LTC1435ATRANSIENT VOLTAGE

SUPPRESSORGENERAL INSTRUMENT

1.5KA24A

VIN

12V

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LTC1435A


conduct during double battery operation, but must stillclamp the input voltage below breakdown of the converter.Although the LTC1435A has a maximum input voltage of

36V, most applications will be limited to 30V by theMOSFET BVDSS.

Design Example

As a design example, assume VIN= 12V(nominal), VIN=22V(max), VOUT = 1.6V, IMAX = 3A and f = 250kHz, RSENSEand COSCcan immediately be calculated:

RSENSE= 100mV/3A = 0.033

COSC = 1.37(104)/250 11 = 43pF

Referring to Figure 3, a 4.7H inductor falls within the rec-ommended range. To check the actual value of the ripplecurrent the following equation is used:

I Vf L

V

VL

OUT OUT

IN=

( )( )

1

The highest value of the ripple current occurs at the maxi-mum input voltage:

I V

kHz H

V

VL=

( )

=

1 6

250 4 71

1 6

221 3

.

.

.. A

The lowest duty cycle also occurs at maximum input volt-age. The on-time during this condition should be checkedto make sure it doesnt violate the LTC1435As minimumon-time and cause cycle skipping to occur. The required on-time at VIN(MAX)is:

t V

V f

V

V kHznsON MIN

OUT

IN MAX( )

( )

.=

( )( )=

( )( )=

1 6

22 250291

TheILwas previously calculated to be 1.3A, which is 43%

of IMAX. From Figure 7, the LTC1435A minimum on-timeat 43% ripple is about 235ns. Therefore, the minimum on-time is sufficient and no cycle skipping will occur.

The power dissipation on the topside MOSFET can be easilyestimated. Choosing a Siliconix Si4412DY results in:RDS(ON)= 0.042, CRSS= 100pF. At maximum input volt-age with T(estimated) = 50C:

PV

VC C

V A pF kHz mW

MAIN= ( ) +( ) ( )[ ]( )

+ ( ) ( )( )( )=

1 6

223 1 0 005 50 25 0 042

2 5 22 3 100 250 88

2

1 85

.. .

.

.

The most stringent requirement for the synchronousN-channel MOSFET occurs when VOUT= 0 (i.e. short cir-cuit). In this case the worst-case dissipation rises to:

P I RSYNC SC AVG DS ON=( ) +( )( ) ( )2

1

With the 0.033sense resistor ISC(AVG)= 4A will result,increasing the Si4412DY dissipation to 950mW at a die tem-perature of 105C.

CINis chosen for an RMS current rating of at least 1.5A attemperature. COUTis chosen with an ESR of 0.03for lowoutput ripple. The output ripple in continuous mode will behighest at the maximum input voltage. The output voltageripple due to ESR is approximately:

VORIPPLE= RESR(IL) = 0.03(1.3A) = 39mVP-P

PC Board Layout Checklist

When laying out the printed circuit board, the followingchecklist should be used to ensure proper operation of the

LTC1435A. These items are also illustrated graphically inthe layout diagram of Figure 10. Check the following in yourlayout:

1. Are the signal and power grounds segregated? TheLTC1435A signal ground pin must return to the () plateof COUT. The power ground connects to the source of thebottom N-channel MOSFET, anode of the Schottky di-ode, and () plate of CIN, which should have as short leadlengths as possible.

2. Does the VOSENSEpin connect directly to the feedback

resistors? The resistive divider R1, R2 must be con-nected between the (+) plate of COUTand signal ground.The 100pF capacitor should be as close as possible tothe LTC1435A.

3. Are the SENSEand SENSE+leads routed together withminimum PC trace spacing? The filter capacitor betweenSENSE+and SENSE should be as close as possible tothe LTC1435A.

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4. Does the (+) plate of CINconnect to the drain of thetopside MOSFET(s) as closely as possible? This capaci-tor provides the AC current to the MOSFET(s).

5. Is the INTVCCdecoupling capacitor connected closelybetweenINTVCCand the power ground pin? This capaci-tor carries the MOSFET driver peak currents.

6. Keep the switching node SW away from sensitive small-signal nodes. Ideally the switch node should be placedat the furthest point from the LTC1435A.

7. SGND should be exclusively used for grounding exter-nal components on COSC, ITH, VOSENSEand SFB pins.

8. If operating close to the minimum on-time limit, is thesense resistor oriented on the radial axis of the induc-tor? See Figure 8.

COSC TG1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

LTC1435A

100pF

1000pF

CC1

CC2

RC

COSC

CSS

+4.7F

DB D1CB

0.1F

CIN

M1

M2

L1

R1

R2 RSENSE

+

+

VOUT

VIN

1435A F10

BOLD LINES INDICATE

HIGH CURRENT PATHS

+COUT

+

RUN/SS

ITH

SFB

SGND

VOSENSE

SENSE

SENSE+

SW

VIN

INTVCC

BG

PGND

EXTVCC

BOOST

ITH

TG

1

2

3

5

6

10

11

15

12

14

16

13

7 8

COSC43pF

CC1000pF

CC2220pF

50pF

RC10k

CSS0.1F

SENSE SENSE+

LTC1435A

1000pF

+4.7F

CF0.1F

DBCMDSH-3

0.22F

M2Si4410

D1MBRS140T3

L13.3H

M1Si4410 RSENSE

0.015

4.7

+ CIN10F30V2

VIN4.5V TO 22V

+ COUT820F4V2

VOUT

1.3V TO 2.0V7A

PGND

BG

COSC VIN

VOSENSE

BOOST

1435A TA07

SGND

RUN/SS

SW

INTVCC

VID0 1 2 3 GND

FROM P

SENSE

LTC1706-19

FB

36

42187

5

VCC

Intel Mobile CPU VID Power Converter

Figure 10. LTC1435A Layout Diagram

TYPICAL APPLICATIONSU

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LTC1435A


COSC

RUN/SS

ITH

SFB

SGND

VOSENSE

SENSE

SENSE+

TG

BOOST

SW

VIN

INTVCC

BG

PGND

EXTVCC

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

LTC1435A

1000pF

CC1470pF

CC251pF

100pF

RC10k

COSC68pF

CSS0.1F

+4.7F

T1: DALE LPE6562-A236

CMDSH-3 0.1F

0.01F

M1Si4412DY

M2Si4412DY

1435A TA04

100

100

+ CIN22F35V2

MBRS140T3

T110H1:1.42

RSENSE0.033

R135.7k1%

R220k1%

+ COUT100F10V

2

SGND

VOUT5V/3.5A

VIN5.4V TO 28V

+ CSEC3.3F35V

4.7k

IRLL014

11.3k1%

100k1%

VOUT212V120mA

Dual Output 5V and Synchronous 12V Application

3.3V/4.5A Converter with Foldback Current Limiting

COSC

RUN/SS

ITH

SFB

SGND

VOSENSE

SENSE

SENSE+

TG

BOOST

SW

VIN

INTVCC

BG

PGND

EXTVCC

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

LTC1435A

1000pF

CC1330pF

CC251pF

100pF

RC10k

COSC68pF

CSS0.1F

+4.7F

CMDSH-3 0.1F

M1Si4410DY

M2Si4410DY

1435A TA01

INTVCC

+ CIN22F35V2

MBRS140T3

L110H

OPTIONAL:CONNECT TO 5V

RSENSE0.025

R135.7k1%

R220k1%

+ COUT100F10V2

SGND(PIN 5)

VOUT3.3V/4.5A

VIN4.5V TO 28V

100pF

IN4148

ITHPIN 3

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Constant-Current/Constant-Voltage High Efficiency Battery Charger

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

TG

BOOST

SW

VIN

INTVCC

BG

PGND

EXTVCC

COSC

RUN/SS

ITH

SFB

SGND

VOSENSE

SENSE

SENSE+

LTC1435A

1

2

3

4

8

7

6

5

AVG

PROG

VCC

PIN

SENSE

IOUT

GND

NIN

LT1620

C1156pF

C9100pF

D2C141000pF

C10100pF

C150.1F

C60.33F

C50.1F

C130.033F

E3SHDN

R51k

C160.33F

E4IPROG

*CONSULT CAPACITOR MANUFACTURER FOR RECOMMENDEDESR RATING FOR CONTINUOUS 4A OPERATION

E5GND

C180.1F

R610k1%

RPROG

C170.01F

C120.1F

C40.1F

E3GND

E1V

IN

D1

C74.7F16V

+

L127H

C322F35V

R10.025

E6BATT

Q1Si4412DY

Q2Si4412DY

E7GND

C8100pF

R21M

0.1%

R476.8k0.1%

JP1B

1435A TA06

+

R3105k0.1%

JP1A

C2*22F35V

R71.5M

C1*22F35V

+ +

Efficiency

BATTERY CHARGE CURRENT (A)

0

E

FFICIENCY(%)

90

95

100

4

1435A TA05

85

80

751 2 3 5

VIN= 24V VBATT= 16V

VBATT= 12V

VBATT= 6V

Current Programming Equation

IBATT=(IPROG)(R6) 0.04

10(R1)

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LTC1435A


Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

0.016 0.050

0.406 1.270

0.010 0.020

(0.254 0.508)45

0 8TYP

0.008 0.010

(0.203 0.254)

S16 0695

1 2 3 4 5 6 7 8

0.150 0.157**

(3.810 3.988)

16 15 14 13

0.386 0.394*

(9.804 10.008)

0.228 0.244

(5.791 6.197)

12 11 10 90.053 0.069

(1.346 1.752)

0.014 0.019

(0.355 0.483)

0.004 0.010

(0.101 0.254)

0.050

(1.270)TYP

DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASHSHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEADFLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

*

**

S Package

16-Lead Plastic Small Outline (Narrow 0.150)(LTC DWG # 05-08-1610)

G16 SSOP 1197

0.005 0.009(0.13 0.22)

0 8

0.022 0.037(0.55 0.95)

0.205 0.212**(5.20 5.38)

0.301 0.311(7.65 7.90)

1 2 3 4 5 6 7 8

0.239 0.249*(6.07 6.33)

14 13 12 11 10 915160.068 0.078(1.73 1.99)

0.002 0.008(0.05 0.21)

0.0256(0.65)BSC 0.010 0.015

(0.25 0.38)DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASHSHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEADFLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

*

**

G Package16-Lead Plastic SSOP (0.209)

(LTC DWG # 05-08-1640)

PACKAGE DESCRIPTIONU

Dimensions in inches (millimeters) unless otherwise noted.

Dual Output 5V and 12V Application

COSC

RUN/SS

ITH

SFB

SGND

VOSENSE

SENSE

SENSE+

TG

BOOST

SW

VIN

INTVCC

BG

PGND

EXTVCC

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

LTC1435A

1000pF

CC1510pF

CC251pF

100pF

RC10k

COSC68pF

CSS0.1F

+4.7F

CMDSH-3 0.1F

M1IRF7403

M2IRF7403

1435A TA02

100

100

+ CIN22F35V2

MBRS140T3

T110H1:2.2

RSENSE0.033 R1

35.7k1%

R220k1%

+ COUT100F10V

2

SGND

VOUT5V/3.5A

VIN5.4V TO 28V

MBRS1100T3

+ CSEC3.3F25V

10k

T1: DALE LPE6562-A092

90.9k VOUT212V

24V

8/13/2019 1435af

20/20

LTC1435A

TYPICAL APPLICATIONU

Low Dropout 2.9V/3A Converter

COSC

RUN/SS

ITH

SFB

SGND

VOSENSE

SENSE

SENSE+

TG

BOOST

SW

VIN

INTVCC

BG

PGND

EXTVCC

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

LTC1435A

1000pF

CC1330pF

CC251pF

100pF

RC10k

COSC68pF

CSS0.1F

+4.7F

CMDSH-3 0.1F

M11/2 Si9802DY

M21/2 Si9802DY

1435A TA03

INTVCC

+ CIN22F35V2

MBRS140T3

L110H

OPTIONAL:CONNECT TO 5V

L1: SUMIDA CDRH125-10

RSENSE0.033

R135.7k1%

R224.9k1%

+ COUT100F10V

2

SGND

VOUT2.9V/3A

VIN3.5V TO 20V

100pF

PART NUMBER DESCRIPTION COMMENTS

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LTC1436A/LTC1436A-PLL/ High Efficiency Low Noise Synchronous Step-Down Full-Featured Single ControllerLTC1437A Switching Regulators

LTC1438/LTC1439 Dual High Efficiency, Low Noise, Synchronous Step-Down Full-Featured Dual ControllersSwitching Regulators

LT1510 Constant-Voltage/ Constant-Current Battery Charger 1.3A, Li-Ion, NiCd, NiMH, Pb-Acid ChargerLTC1538-AUX Dual High Efficiency, Low Noise, Synchronous Step-Down 5V Standby in Shutdown

Switching Regulator

LTC1539 Dual High Efficiency, Low Noise, Synchronous Step-Down 5V Standby in ShutdownSwitching Regulator

LTC1706-19 VID Voltage Programmer Creates a Programmable 1.3V to 2V Supply for IntelMobile PentiumII Processor When Used with theLTC1435A

Pentium is a registered trademark of Intel Corporation.

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