134 IEEE TRANSACTIONS ON COMPONENTS ......134 IEEE TRANSACTIONS ON COMPONENTS, PACKAGING AND MANUFACTURING TECHNOLOGY, VOL. 4, NO. 1, JANUARY 2014 A Comprehensive Investigation of

134 IEEE TRANSACTIONS ON COMPONENTS, PACKAGING AND MANUFACTURING TECHNOLOGY, VOL. 4, NO. 1, JANUARY 2014

A Comprehensive Investigation of a Common-ModeFilter for Gigahertz Differential Signals Using

Quarter-Wavelength ResonatorsGuang-Hwa Shiue, Member, IEEE, Che-Ming Hsu, Chi-Lou Yeh, and Cheng-Fu Hsu

Abstract— This paper presents a comprehensive investigationon an inexpensive and wideband common-mode noise suppressionfilter that uses a quarter-wavelength resonator. An equivalenttransmission line model is also used to evaluate the effectivenessof the proposed structure in a common-mode filter (CMF).Moreover, more important parameters, such as coupling coef-ficients are presented to investigate the characteristics of CMFunit. The performance of CMF is then characterized usingtwo coupling coefficients between two differential traces andbetween differential traces and a quarter-wavelength resonator.A simple design approach is also presented for a wideband CMF.Numerical results demonstrate that CMF with three differentlengths of quarter-wavelength resonators have rejection bandsof 3.94–8.74 GHz and 3.53–10.1 GHz at cutoff frequencies of−20 dB and −10 dB, respectively Time-domain common-modenoise is decreased by approximately 60%. Moreover, analysesof differential insertion loss and group delay indicate that thedifferential signals maintain sufficient signal integrity when thewideband CMF is used in single- and two-pair differentialinterconnects. Finally, CMF is validated by an eye diagram andby measurements of time/frequency-domain common-mode noiseand I/O cable common-mode current.

Index Terms— Common-mode filter (CMF), differentialsignals, eye diagram, multiple differential signaling pairs, openstub, quarter-wavelength resonator, signal integrity (SI).

I. INTRODUCTION

IN MODERN high-speed digital circuits, signals areconventionally transmitted by differential signaling [1].

Common applications include serial advanced technologyattachment III (SATA III/6 Gb/s), high definition multime-dia interface (HDMI/5 Gb/s), and universal serial bus (USB3.0/5 Gb/s) devices. Furthermore, high-speed digital circuitshave harmonics of tens of GHz (gigahertz). The benefits ofusing differential signals include improved SI, good EMIcancellation, and strong immunity to noise [2]. However, an

Manuscript received August 9, 2012; revised May 1, 2013 and June 9,2013; accepted August 3, 2013. Date of publication August 28, 2013; date ofcurrent version December 30, 2013. This work was supported by the NationalScience Council of China under Grant NSC 100-2221-E-033-078, GrantCYCU-EECS, and Grant CYCU-EECS-9903. Recommended for publicationby Associate Editor L.-T. Hwang upon evaluation of reviewers’ comments.

G.-H. Shiue and C.-F. Hsu are with the Electronic Engineering Department,Chung Yuan Christian University, Taoyuan 32023, Taiwan (e-mail:[email protected]; [email protected]).

C.-M. Hsu and C.-L. Yeh are with the Master Program in CommunicationEngineering, Chung Yuan Christian University, Taoyuan 32023, Taiwan(e-mail: [email protected]; [email protected]).

Color versions of one or more of the figures in this paper are availableonline at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TCPMT.2013.2278444

imbalance between differential signals and common modenoise can exacerbate EMI problems by generating common-mode current radiation through the I/O cables [1], [3]. There-fore, the maximum common-mode voltage must be specifiedfor high-speed GHz differential transmission systems, such asUSB 3.0. Recently, the mitigation of common-mode noise hasattracted substantial attention.

The bandwidths of common-mode choke and low-temperature co-fired ceramic (LTCC) filters used to suppresscommon-mode noise in differential signals are limited toseveral MHz [3], [4]. Since the frequency range of common-mode noise currently exceeds the GHz range, a CMF withseveral GHz bandwidths and a satisfactory SI performance inhigh-speed digital circuits is needed [5].

Recent common-mode filter designs include defectedground structure (DGS) filters [5] and electromagneticbandgap (EBG) filters [6]. Since the shielding metal planeunder a DGS structure greatly degrades CMF performance, theDGS structure cannot be implemented in multilayer structures.Although the use of an EBG structure enables application ofa CMF in a multilayer structure, many EBG units are neededto realize a CMF for a single pair of differential traces ona PCB scale. Another recently developed CMF solution is touse the differential signals that cross a planar EBG [7], [8]to suppress the harmonic components of the portion of theCM signal generated by skewed differential signals. Resonantfrequency must be decreased by increasing the planar EBGsize. Applications of the planar EBG are limited because itis a large and incomplete structure, and it is not a widebandfilter.

An inexpensive wideband CMF based on quarter-wavelength open stub resonators was recently proposed forfiltering high-speed digital differential signals [9]. The authorsused many cascading stopband filter units based on a quarter-wavelength openstub resonator to realize a wideband CMF.The special shape of the CMF also facilitates its imple-mentation in multilayer standard printed circuit board (PCB)structures and multiple differential signaling pairs. The eyediagram of the differential signals is not degraded within thewide bandwidth of the proposed CMF. However, the authorsin that work [9] did not describe these schemes in detail.

Therefore, this paper extends the concepts introducedin [9] by comprehensive studies of a CMF based onquarter-wavelength open-stub resonators. In this paper, thesignificantly additional investigations for the CMF are simply

2156-3950 © 2013 IEEE

SHIUE et al.: COMPREHENSIVE INVESTIGATION OF A CMF FOR GHz DIFFERENTIAL SIGNALS 135

summarized as follows: 1) an appropriate equivalent trans-mission line model is developed. The model is then used toshow how the proposed structure can be used to form a CMF.The CMF is then evaluated by measurements of its electricalfield distribution; 2) additional properties used to evaluatethe properties and performance of the CMF unit includecoupling coefficient between differential traces, capacitivecoupling coefficient between common-mode signal tracesand quarter-wavelength resonator, width/height of a quarter-wavelength resonator, and group delay; 3) a simple designapproach is presented for the wideband CMF; 4) reductionof time-domain common-mode noise by the wideband CMFis demonstrated; and 5) numerous measurement results arepresented to confirm the efficacy of the wideband CMF.

This paper is organized as follows. Section II presents atransmission line equivalent circuit model of the CMF. Theresults of a simulation of the transmission line equivalentcircuit model and full-wave simulator are presented to validatethe implementation of the CMF unit. The key parametersof the CMF are then investigated. Section III examines theimplementation of the wideband CMF by cascading severalCMF units The unique shape of the wideband CMF enablesits implementation in many differential lines pairs. Section IVverifies the performance of CMF by comparing simulated andmeasured results and by comparing measured eye diagramsand the common-mode current of the I/O cable. Section Vbriefly presents the conclusions of this paper.

II. COMMON-MODE FILTER UNIT

A. CMF Unit Structure

Fig. 1 shows the basic CMF, which incorporates an openstub, a quarter-wavelength resonator, and differential tracesin a three-layer structure [9]. Differential signals of coupledmicrostrip lines appear above the open stub on the top (first)layer. The trace of the open stub is located on the secondlayer while one end of the open stub is connected to thethird layer by a via. Since both the second and third layersare ground layers, the differential traces are broadside-coupledwith the open stub trace. In Fig. 1, the dimensions of the CMFstructure are denoted by Ws , S, h1, Wr , g, r , h2, and Lr ,and the dielectric constant of the structure is denoted by εr .The oddmode and evenmode characteristic impedances forSections 1 and 2 of differential traces are (Z0o1, Z0e1) and(Z0o, Z0e), respectively. The characteristic impedance of theopen stub is Z0q .

B. Design Concept and Equivalent Circuit Model

Fig. 2(a) presents the equivalent transmission line model ofthe CMF unit, which is shown in Fig. 1. Fig. 2(b) and (c)shows the odd mode and even mode equivalent circuit modelscorresponding to a structures with a perfect electrical wall(E-wall) and one with a perfect magnetic wall (H-wall),respectively, on the center line between the differential tracesshown in Fig. 1 [5], [12]. Fig. 2(b) shows that the differentialsignals propagate in odd mode and that the current returnpassing through the ground trace (i.e., the open stub trace)is relatively low. The open stub only slightly degrades the

Fig. 1. Configuration of common-mode filter unit in 3-D view.

(a)

(b)

(c)

Fig. 2. (a) Equivalent circuit model of the common-mode filter unit.Equivalent circuit models of (b) odd mode and (c) even mode.

differential signals. However, Fig. 2(c) shows that most ofthe return current of the even mode signal (or common-modenoise) passes through the ground plane and open stub, whichexplains the large effect of the open stub quarter-wavelengthresonator on common-mode noise [5]. A conventional quarter-wavelength openstub resonator can be used to implement anarrow-band bandstop filter [9]–[11]. Accordingly, Fig. 2(c)shows a narrow-band bandstop filter for common-mode noise(or an even-mode signal). The desired rejection frequency, orresonant frequency, of the CMF design determines the length(Lr ) of the quarter-wavelength open stub. The equation for the


Fig. 3. Comparison of full-wave simulation results for |Sdd21| and |Scc21|obtained by the CMF unit and by the equivalent circuit model.

length of the openstub quarter-wavelength resonator [10] is

f0 ≈ c

4(Lr + h2)√

εr(2n + 1) (1)

where c denotes the speed of light in free space; εr is thedielectric constant of the open stub, and n is the resonancenumber (n = 0, 1, 2, 3…). The grounded via of the open stubis modeled as a series inductance [13]

Lvia ≈ μ0h2

2π

⎡⎣ln

⎛⎝2h2

r+

√1 +

(2h2

r

)2⎞⎠

−√

1 +(

r

2h2

)2

+ r

2h2+ 1

4

⎤⎦. (2)

Consider a CMF unit in a multilayer PCB environment(Fig. 1) in which the geometrical parameters Wr , h1, 2r , h2,Lr , g, and Ws S are 16.5 mil, 4 mil, 16.5 mil, 20 mil, 270 mil,8 mil, 5 mil, and 6.5 mil, respectively. The dielectric constantεr and loss tangent of the substrate material are 4.4 and0.02, respectively. According to formula (2), inductance Lr isapproximately 0.175 nH. Additionally, the relevant character-istic impedance values Z0o1, Z0e1, Z0o, Z0e, and Z0q obtainedby Q3-D simulator [14] for the equivalent circuit (Fig. 2)are 66.2 �, 171 �, 51.5 �, 68 �, and 68.6 �, respectively.Notably, the use of mixed-mode S-parameters is increased todescribe the performance of differential interconnects in termsof pure, differential, common and mixed modes, common-to-differential mode, and differential-to-common mode [15].Fig. 3 shows the magnitudes of differential (|Sdd21|) and com-mon (|Scc21|) insertion losses obtained for the CMF unit byfull-wave simulator (HFSS [16]) and by the equivalent circuitmodel (ADS environment [17]). Notably, a 0.5 dB differentialinsertion loss occurs from low frequency to 10 GHz. Thecommon-mode noise is greatly reduced at frequencies nearthe fundamental resonant frequency (5 GHz) of the quarter-wavelength open-stub resonator. A good agreement betweenthe two simulation results is also noted.

Fig. 4 presents the HFSS full-wave simulation results for theelectrical field distribution of the open stub traces of a CMF

(a)

(b)

Fig. 4. Electrical field distribution of open stub traces of CMF unit atfrequency of 5 GHz for (a) odd mode (differential) and (b) even mode(common) signals.

Fig. 5. Comparison of |Scc21| values obtained in simulations of CMF unitswith quarter-length resonators of varying widths (Wr ).

unit at 5 GHz for odd mode (differential) and even mode (com-mon) signals. Fig. 4 confirms that most of the return currentof the common-mode signal passes through the open stub andsubstantially affects the common-mode signal, especially whenits frequency approximates the resonant frequency of 5 GHz.Note the negligible effect of the open stub on the differentialsignals.

C. Analysis of Important Parameters

Since the differences in the previous comparisons betweenthe equivalent circuit model and the full-wave simulation aresmall, the accuracy of further analyses of key CMF parameterswas enhanced by using HFSS full-wave simulator. Fig. 5shows the |Scc21| values obtained in simulations of a CMFunit with a quarter-wavelength resonator of varying widths(Wr ). The similar |Scc21| values obtained for widths (Wr ) of16.5 mil and 32.5 indicate that the return current area in theCMF structure (Wr = 16.5 mil) is sufficient for the common-mode signal. A width (Wr ) less than 16.5 mil decreases thestopband bandwidth and the band attenuation depth. Therefore,the width (Wr ) of the quarter-wavelength resonator with thefollowing simulated structure is set to 16.5 mil. Notably,the total width (2W + S) of the differential traces equalsthe width (16.5 mil) of the open stub trace. Therefore, afavorable performance is ensured by setting the minimumquarter-wavelength resonator width (Wr ) to 2W + S.


TABLE I

COUPLING COEFFICIENT (K ) AND CAPACITIVE COUPLING

COEFFICIENT (k′c) FOR CMF UNIT WITH DIFFERENTIAL

TRACES OF VARYING DIMENSIONS

The coupling coefficient (also called the coupling factor) isa convenient way to represent the coupling between circuits.The coupling coefficient (k) of the coupled microstrip lines(differential traces) is defined by [18]

k = Zeven − Zodd

Zeven + Zodd(3)

where Zeven and Zodd are the even-mode and odd-mode char-acteristic impedances, respectively. The capacitance matrixequation [14], [19] for the CMF unit of a three-conductorsystem in which all three ends (#1, #2, and #3) match asshown in Fig. 1 is

Qi = [Cij ] · Vj , i, j = 1, 2, 3. (4)

Consider a situation in which conductors #1 and #2 are drivenby common-mode signals with V2 = V1 and Q2 = Q1.The self-capacitance of the common-mode signals of differen-tial traces and the mutual capacitance between the common-mode signals of the differential traces and the trace of thequarter-wavelength resonator are easily calculated by Cs =(C11 + C22 + C12 + C21)/2 and Cm = (C13 + C23)/2,respectively. The capacitive coupling coefficient (k′

c) betweenthe common-mode signals of the differential traces and thetrace of the quarter-wavelength resonator is [20]

k ′c = Cm√

CsC33(5)

where C33 is the self-capacitance of the trace of the quarter-wavelength resonator.

Table I presents the coupling coefficient (k) and capacitivecoupling coefficient (k ′

c) of the CMF unit with various dimen-sions and differential traces of various dimensions. Fig. 6further compares the simulated |Scc21| values obtained forCMF units with differential traces of different coupling coef-ficients (k). Fig. 6 also shows that the coupling coefficient (k)correlates negatively with the stopband bandwidth and withthe band attenuation depth. However, the capacitive couplingcoefficient (k ′

c) of the CMF unit correlates positively with thestopband bandwidth and with the band attenuation. However,the capacitive coupling coefficient (k ′

c) of the CMF unitcorrelates positively with the stopband bandwidth and withthe band attenuation capacitive coupling coefficient (k ′

c). Thesmall variation in capacitive coupling coefficient (k ′

c) for theCMF unit results in a small variations in stopband bandwidthand in band attenuation depth (Fig. 6). Coupling coefficient(k) indicates the integrity of differential signals. However,coupling coefficient (k) of differential signals is negatively

Fig. 6. Comparison of simulated |Scc21| for CMF units with differentialtraces and varying coupling coefficients (k).

Fig. 7. Comparison of simulated |Scc21| for CMF units with varying h1.

associated with the performance of the CMF unit. Therefore,the coupling coefficient (k) for the differential signals is setto 0.12.

Fig. 7 compares the simulated |Scc21| values obtained forCMF units with varying h1. The inset in Fig. 7 presents thecoupling coefficient (k) and capacitive coupling coefficient(k ′

c) of the CMF unit with various h1. As h1 decreases,capacitive coupling coefficient (k ′

c), stopband bandwidth andband attenuation depth increase. Although h1 = 2 mil providesa better |Scc21| compared to all other values for h1 owing toPCB manufacturing limitations, h1 is set to 4 mil.

The favorable integrity of differential signals can be main-tained only if insertion loss |Sdd21| is small and if the groupvelocity in differential mode is not distorted by the quarter-wavelength resonator of the CMF unit [5]. Fig. 8 shows thesimulated group delay for the differential signal with theCMF unit (CMF board) and the solid ground plane (referenceboard). The group delay for the differential signals at thecentral frequency with and without the filter is approximately0.0409 and 0.0406 ns, respectively. The maximum differencewithin the designed stopband is only approximately 0.69%.

III. WIDEBAND CMF ANALYSIS

The performance of a wideband bandpass filter can beimproved by cascading several bandpass units [10], [11].Therefore, the proposed wideband CMF cascades several


Fig. 8. Comparison of simulated group delays for differential signals withand without the CMF unit.

Fig. 9. Comparison of |Scc21| values for a cascade of two different CMFunits with different gaps and individual CMF unit.

CMF units [9]. Fig. 9 shows the |Scc21| values obtained insimulations of two cascaded CMF units with different gaps(g = 4 mil, 8 mil, and 12 mil). Fig. 9 also presents thesimulated |Scc21| for different CMF units. The gap of 4 mil isseen to be too small for the common-mode signal because theband attenuation depth of |Scc21| is small. The gap betweenthe two CMF units must be large enough to allow the currentof the common-mode signal to return through the quarter-wavelength resonator. Fig. 9 shows that gap sizes of 8 miland 12 mil yield very similar |Scc21| values. Therefore, thegap between the two CMF units is set to 8 mil.

Fig. 10 presents the |Scc21| values obtained in simulatedcascades with one to four identical CMF units. Clearly, acascade of two identical CMF units substantially increases thestopband bandwidth and the band attenuation depth comparedto a cascade of one unit. However, a cascade of three or fourCMF units does not substantially increase stopband bandwidthor band attenuation depth. The above analyses show that, for agiven number of CMF units, a cascade of different CMF unitscan yield a wider bandstop CMF with the shortest possibletotal length and a large (20 dB) attenuation depth.

Fig. 9 shows the simulated |Scc21| values obtained for twocascaded CMF units, one with Lr = 278 mil and one withLr = 196.5 mil. Clearly, the −10 dB crossing point ofthe individual values of |Scc21| for the CMF units reveals

Fig. 10. Comparison of |Scc21| values for different numbers of identicalcascaded CMF units.

Fig. 11. Comparison of simulated |Sdd21| (|Sdd12|) and |Scc21| (|Scc12|)in wideband CMF comprising three different cascaded CMF units and inindividual CMF units.

that a wideband CMF with a band attenuation depth below−20 dB can be formed by cascading two different CMF units.Accordingly, a simple design approach that involves a −10 dBcrossing point between the individual |Scc21| values of the twoCMF units is adopted in the following wideband CMF design.

An example of wideband CMF using three different cas-caded CMF units. The lengths of the three CMF units areLr = 170 mil, 240 mil, and 330 mil, are shown in Fig. 11.Fig. 11 presents not only the simulated |Sdd21| (|Sdd12|) and|Scc21| (|Scc12|) of the wideband CMF but also the individual|Scc21| (|Scc12|) for each CMF unit. In the simple designapproach above, the wide stopband CMF has a bandwidth of3.94 GHz to 8.74 GHz at a cutoff frequency of −20 dB anda bandwidth of 3.53 GHz–10.1 GHz at a cutoff frequency of−10 dB. For the differential signals, the insertion loss |Sdd21|is also less than −0.69 dB up to 10 GHz, indicating thatcascading different CMF units efficiently suppresses common-mode noise in a wideband frequency range while maintainingfavorable integrity of the differential signals. Comparing thecorresponding |Scc21| in Figs. 9 and 11, it also shows that theminimum attenuation depth between two lower transmission-


Fig. 12. Comparison of simulated group delays for differential signals withand without the wideband CMF comprising three different CMF units.

zero points of cascaded wideband CMF may be smaller than20 dB. It is because that the combined effects [9] using theproposed simple design guideline is below 4 GHz. However,the CMF with 10 ∼ 15 dB reduction of common-mode noiseis sufficient for solving the signal integrity and EMI issues[5], [6]. Notably, the asymmetry of the CMF unit in Fig. 11diminishes the performance of CMF units cascaded in dif-ferent directions. Restated, the wideband CMF with differentcascaded CMF units is a directional structure. Therefore, tomaintain the good performance of the proposed widebandCMF, all CMF units must cascade in the same direction.

Fig. 12 shows the simulated group delay of the differentialsignals with and without the wideband CMF. The groupdelays of the differential signals at the central frequencywith and without the CMF approximate 0.1099 and 0.108 ns,respectively. The maximum difference in the designedstopband is limited to approximately 1.78%. Therefore, thefavorable signal integrity of the differential signals can bemaintained as they pass through the wideband CMF becauseboth the insertion loss |Sdd21| and group velocity of thedifferential signals are small.

Common-mode noise suppression is also verified in thetime domain. For comparison, the differential traces obtainedusing the reference and the wideband CMF boards are bothsimulated in the time domain. A differential signal with askew of 25 ps is generated to excite common-mode noise.The common-mode noise (Vcommon) is defined as half the sumof the differential voltages at two of the output ports of thedifferential traces. Fig. 13 presents the simulated common-mode noise at the receiving end of differential traces with thereference board and wideband CMF board. The peak-to-peakoutput common-mode voltage obtained using the referenceboard is 0.164 V, but that of the wideband CMF board isonly 0.061 V. An improvement of over 60% is achieved.

The transmission interconnects of many currently usedhigh-speed digital signal standards, e.g., PCI-E, SATA III,USB 3.0, and HDMI, comprise many differential line pairs.Therefore, the development of a wideband CMF that canbe implemented in many differential line pairs would haveimportant practical industrial applications. Thus, embeddingthe wideband CMF in quarter-wavelength openstub resonators

Fig. 13. Comparison of simulated common-mode noise at receiving end fordifferential signals with common-mode noise that pass through the referenceboard and the wideband CMF board.

Fig. 14. Comparisons of simulated |Sdd21| and |Scc21| of two differentialpairs with proposed wideband CMF for various separations, Sp.

underneath and parallel to the differential traces greatly facil-itates its implementation using multiple differential signalingpairs. Fig. 14 compares the simulated |Sdd21| and |Scc21| oftwo differential pairs with the wideband CMFs for variousseparations Sp between the two pairs. A comparison witha single differential pair with a wideband CMF shows thatthe small separation Sp is associated with a small stopbandbandwidth and a small band attenuation depth |Scc21| becauseof the strong mutual coupling effect [9]. Furthermore, a smallseparation Sp is also associated with a small magnitude of|Sdd21|. Fortunately, favorable signal integrity is maintainedbecause the small separation Sp limits degradation of |Sdd21|to 1.2 dB at frequencies up to 12 GHz.

Notably, a common design guideline to avoid crosstalkbetween two differential pairs is to ensure that the separation(Sp) between the two differential pairs of lines is not less


(a)

(b)

Fig. 15. (a) |Sdd31|, |Sdd41| and (b) |Scc31|, |Scc41| in simulations oftwo differential pairs with and without proposed wideband CMF for variousseparations, Sp.

than twice the spacing between the differential traces (S).Therefore, in Fig. 14, the separation (Sp = 13 mil) is justtwice the spacing between differential traces (S = 6.5 mil).A single differential pair with wideband CMF has smallvariations in stopband bandwidth but large variations in bandattenuation depth of |Scc21| when two differential pairs areseparated by Sp = 13 mil. Despite the large variation in theband attenuation depth of |Scc21|, attenuation depth approaches−20 dB. Certainly, a separation of Sp = 19.5 mil yields abetter performance than Sp = 13 mil case. Fig. 14 confirmsthe adequate performance of the wideband CMF for multipledifferential signaling pairs under the common design guide-line.

Fig. 15(a) further shows the HFSS simulation results fordifferential-mode backward crosstalk (Sdd31) and forwardcrosstalk (Sdd41) of two differential pairs with and with-out proposed wideband CMF for various separations (Sp).Notably, the values of differential-mode forward crosstalk fortwo differential pairs with and without wideband CMF do not

Fig. 16. Simulation results for common-mode noise at receiving end (farend) for two differential signals with common-mode noise passing throughtwo differential pairs with and without proposed wideband CMF for variousseparations, Sp.

substantially differ at 0 ∼ 12 GHz. Additionally, the valuesof differential-mode backward crosstalk for two differentialpairs with and without wideband CMF are almost identicalat frequencies below 8.2 GHz. Differential-mode backwardcrosstalk only slightly differs between 8.2 GHz and 12 GHz.Therefore, differential-mode forward and backward crosstalksfor two differential pairs with and without wideband CMFare very similar regardless of the separation (Sp). That is, theresults again confirm that the proposed wideband CMF hasalmost no effect on differential signal performance Addition-ally, the values of differential-mode backward and forwardcrosstalks for two differential pairs under Sp = 2S (13 mil)are both small enough because they are below −30 dB and−20 dB, respectively.

Fig. 15(b) shows the common-mode backward crosstalknoise (Scc31) and forward crosstalk noise (Scc41) obtainedin simulations of two differential pairs with and without theproposed wideband CMF for various separations (Sp). Clearly,the filtering properties of the proposed wideband CMF canaffect forward crosstalk but not backward crosstalk. However,comparing to the without one, the common-mode backwardcrosstalk is increased for two differential pairs with proposedwideband CMF. Additionally, separation is small when |Scc31|is large. The variations in the stopband bandwidth and the bandattenuation depth of |Scc41| are both small when the separationof two differential pairs is small. For Sp = 2S (13 mil), theband attenuation depth of |Scc41| still approaches −20 dB.Therefore, the proposed wideband CMF can still filter theforward crosstalk between the designed bandwidth. However,compared to without one, there are some increased amountsof forward crosstalk below 4 GHz for two differential pairswith proposed wideband CMF.

Fig. 16 compares the simulated common-mode noise atthe receiving end (far end) for two differential signals withcommon-mode noise (generated by time delay) that passthrough two differential pairs with and without the proposedwideband CMF for various separations (Sp). Compared tothe large separation one the common-mode noise for two


Fig. 17. Comparison of simulated |Sdd21| and |Scc21| of the wideband CMFwith four different cascaded CMF units and individual CMF units.

differential pairs without wideband CMF is slightly increasedfor small separation. In other words, most of the common-mode noise at the received end is generated by time delaysin the two differential signals. Hence, common-mode forwardcrosstalk generates a minimal amount of common-mode noise.For two differential pairs with wideband CMF, separationdecreases as common-mode peak-to-peak voltage increases.The simulation results in Fig. 16 also show that this resultsfrom the larger difference in common-mode peak-to-peakvoltage amount between Sp = 6.5 mil and Sp = 13 milcompared to that between Sp = 13 mil and Sp = 19.5 mil.Additionally, the peak-to-peak common-mode voltages areclose for Sp = 13 mil and Sp = 19.5 mil. Hence, accordingto above analysis, the performance of the wideband CMFfor two differential signaling pairs under the common designguideline, Sp = 2S, is acceptable.

Fig. 17 shows the example of a CMF with a wider band-width and four different cascaded CMFs. CMF units withlengths Lr = 120 mil, 170 mil, 240 mil and 330 mil.Fig. 15 presents the simulated |Sdd21| and |Scc21| for thewideband CMF and the individual |Scc21| for each CMF unit.When applying the simple design approach above, the widestopband CMF has a bandwidth of 3.98–13 GHz under acutoff frequency of −20 dB and a bandwidth of 3.54–15 GHzunder a cutoff frequency of −10 dB. For differential signals,insertion loss is less than −0.14 dB at frequencies up to15 GHz. Notably, the second resonant frequency of the long(Lr = 33 mil) quarter-wavelength resonator can help to extendthe bandwidth for the cascading CMF. The simulation resultsconfirm that a wider band CMF implemented by cascadingdifferent CMF units can maintain integrity in differentialsignals.

IV. EXPERIMENTAL VALIDATION

Since the via does not extend to the top layer of the PCB,it must be considered a buried via in the above structure.To ensure low cost and ease of manufacture in the laboratory,test boards are fabricated with plate-through-holes (PTHs).Fig. 18 shows photographs of two boards for testing dif-ferential traces that pass through the wideband CMF and

(a) (b)

Fig. 18. Photographs of two boards used to test differential signals passingthrough (a) wideband CMF board and (b) reference board.

Fig. 19. Comparison of simulated and measured |Sdd21| and |Scc21| of thewideband CMF with three different cascaded CMF units.

the reference boards. The wideband CMF is implemented bycascading three CMF units with different lengths. Additionallyowing to the PTH via, the differential traces must followdetours through the vias of the wideband CMF, as presentedin Fig. 18. Fig. 18(a) shows that the geometrical parametersof a CMF unit of the test board in an FR4 multilayer PCBenvironment are (Wr , h1, 2r , h2, g, Ws , S) = (1.8 mm,0.115 mm, 0.5 mm, 0.495 mm, 0.3 mm, 0.2 mm, 0.5 mm).The substrate material has a loss tangent of 0.02. To minimizemanufacturing costs, a different dielectric constant εr of thesubstrate material is used (inset, Fig. 19). Based on the simpledesign approach above, the lengths of the three CMF units areLr = 6 mm, 7.75 mm, and 10 mm.

Fig. 19 shows the simulated and measured |Sdd21| and|Scc21| for the wideband CMF and the individual |Scc21|for each CMF unit. A frequency-domain network analyzerAgilent/E5071B is used to verify the experimental results.Fig. 19 shows the good agreement between the simulatedand measured results. As evident from the measurements inFig. 19, the bandwidth of the wide stopband CMF is from3.7 GHz to 7.3 GHz under the cutoff frequency by −20 dBand from 3.34 GHz to 7.66 GHz under the cutoff frequency by−10 dB. Moreover, for the differential signals, the insertionloss |Sdd21| is less than −4.48 dB at frequencies up to 10 GHz.

Fig. 20 compares the measured eye diagrams for the differ-ential signals with the reference board and the wideband CMFboard Data ratios of 5 Gb/s are provided by a pattern generator(Anritsu MP1763C) and TDR (Tektronix CSA8000B). Fig. 20shows that, compared to the reference board the widebandCMF board provides sufficiently favorable performance of


(a) (b)

Fig. 20. Comparison of measured eye diagrams of differential signals thatpass through (a) reference board and (b) wideband CMF board.

Fig. 21. Comparison of measured common-mode noise at receiving end fordifferential signals with common-mode noise passing through the referenceboard and the wideband CMF board.

eye diagram. This finding proves that the wideband CMFadequately retains the integrity of the differential signals.

The two differential signals with a skew of 25 ps,derived from a time-domain reflectometer TEK/CSA8000,are regarded as a time-domain common-mode noise source.Common-mode noise is introduced to differential traces ofthe two test boards. Fig. 21 shows the common-mode noisemeasured at the receiving end of the differential traces forthe reference board and wideband CMF board used in thetests. Notably, the peak-to-peak output common-mode voltagefor the reference board is 46.8 mV, but that for the wide-band CMF board is only 23.9 mV. The improvement ratioapproaches 50%.

Differential I/O cables that connect two PCBs must beutilized for serial high-speed digital data transmission, as inSATA III, USB 3.0, and HDMI. Common-mode current indifferential I/O cables can cause significant radiation emissionor EMI in electronic systems. This paper investigates the sup-pression of common-mode current in attached I/O cables [4].The measurement equipment is a vector network analyzerAgilent/E5071B. Fig. 22(a) shows the measurement setup. Theport 1 cable and a 10 cm conductor are connected to the leftand right ends, respectively, of the differential traces for twotest boards. Since the port 1 signal that is excited by VNA isdirectly launched onto the two traces of differential lines, thesignal is regarded as common-mode noise. The common-modenoise passes through the differential traces with and withoutthe CMF to the attached I/O cables with length 10 cm. The

(a)

(b)

Fig. 22. (a) Measurement setup using vector network analyzer and currentprobe. (b) Comparison of measured |S21| values or common-mode currentstrengths between differential traces in the reference board and widebandCMF board.

port 2 cable is connected to a current probe to measure thecommon-mode current strength at the 10 cm conductor.

Fig. 22(b) compares the measured |S21|, which is used todetermine the common-mode current strength between thedifferential signal that passes through the reference board andthat which passes through the wideband CMF board. Fig. 22(b)shows the substantial suppression of common-mode currentby the wideband CMF at 3.5 to 7.8 GHz. The common-modecurrent in the attached I/O cables is reduced by approximately20 dB in the CMF stopband.

V. CONCLUSION

An inexpensive wideband common-mode noise suppressionfilter with a quarter-wavelength resonator was proposed andcomprehensively evaluated. An equivalent transmission linemodel and the electrical field distribution were also used todemonstrate how the presented structure forms a CMF. Otherimportant parameters, including coupling coefficients werepresented to evaluate the properties and performance of CMFunit in this paper. The capacitive coupling coefficient (k ′

c)of the CMF unit correlates with its stopband bandwidth andband attenuation depth. As is well known, a larger couplingcoefficient (k) of differential signals corresponds to bettersignal integrity. However, a larger coupling coefficient (k) of


the differential signals implied a lower performance of theCMF unit.

The feasibility of cascading several quarter-wavelength res-onators to achieve a wide bandwidth over a broad GHz rangewas established. A simple design approach was presentedfor a wideband CMF. The simulation results revealed therejection band ranges of the CMF with quarter-wavelengthresonators of three lengths are from 3.94 GHz to 8.74 GHzand from 3.53 GHz to 10.1 GHz under the cutoff frequencyby −20 dB and −10 dB, respectively. The novel structure ofthe CMF greatly facilitated its implementation in a multilayerstructure and multiple differential signaling pairs for high-speed digital systems. Under the common design guidelinethat the separation between two differential pairs must not beless than double the spacing between the differential traces,the simulation results confirmed the satisfactory performanceof the wideband CMF for multiple differential signalingpairs. Importantly, the analysis of differential insertion lossand group delay confirmed the adequate signal integrity ofthe differential signals when the wideband CMF is used insingle and two differential pairs interconnects. The CMF wasfurther validated by measurements of time/frequency-domaincommon-mode noise, an eye diagram, and measurements ofcommon-mode current in the I/O cable.

ACKNOWLEDGMENT

The authors would like to thank NTUEE SI Laboratoryand Ansoft Taiwan for providing the measurement equipmentand simulation software. They would also like to thank MPICorporation for manufacturing the measured board.

REFERENCES

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Guang-Hwa Shiue (M’07) was born in Tainan,Taiwan. He received the M.S. degree in electricalengineering from the National Taiwan University ofScience and Technology, Taipei, Taiwan, and thePh.D. degree in communication engineering fromNational Taiwan University, Taipei, in 1997 and2006, respectively.

He joined the Faculty with the Department ofElectronic Engineering, Chin Min Institute of Tech-nology, Miaoli, China, in 1999, where he was aLecturer. In 2001, he joined the Faculty with the

Department of Electronic Engineering, Jinwen University of Science andTechnology, Taipei, where he was a Lecturer from 2001 to 2006 and anAssistant Professor from 2006 to 2008. In 2008, he joined the Faculty withthe Department of Electronic Engineering, Chung Yuan Christian University(CYCU), Taipei, where he is currently an Associate Professor. He has been aVice-Director of the Automotive Electronics and Reliability Research Center,CYCU, since 2013. His current research interests include numerical electro-magnetic field, microwave planar circuits, signal/power integrity for high-speed digital systems, electromagnetic interference/compatibility for high-speed/frequency electronic systems, and electrical characterization of system-in-package.

Che-Ming Hsu was born in Taoyuan, Taiwan, in1987. He received the B.S. degree in electricalengineering from Chung Yuan Christian University,Taoyuan, Taiwan, in 2010, where he is currentlypursuing the M.S. degree in communication engi-neering.

His current research interests include signal/powerintegrity and electromagnetics compatibility designfor high-speed digital circuits.


Chi-Lou Yeh was born in Miaoli, Taiwan, in 1989.He received the B.S. degree in electrical engineeringfrom Chung Yuan Christian University, Taoyuan,Taiwan, in 2012, where he is currently pursuing theM.S. degree in communication engineering.

His current research interests include signal/powerintegrity and electromagnetic compatibility designfor high-speed digital circuits.

Cheng-Fu Hsu was born in Taoyuan, Taiwan, in1985. He received the B.S. degree in electricalengineering and the M.S. degree in communicationengineering from Chung Yuan Christian University,Taoyuan, in 2008 and 2010, respectively.

His current research interests include signal/powerintegrity and electromagnetic compatibility designfor high-speed digital circuits.

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