0

cJo

=1. c-)000wz

=.3CD 1-

-I 0 (D(I)F-4-cCO

c13

1t

cc CO

in (D4b

(Y1 -8774

P \oo

p<I>

0

advert issm sot elektor septembet 1978 - UK 3

MK14-the only low-costkeyboard -addressablemicrocomputer!The new Science of CambridgMK14 Microcomputer kitThe MK14 National Semiconductor Scamp basedMicrocomputer Kit gives you the power and performanceof a professional keyboard -addressable unit - forless than half the normal pnce. It has a specificationthat makes it perfect for the engineer who needs tokeep up to date with digital systems Or for usein school science departments. It'sideal for hobbyists and amateur electronicsenthusiasts. too.

But the MK14isn t just a training aidits beendesignedforpractical performanceso you can use it as a working componentof. even the heart of, larger electronicsystems and equipment

MK14 Specification* Hexadecimal keyboard* 8 -digit. 7 -segment LED display* 512 x 8 Prom, containing monitor

program and interface instructions* 256 bytes 'RAM* 4MHz cr.,- :1

** , . wer supply* Space available for extra 256 byte

RAM and 16 port I/O* Edge connector access to al

lines and l(0 portsFree Manual

3corrouter ka inctudec 3

Fo Science of Cambridge Ltd,- Imbndge.

3se send me an MK14 Standard,,tr3,1<tt I enclose cheque/

'or E43 551E39 958p)

operational instructions andexamples for training applications. andnumerous programsincludingmathroutines(square root. etc) digital alarm clock.single-step music box. mastermind andmoon landing games, self -replication,general purpose sequencing. etc

Designed for fast, easy assemblyEach 31 pleCe kit includes everyth ng youneed to make a full-scale workingmicroprocessor. from 14 chips. a 4 -partKeyboard. display interface components,lo PCB, switch and fixings Further softwarepackages. including serial interface to TTYand cassette, are available, and areregularly supplemented

Ti, MK14 ran anyone

T,

Name

Aodress (please print)

The low-cost computing power of themicrcproceSSOr is already being used toreplace other forms of digital. analogue.electro-mechanical, even purelymechanical forms of control systems

The Science of Cambnclge MK14 StandardMicrocomputer Kit allows you to learn moreabout this exciting and rapidly advancingarea of technology It allows you to useyour awn microcomputer in practicalapplications of your own design And itallows you to do it at a fraction of theprice you d have to pay elsewhere

Getting your MK14 Kit is easy. Just fill inthe nnt.ry)n below and PCSt it to us today,

- payable to: of course. it

Science of Cambridge Ltd,6 Kings Parade,Cambridge,Cambs.. C82 15N.Telephone: Cambridge (0223) 311488

Science ofCambridge]

For further information please tick SOC41

UK 4 - elektor september 1978 decoder

elektor 41Volume 4 Number 9

Editor : W. van der HorstDeputy editor : P. HolmesTechnical editors : J. Barendrecht, G.H.K. Dam,

E. Krempelsauer, G.H. NachbarA. Nachtmann, K.S.M. Walraven

Subscriptions : Mrs. A. van MeyelInternational head offices: Elektuur Publishers Ltd.

Bourgognestr. 13aBeek IL), NetherlandsTel. 044024200Telex: 56617 Elekt NL

U.K. editorial offices, administration end advertising:Elektor Publishers Ltd.. Elektor House,10 Longport Street, Canterbury CT1 1PE, Kent. U.K.Tel.: Canterbury 10227154430. Telex 965504.Please make all cheques payable to Elektor Publishers Ltd.at the above address.Bank: 1. Midland Bank Ltd.. Canterbury, A/C no. 11014587

Sorting code 40-16.11, Giro no. 3154254.2. U.S.A. only: Bank of America, c/o World Way

Postal Center, P.O. Box 80689, Los Angeles,CA 90080, A/C no. 12350-04207.

3. Canada only: The Royal Bank of Canada,c/o Lockbox 1969. Postal Station A, Toronto,Ontario, M51111 1W9. A/C no. 160-269-7.

Assistant Manager and Advertising R.G. KnappEditorial : T. Emmons

ELEKTOR IS PUBLISHED MONTHLY on the third Friday of eachmonth.1. U.K. and all countries except the U.S.A. and Canada:

Cover price f 0.50.Number 39/40 (July/August). IS a double issue,'Summer Circuits', price E 1.-.Single copies line. back issues) are available by post from ourCanterbury office, at f 0.60 (surface mail) or £ 0.95 lair marl).Subscriptions for 1978, January to December incl..f 6.75 (surface mail) or f 12.00 lair mail).

2. For the U.S.A. and Canada:Corer price S 1.50.Number 39/40 (July/August(, Is a double issue,'Summer Circuits', price S 3.-.Single copies (incl. back issues) S 1 50 (surface mad) orS 2.25 lair mail(.Subscriptions for 1978, January to December incl.,S 18.- (surface mail) or S 27.- lair mail).All prices include post & packing.

CHANGE OF ADDRESS. Please allow at least six weeks for change ofaddress. Include your old address, enclosing, if possible. an address labelfrom a recent issue.

LETTERS SHOULD BE ADDRESSED TO the department concerned.TO - Technical Queries, ADV - Advertisements. SUB Subscriptions.ADM - Administration; ED Editorial (articles submitted forpublication etc.); EPS - Elektor printed circuit board service.For technical queries. please enclose a stamped, addressed envelope ora self addressed envelope plus an IRC.

THE CIRCUITS PUBLISHED ARE FOR domestic use only. The sub-mission of &Signs or articles to Elektor implies permission to thepublishers to alter and translate the text and design, and to use thecontents in other Elektor publications and activities. The publisherscannot guarantee to return any mineral submitted to them. Alldrawings, photographs, printed circuit boards and articles published inElektor are copyright and may not be reproduced or imitated in wholeor pert without prior written permission of the publishers.

PATENT PROTECTION MAY EXIST in respect of circuits, devices.components etc. described in this magazine.The publishers do not accept responsibility for failing to identify suchpatent or other protection.

National ADVERTISING RATES for the English -language edition ofElektor and/or international advertising rates for advertising at the sametime in the English, Dutch and German issues are available on request.DISTRIBUTION in U.K.: Spotlight Magazine Distributors Ltd.,Spotlight House 1, Bentwell Road, Holloway, London N7 7AX.DISTRIBUTION in CANADA: Gordon and Gotch ICan.1 Ltd.,55 York Street, Toronto. Ontario, M5J 1S4.Copyright Elektor publishers Ltd - Canterbury.Printed in the Netherlands

decoderWhat is a TUN?What is 10 n7What is the EPS service?What is the TO service?What ell a missing link?

Semiconductor typesVery often, a large number of quivalent semiconductors existwith different type numbers. Forthis raison, 'abbreviated' typenumbers are used ,n Elektorwherever possible '741' stand for 14A741,

LM741, MC641. MIC741,RM741. SN72741. etc.

'TUP' or 'TUN' (Transistor.Universal, PNP or NPN respect-ively) stand for any low fre-quency silicon transistor thatmeets the following soma i-cat ions:

U- CEO. max1C, maxMe, mmPlot, maxfT min

20V -1100 mA100100 mW100 MHz

Some 'TUN's are, BC107. BC108and BC109 families, 7N3856A,2N3859, 2N3860. 2N3904.2N3947, 2N4174 Some 'TUP'sare' BC177 and fiC178 families.BC179 family with the possiblemotion of BC159 and 8C179,2N2412. 2N3251, 2N3906,2N4126. 2N4291

'DVS' or 'DUG' (Diode univoc-al, Silicon or Germaniumrespectively) stands for anydiode that meets the followingspecifications

:DUS 'DUG*UR. max 25V 20VIF, max 100mA 35mAIR, max I she1 100 ssA'Ptot. max 250mW 750rTIWI I'co max ,5pF 10pF

Some 'OUS's are BA127. BA2178A218, 8A221. BA222. BA317.8A318, BAX13.BAY61, 1N914,1N41410Some VUG's are 0A85. 0A91,0A95. AA116

'BC107E1'. 'BC23713', 'BC547B'all refer to the same 'family' ofalmost identical better -qualitysilicon transistors In general,any other member of the samefamily can be used instead.

BC107 1-8. 91 families:BC107 (S, .9). BC147 (8. -9).BC207 18, -91, BC237 1.8. -9/,BC317 1-8, 4/, BC347 18, .9).8C547 Ia. .9), BC 171 I.2, -3).BC182 1-3. 4), BC382 1.3, 4),BC437 I-8. 8C414

BC177 1.8, -91 families.BC177 I8. 8). BC157 -9),BC204 1-5. 61. BC307 (8, -91.BC320 1.1, -2), BC350 1-1.BC557 -9), BC251 1.2,BC212 1.3. 41. BC512 1.3, 41,BC261 BC416

Resistor and capacitor tittlesWhen giving component values.decimal points and large numbers

of zeros are avoided whereverpossible The decimal point isusually replaced by one of thefollowing abbreviations

Pn Ilna"Drso.!) : 2-.2

um :,n7,o.,.): ;QV')k lkilo-1 101M (maga-) 10'G (gigs-) - 10.A few examPleSResistance value 2k7 2700 11.Resistance value 470. 470 11Capacitance value 4p7. 4 7 pF, or0.000 000 000 004 7 FCapecotance value 10n this is theinternational way of writing10,000 pF or 01 0F, since 1 n is10-6 farads or 1000 pFResistors are ',. Watt 5% carbontypes, unless otherwise specifiedThe DC working voltage ofcapacitors (other than electro-lyt ics1 is normally assumed to beat least 60 V As d mule of thumb,a safe value is usually aporoximately twice the DC supplyvoltage

Test voltagesThe DC test voltages shown aremeasured with a 20 k11 V instru-ment. unless otherwise specified.

U, not VThe international letter symbol'U' for voltage is often usedinstead of the ambiguous 'V''V' is normally reserved for Volts'For instance. Ub - 10 V,not Vb 10 V.

Mains voltagesNo mains (power lino, voltagesare listed in Elektor circuits It isassumed that our feeders knowwhat voltage is standard in theirpart of the worldsReaders in countries that use60 Hz should note that Elektorcircuits are designed for 50 Hzoperation. This will not normallybe a problem; however, in caseswhere the mains frequency is usedfor synchronisation some model 1rcation may be required.

Technical services to readers EPS service. Many Elektorarticles include a layout for aprinted circuit board. Some - butnot all - of these boards are avail-able reedy -etched and predrilled.The 'EPS print service list' in thecurrent issue always gives a Commplate list of available boards Technical queries. Members ofthe technical staff are available toanswer technical queries Irelatingto articles published in Elektor)by telephone on Mondays from14.00 to 16.30 Letters withtechnical queries should beaddressed to. Dept. TO. Pleaseenclose a stamped, self addressedenvelope, feeders outside U Kplena enclose an I RC instead ofstamps. Mining link. Any importantmodifications to, additions to.improvements on or correctionsin Elektor circuits are generallylisted under the heeding 'MissingLink' at the earliest opportunity.

contents @folder september 1978 - UK 5

IV

The puffometer, whilstmaking no claims toproviding accuratemeasurements, howeverdoes represent arelatively simple (andamusing) means ofmeasuring just how long-winded some peopleare. Its great for parties!

p. 9-02

Electronic temperature -controlled soldering ironsoffer many advantagesover the continuous heattypes. The Elektor circuitis a thermostaticcontrol unit, which isboth easy to build anduses standard parts.

p. 9-24

The car start booster offers thathelping hand that may make

the difference on thosecold winter mornings.

p. 9-30

elektor'la

;so. ..i-

An artist's impression ofthe Elektor piano, whichmay not replace theSteinway but does havemany applications wherecost and portability areimportant factors.

contentsselektor UK 14

puffometer 9-02

osci I lographics 9-06This little circuit will make your scoperelatives and friends will appreciate

something your

one chip does not make a piano 9.08

master tone generator 9-09Although primarily intended for use in the Elektor Pianothe design of this master tone generator lor top octavegenerator) is sufficiently universal to permit its use in a widevariety of electronic keyboard instruments.

piano 9-12

temperature -controlled soldering iron 9-24

car start booster 9-30

applikator 9-32

24 dB VCF - C. Chapman 9.34In response to requests from readers who have built theFormant synthesiser the following article presents a designfor a voltage -controlled filter whose slope is considerablysteeper than that of the original VCF, in fact 24 dB/octave asopposed to 12 da'octave. The filter offers a choice of high-pass or lowpass modes and slopes of 6. 12. 18 or 24 d13/octave.

buffered/unbuffered CMOS 9-43Many CMOS digital ICs are now available in two versionsbuffered and unbuffered. The lack of information availableto the amateur on the differences between them has givenrise to some confusion as to the compatibility of the twotypes. This article examines both types of CMOS and com-pares them with TTL

market 9-50

advertisers index UK -24

UK 14 - elektor sapternbar 1978 sokokior

Blue FoxBlue Fox is an integral part of theweapons system of the Royal Navy'sSea Harrier vertical take off fighter/strike/reconnaisance aircraft.A distinctive feature of the radar is itsflat aperture aerial designed to increasedetection range and produce minimumsidelobes. Another distinctive feature isthat the scanner is stabilised in bothpitch an roll. Roll stabilisation assists inachieving the maximum accuracy inair -to -surface attacks.Blue Fox is a lightweight radar weighingless than 190 I bs (86 kgs). It isdesigned to fulfill a dual role: that ofairborne search and interception; andair -to -surface search -and -strike.Having successfully completed its initialground trials, Blue Fox is currentlybeing prepared for flight trials later thisyear ( 1978).Blue Fox is a monopulse radar operatingin the X -band. Frequency agilitymaximises its ability to detect smalltargets in clutter conditions and alsoimprov-s its immunity to electroniccountermeasures.For air-to-air interceptions Blue Foxprovides directional and range data tothe Sea Harrier's weapon -aiming

computer for lead -pursuit or chaseattacks. In the strike role it will be usedfor aiming air -to -surface weapons, theradar derived information providingweapon -release data.Blue Fox employs monopulsetechniques in both the horizontal andvertical planes. When searching fordistant targets a PPI scan can be usedand changed to a sector B scan ifrequired. The pilot may choose either asingle -bar or multiple -bar search scanpattern depending on the amount of skyto be covered.The main radar unit comprises a numberof line replaceable units (LRUs).Built-in test facilities are provided toenable faulty LRUs to he identified andreplaced at the first -line servicing level.Modular construction and detail designsimplify the replacement of anyunserviceable components at second andthird -line maintenance levels. Predictedmeantime between mission failures is inexcess of 120 hours.Because Blue Fox is designed for arelatively small aircraft, is of modularconstruction, and light in weight, it isexpected to prove suitable forinstallation in many other types ofaircraft and, as such, possessesconsiderable development potential.

Electronic Systems Department,Ferranti Limited. Ferry Road,Edinburgh, ENS 2XS, England

(347 Sl

Unmanned submersibleA completely new type of micro-computer -controlled unmannedinspection system - built to operate inthe poor visibility and hostile operatingconditions of the North Sea - has been

launched by Richmond -based MarineUnit Technology Limited.The new vehicle - which is supportedby the Department of Energy throughthe offshore Energy Technology Board- is lighter, smaller, more versatile, andfar more controllable than otherunderwater inspection systems currentlyavailable.The new system is code -namedSMARTIE (Submarine AutomaticRemote Television InspectionEquipment). It is elliptical in cross-section and is basically a highly mobileunderwater vehicle equipped with abattery of underwater televisioncameras. These will consist of at leastone low -light silicon intensified target(SIT) camera and a high resolutionvidicon camera. The vehicle is driven byan electrically -powered submersiblepump and is therefore propellerless.

On -Board Computer

The addition of on -board computerfacilities has enabled Marine Unit toprovide the offshore industry with asubmersible which is much morepowerful and versatile than has beenavailable to the industry in the past. Themicrocomputer was designed anddeveloped by Marine Unit Technology'sresearch and development team headedby Dr. Brian Ray. This is the firstoccasion on which a microcomputer hasbeen installed on an underwater vehicleof this type.Apart from the relatively straight-forward procedures of interpretingmanual input control signals from theoperator's console, and controllingvehicle speed and direction, thecomputer is also capable of makingSMARTIE a good deal easier to operate.For low visibility work, the computercan accept input from the submersible'smagnetic compass and gyro, and projectan artificial navigation 'target' which theoperator can follow on his video screeneven though the craft may be passingthrough an area of zero visibility.Similarly, if the submersible is operatingin fast currents, the operator is able tomaintain a fixed position in the waterby simply pressing a 'hold' button: thecomputer will automaticallycompensate for the effects of thecurrent, keeping the vehicle in therequired position without operatorintervention. The 'hold' facility is alsouseful for keeping a steady course atspeed.

Slim Umbilical CableThe vehicle will be supplied with powerand control signals by a single umbilicalcable under half a centimetre indiameter. The video signal will becontinuously transmitted hack to thesurface via the same cable.Most unmanned submersibles aresupplied by very bulky multi -core cableshearing separate conductors for the

w elektor septembor 1978 - 9-01

y

Pr

IC

's

IS

e.

's:tto

ig

tl

.es

ailektee

IJ

control circuits, power circuits andvideo controls and signals. Thisconfiguration can affect theperformance of the vehicle adversely ina number of different ways.Firstly, there is a very real problem ofphysical drag exercised by the cable onthe vehicle; secondly, there is theproblem of interaction between thevideo, control and power circuits, aproblem which can cause interference orother deterioration in video picturequality.Careful design of the electronics hasensured that these problems do notoccur, and at the same time has made itpossible to keep the cable dimensions toa minimum.

Projected DeyelopmintsMarine Unit will eventually be offering anumber of different services to theoffshore industry, and the next phase isalready under development. One facilitywill be to enable the vehicle to lock -on'and follow closely a pipeline or othersubmerged structure at a fixed distancefrom it. It is also hoped to fit sonarequipment at some later date.

Offshore Survey ServiceSMARTIE will not be sold to theoffshore industry for the time being.Instead, MUT's Associate CompanyMarine Unit Limited will offer acomplete inspection service to theoffshore industry. It is expected thatthe main applications will be insurveying pipelines, underwaterconstruction work, and other workwhere the cost of putting a man on thesea-bed is becoming very expensive.SMARTIE will be made available in oneof two operating modes. Firstly, it willbe available anywhere in the world in anemergency model air transport kit,consisting of the SMARTIE submersibleand electrical generator, operator'sconsole and cable. Three trainedSMARTIE operaturs will accompanythe submersible. SMARTIE will also be

available for long-term contracts with acustom-built winch and a robustlauncher which will be designed towithstand very rough surface conditions;in this mode, SMARTIE will belaunched at depth and a prefabricatedbuilding will house all the controls.Marine Unit will have available twoSMARTIE units by the end of July,with others following later in the year.All the production models will bemanufactured at the Group's newPlymouth factory.SMARTIE should provide the offshoreoperator with a very realistic low-costalternative to the diver used as anunderwater observer, even in shallowwater. Over the years it has been proventhat an operator on the surface actuallysees more on his screen than the diveron the sea-bed. Now SMARITE givesthe Television camera the mobility, theprecision and the manoeuvrability ofthe diver. In addition, SMARTIE canoperate in more hazardous conditionsthan the diver.

Marine Unit Technology Ltd3 Friars Lane, RichmondSurrey TN9 INL, England

1348 S1

9-02 - elector 'optimiser 1978 puffometer

puffometer

Obviously, the length of time one cancontinue to whistle a particular notedepends on how loud one tries towhistle and on how much air from thelungs is available to sustain the note.These factors determine the basicprinciple of the puffometer, whichprovides a comparative indication ofdifferent people's lung capacity bymeasuring how long they can sustaina note above a particular level. Amultimeter is used to provide an opticalindication of the duration of the signal.It goes without saying that thepuffometer is not intended to he a

serious scientific instrument, but ratheris suited for 'party -piece' applications(where the reader may well find that hisperformance on the puffometer isdetermined more by his ability to keepa straight 'pucker' than the capacityof his lungs!).

Block diagramThe block diagram of the puffometeris shown in figure I . As can be seen, thecircuit is comprised of a number ofdifferent stages, each of which will beexamined separately.The whistled note is picked up by amicrophone, which of course, convertsit into an electrical signal. This is thenfed to the lowpass filter, blockwhich removes the high frequencyinterference components from the signalwhich could be generated by nearbyelectrical appliances.Block 2 contains an amplifier with again of between 50 and 500. Once thesignal has been amplified it is fed to ahandpass filter with a centre frequencyof 1900 Hz. This means that signalswith a frequency of 1900 Hz are passedunaffected, whilst frequencies aboveand below this figure are attenuated.The filter output signal is then rectifiedby the circuit in block 4. The waveformof the whistled note is basically a

sinewave, i.e. it contains positive andnegative half cycles. During the negativecycles when the diode is conducting, thecapacitor, C6, rapidly discharges (underquiescent conditions the capacitor isheld charged via R9).The output of the rectifier is fed to acomparator, which, as its name suggests,

Spirometers, which areinstruments designed to measurea person's lung capacity, are bothfairly complicated and expensivedevices, and would be difficult toconstruct at home. The'puffometer' described in thisarticle, whilst making no claimsto providing accurate volumemeasurements, nonethelessrepresents a relatively simple(and amusing) means ofcomparing how much air differentpeople can expel in a singlebreath by measuring how longthey can hold a whistled note.

Figure 1. The block diagram of the puff o -meter The numbering of the blocks isrepeated in the circuit diagram of figure 2.

Figure 2. The complete circuit diagram ofthe puffometer. The input signal is pickedup via a crystal microphone and a multimeteris used for the readout.

Figure 3. The power supply for the puffometer. The power supply circuit is also usedto derive the 5 V reference voltage. Withthe exception of the transformer, bridgerectifier and smooting capacitor, the powersupply is mounted on the p.c.b.

compares the capacitor voltage with apreset reference voltage, UR.As soon as the capacitor voltage fallsbelow the reference voltage, the outputof the comparator swings low. Thiscauses the LED to light up. therebyindicating that the amplitude of thewhistled note is sufficiently large andof the correct tone to be processed bythe puffometer. As long as the compara-tor output is low, the capacitor inblock 6 remains discharged; the startbutton is pressed, the output voltage ofthe integrator starts to rise. This voltageis then displayed on a meter.When the input signal to thephone stops, the capacitor in block 4once more charges up, taking the outputof the comparator high, extinguishingthe LED and causing the capacitor inblock 6 to also charge up. The timetaken for this capacitor to charge fullyis determined by the value of thecapacitor itself and of the series resistor;block 6 is therefore basically a delaynetwork. When the capacitor voltagereaches a certain value, the outputvoltage of the integrator ceases to riseand is held at whatever level it hasreached at that moment. This meansthat the meter reading will be helduntil it is reset.When a second person wishes to testhis 'puff power', the pushbutton isdepressed, resetting the meter.

Circuit diagramThe complete circuit diagram of thepuffometer is shown in figure 2. Themicrophone signal is fed via the DCdecoupling capacitor, CI. to the lowpassfilter formed by R2 and C2. Signalswith a rise time shorter than 470 ns, i.e.frequencies greater than 340 kHz. aresuppressed.Al is the amplifier of block 2. The gain(A) of the amplifiers equals

+ PiA =R

+3

which means that PI can be used tovary the sensitivity of the puffometcr.The output voltage of the amplifier isdivided across R5 and Ito, and fed viaC5 to the bandpass filter circuit roundA2. The negative feedback loop (i.e. a

putfonieter elektor septembor 1978 - 9-03

2

it 18

Al A4 ICI - 324N1 ... N4 IC2 4031D1.03 D5 - 1N4148D2 LED

=10A 101c- 1900 Hz

See text9941

SY

UR

9-04 - elektor september 1978 puff °meter

Figure 4. Track pattern and componentlayout of the p.c.b. for the putfometer(EFS 9661).

Figure 5. The interior of a finished modelof the puflometer. Care should be taken toensure that the 240 V connections are safelyinsulated.

Figure 6. An inexpensive alternative to a

multimeter can be formed by using a 100 as -meter and a current limiting resistor. Full-scale deflection should be obtained at anoutput voltage of roughly 7 V. If a 1 mAInstrument is used, the value of the currentlimiting resistor should be 6k8.

Figure 7. A modern and attractive case forthe puttometer can be constructed usingperspex and a little skill.

portion of the output signal is fed inantiphase back to the input) via C4 andR7 ensures that, just as the gain of Alcould be varied by altering the value ofthe feedback components. so the gainof A2 varies with the frequency of theinput signal. The values of the variouscomponents were chosen so that atapprox. 1900 Hz A2 would have maxi-mum gain.The output of the filter is fed to thepeak rectifier circuit of block 4. Underquiescent conditions C6 is kept chargedvia R9. When the microphone picks upan input signal, C6 will be dischargedvia DI whenever the output of the1900 Hz filter goes low (i.e. negative).Since R8 is much smaller than R9, thecapacitor discharges much more rapidlythan it charges, so the voltage on C6 islower than that of the reference voltage.The comparator, A3, has no feedback,which means that it has a very high gain.The difference between the referencevoltage on the inverting input and thevoltage across C6, on the non -invertinginput, is amplified to such an extentthat it is either 'high' or 'low'. When itis low current flows through RIO andthe LED, D2, which causes it to lightup.

If the voltage across C6 exceeds thereference voltage of the comparator,however, the output of the comparatorwill swing up to plus supply. The resultis that there is no longer a voltage dropacross RIO and D2. hence the LEDgoes out. This indicates that either thewhistler has stopped whistling or iswhistling at the wrong frequency.The circuit round NI, N2 and thepushbutton SI forms a flip-flop. Thevoltage level at pin 4 is determined bythe voltage levels at pins I and 6. Underquiescent conditions, the output of A3is high, so that capacitor C7 is chargedup via RI I. Pin I of N2 is thereforehigh, whilst pin 6 of NI is low (SI isnormally open), with the result that theoutput of NI (pin 4) is also high.As soon as the microphone picks upan input signal, the voltage on C7 falls;if the start button is then pressed,pin 6 of NI is taken high, with theresult that the output of NI goes low, asdoes pin 2 of N2. This means that, aslong as an input signal is present and C7is discharged, the output of N2 is heldhigh, so that even after the pushbuttonhas been released, the output of NIremains low.N3 monitors the output states of the

flip-flop and of the comparator, A3.Initially these are both high, so thatthe output of N3 is low, with theresult that the output of inverter, N4,is high. Although a current flowsthrough P2, D4 is reverse biased andensures that no current flows to A4.When an input signal is present, theoutputs of A3 and of the flip-flop arelow, so that the output of N3 is high,and that of N4 is low.With no input signal the voltage at bothinputs of A4 is virtually the same(UR). The input bias current setting ofthe op -amp is such that a small currentflows from the inverting input via R15,P3 and R16 to earth. This current isjust sufficient to keep C8 charged.When the output of N4 goes low, i.e.falls below UR, current attempts toflow from the inverting input of A4through R14, D4 etc. Due to the highinput resistance of the op -amp, how-ever, this current can only be suppliedround the negative feedback loop. Theoutput voltage of the op -amp thereforeswings positive to hold the voltage atboth inputs the same. C8 first dischargesthen charges up with reverse polarity.The time taken for the output of theop -amp to swing high is determined by

pad/omens etektor september 1978 - 9-05

parts list to figures 2 and 3

Resistors:

R1,R4,R9 = 47 kR2,R3,R8,R17,R18 1 kR5 4k7R6 - 2k2R7 100 kRIO = 390 SIR11= 1 M 1100 kl*R12,R13 - 10kR14,R15- 10MR16 - 22 kP1 - I M preset potentiometerP2 50 k preset potentiometerP3 2k5 preset potentiometer

Capacitors:CI,C10,C11 = 100nC2 - 470 pC3 1µ5/10 VC4,C5' 10 nC6,C9 - 10µl10 vCl-470 n110 nlC8 - 1 s polycarbonate or

Polyester

Semiconductors:Al ... A4 - IC1-- 324NI ... N4 1C2 - 4001IC3 78L10D1,133 .. D5 1N414802 LED

Miscellaneous:

transformer 12 V, 500 mA secondarybridge rectifier 40 V/400 mA or

4 x 1N4001electrolytic capacitor 220 u/25 VSi pushbutton, push-tomakeFl - fuse 100 mAdouble -pole on/off switchmicrophoneif used: 100 µA -meter68 Is' resistor

sea test

the time constant C8/R14. The rise inthe output voltage of A4 can bemeasured on a multimeter. As soon asthe input signal ceases, the output ofN4 is once more taken high. D4 isreverse biased, and thanks to the smallbias current from the inverting inputof A4, the voltage on C8 is held at thatinstantaneous value (provided P3 is setcorrectly). The meter reading can thenbe checked at leisure. Pressing the startbutton has the effect of returning theinverting input of A4 to UR, so that theoutput falls back to zero, therebyrestoring the original polarity of CH.The reference voltage UR is derivedfrom the supply, the circuit diagram ofwhich is shown in figure 3. As can beseen, a 10 V IC regulator is used. RI7and RI8 divide down the regulated10 V supply to derive a 5 V referencevoltage

ConstructionThe track pattern and component!ayout of the p.c.b. for the puffometerare shown in figure 4. The componentnumbering in this figure corresponds tothe numbering in figures 2 and 3. Thetransformer, four rectifier diodes (or a

single bridge rectifier) and the 220 NFelectrolytic are not mounted on theboard. The photograph shown infigure 5 indicates how these can bemounted safely and compactly besidethe p.c.b.The ICs are best mounted using sockets;this reduces the risk of overheating theIC pins during soldering.The centre frequency of the bandpassfilter, and hence the desired pitch ofthe required whistled note, can bevaried by altering the values of C4and C5. A smaller value will increasethe centre frequency whilst a largervalue will decrease it.There is a certain delay between themoment a person stops whistling andthe needle on the meter settlingat a final reading. This delay is deter-mined by the time taken for C7 tocharge up, It is therefore possible toincrease the duration of this intervalby increasing the value of C7. It is evenpossible for the skilled owner of thepuffometer to cheat a little by selectinga value for C7 which allows him to takein a quick breath in the middle of hisattempt!If the above mentioned cheat functionisn't desired the unit's operation can be

changed by substituting a 10 n capacitorfor R11 and by replacing C7 with a100 k resistor. If this change is incor-porated then the function of SI is

changed. By pressing SI once the meteris reset to zero. Once the whistlingstarts and the LED lights the metershould start to climb, but once thewhistling stops, however briefly, themeter reading is frozen, thus thwartingany cheaters.Any normal type of microphone, suchas those used with cheap cassetterecorders will prove suitable.A multimeter set to the 10 Volt DCrange can be used as the readoutindicator for the puffometer. Or, ifavailable, the LED voltmeter describedin Elektor 12, April 1976 could also beused, or a panel meter with suitablerange resistors like that shown infigure 6 will also work nicely.

CalibrationCalibrating the puffometer is a relativelysimple matter, since there are only threepotentiometers which require adjust-ment.To start, PI and P2 should be turnedfully anticlockwise. LED, D2, shouldnow be lit. Now, whistle very softlyinto the microphone, the meter shouldrise fairly fast, when a reading of about5 Volts is reached, stop whistling. Themeter reading should be stable, i.e. notfalling or rising. If the reading is notstable adjust P3 one way or the otherso that the reading is frozen, notdrifting. Now, P1 and P2 can beadjusted. P1 adjusts how sensitive themicrophone is, and P2 controls the rateor speed the meter climbs to full scale.P2 should therefore be adjusted suchthat the needle is deflected slowlyenough that the most long windedperson just runs out of breath beforethe meter reaches maximum reading.PI is adjusted so the unit isn't toosensitive, otherwise it might respondto background noise.

OperationIf the unit is wired as shown in thecircuit of figure 2 then SI functions asfollows. SI should be pressed and helduntil the whistling starts and the LEDlights, then it should be released.If the unit is modified (R11 and C7changed) then SI should be pressed toreset the unit and then released. Oncethe whistling starts and the LED lights,the meter will start automatically. N

9-06 - elektor september 1978 oscillographscs

oscillographicsFascinating geometrical patterns on an oscilloscope screen

The accompanying illustrations givesome idea of the variety of differentpatterns that can be generated by the'spirator' (spirographics generator). Ascan be seen, they are similar to thepatterns which can be produced byhand using the popular 'SpirographTM'outfit, and also to the type of figuresoften produced by computer graphics.The patterns are derived from certainbasic geometrical functions, and areknown as 'Lissajous' figures. They areto he found in nature, for example inthe path described by an object fixed tothe end of a rope which is oscillating.In geometrical terms a Lissajous figure isobtained when a point describes asinewave on both the X- and Y-axes.The circuit of the spirator produces twosinewave voltages, the frequency ofeach being independently variable. Bothsinewaves are damped. i.e. after thewaveform has been started, the functionwill decay exponentially to zero.

Block diagramThe function of the spirator can beexplained simply be referring to theblock diagram of figure I.The circuit is built round two damped-sinewave oscillators, one of which isresponsible for the vertical deflection1Y -signal) of the spot on the screen, theother for the horizontal deflection(X -signal). Both the frequency and thedegree of damping of the two oscillatorsare independently variable by means ofpotentiometers. It is also possible tomodulate either oscillator frequency byan external signal, so that the patternsare continuously changing.Since the oscillators are not free -runningbut have to be started, there is anastable multivibrator which ensures thatboth oscillators are triggered simul-taneously. The trigger frequency is

60 Hz. Whilst the oscillators are being

An oscilloscope can be used notonly as a test instrument; with theaid of the following circuit it canbe made to generate a multitudeof fascinating and attractivegeometrical patterns.

M. Zirpel

started the spot on the screen is blankedby means of the brightness signal Z,thereby suppressing unwanted linesproduced by the normal scan of thescope.

Circuit diagramThe complete circuit diagram of thespirator is shown in figure 2.The astable multivibrator which pro-vides the trigger pulses for the twodamped-sinewave oscillators is formedby the simple circuit round opamp ICI.As was already mentioned, the fre-quency of the multivibrator signal is60 Hz, sufficiently high to ensure aflicker -free picture on the scope screen.The period for which the output of ICIis high is much longer than the penodfor which it is low. During the latterportion of the signal the trace is blankedvia the brightness input. The nextpositive -going edge at the output ofICI triggers the sinewave oscillators.The two oscillators are identical andconsist of three 741 opamps. IC2, IC3and IC4 comprise the oscillator for thehorizontal (X-lsignal. whilst IC5, IC6and IC7 form the vertical (Y -)signaloscillator. To see how they work let ustake the example of the X -oscillator.Opamps IC2 and IC3 are both connec-ted as integrator and thus, at a certainfrequency. produce a phase -shift of180° in sinewave signals. A furtherphase -shift of 180° is introduced byIC4. which functions as an inverter.The three opamps together thereforehave the total phase -shift of 360°required for oscillation. The total gainof the 3 cascaded opamps can be variedby means of PI and is always less thanunity. Thus, once started, the oscillatorgenerates a damped sinewave. Thedegree of damping can be varied bymeans of PI (P3 in the case of the

elY -signal), and the frequency of the

ostillographies elektor utptember 1978 - 9-07

Figure 1. The spirographics generator consistsof two damped-sinewave oscillators which aretriggered by an astable multivibrator. Boththe frequency and damping of the sinewavescan be varied independently. The OW1114101output signals are used to control the X -and Ydeflection of an oscilloscope trace;the result is a fascinating display of Lissajousfigures.

Figure 2. The complete circuit diagram ofthe spirator. Two extra modulation inputsallow the patterns to be continuously varied.Various different types of oscillator (sine -

wave or triangle-) can be used to provide themodulation inputs. Waveforms with steepedges lsawtooth, squarewave etc.) result inan abrupt transition from one pattern toanother)

240164066

ICI IC7 - 741S/.. S4 - ICI - 4016. 4016

0)0 ,

9-08 - elektor september 1978

Figure 3. Those photograph' give some ideaof the type of patterns which can be obtainedfrom the spirator.

oscillator can be adjusted by meansof P2.In order to obtain a stable picture onthe screen, both oscillators must, ofcourse, be triggered simultaneously.This is ensured by means of electronicswitches SI ... S4. At each positive -going edge of the output signal ofICI they cause the capacitors C2 . C5in the feedback loops of the opamps tobe discharged. This sets the initialconditions of each opamp, i.e. that theoutput always starts from the samevoltage. Thus each successive patternwritten on the screen appears at exactlythe same point thereby producing acompletely stable image. Without thisprecaution, integrator drift would causethe same pattern to appear at slightlydifferent positions on the screen - ahighly undesirable effect.The circuit also offers the particularly

oscillographics

aesthetic possibility of displaying movingpatterns by varying the frequencyand/or damping of one or both oscil-lators. To this end the circuit providesextra control inputs (M1 and M2 ) forlow frequency modulation signals(from, e.g. a sinewave generator).Naturally enough, various types ofsignal (squarewave, triangle etc.) ofvarying amplitude and frequency can beused. The type of waveform will deter-mine the design of the resulting pattern,the frequency will determine the speedat which it changes, whilst the ampli-tude affects the extent to which thepattern varies. The only constraint uponthe modulation signal is that it shouldnot contain a DC component (i.e. itshould be symmetrical and AC coupled).otherwise there is the possibility thatpart of the pattern will be off thescreen. The maximum amplitude of themodulation signal is 15 Vpg. If desiredthe value of R I4 and -R21 can bealtered to suit input levels greater thanthis.Depending upon the type of oscillo-scope used, it may be necessary toinvert the Z -signal, in which case thesignal can be taken from the collectorof T I .

If the picture is not completely flicker -

free, then the value of CI should bereduced accordingly.

chipone does not make a piano

one chipdoes notmakea piano

The General Instruments 'piano -IC' (theAY -I-1320) took some taming . .

Initially, several manufacturers of elec-tronic musical instruments wereinterested in this chip. In particular, theingenious keying system is noteworthy:the loudness of a note depends on theforce with which the key is struck, as ina 'real' piano.It was a great disappointment to discoverthat the designers were apparentlysatisfied with a signal-to-noise ratio thatis more suited to digital circuits than tomusic. To give an idea, the permissibleS/N ratios for a few technologies are asfollows:

CMOS: 10 dB (30% logic level tol-erance)

TTL: 10 dB 110% logic level tol-erance)

stereo: 26 dB (5% crosstalk be-tween channels)

'DIN HiFi': 50 dB (0.3`7'e, S/N ratio)For electronic organs and the like, theunwanted signals (i.e. the output fromall keys except the ones actually de-pressed) should be at least 50 dB down.To our great surprise, a prototype pianobuilt according to the GI applicationnote proved to have a SIN ratio closerto 10 dB! On contacting several elec-tronic organ manufacturers, we dis-covered that they had encountered thesame problem and - as far as ourinformation goes at present - they havenow all given it up as a bad job.However, our designers suffer from astubborn streak. It took some doing,but they finally came up with a satisfac-tory circuit. Feedthrough of unwantedsignals is reduced to the point where itis no longer audible (S/N better than50 dB) by means of an additional elec-tronic switch for each key. This doesincrease the price by about L3 for eachoctave, which is a pity. Perhaps GIcould consider designing an unprovedversion of the original chip, or possiblya second add-on chip for the gating?Anyway, we've finally found a circuitthat works - and that's quite a relief. N

master tone plinorator eleklor september 1978 - 9-09

r

e

0

n

r

e

e

a

3.

it

n

!s

h;Id

y

it

mastertone

generatorKeyboard instruments use the equallytempered scale, in which each octave isdivided into twelve semitones. Any twoadjacent semitones differ in frequencyby a ratio 1:1, or 1.0594631. Notes12 semitones or one octave apart obvi-ously differ in frequency by a factor(kyn12, i.e. 2. Since it is not really prac-tical to use a separate oscillator for eachnote of a keyboard instrument, it hasbeen common practice for a number ofyears to use separate oscillators togenerate the twelve notes of the topoctave, and then by the process of fre-quency division, by powers of two, thelower octaves can be derived.More recently it has become possible toreplace the twelve top octave oscillatorsby a single master oscillator whichgenerates the twelve top notes from asingle (crystal -controlled) clock. Theadvantages of this approach are obvious.Firstly, since the top octave notes arelocked to the clock generator and thelower octaves are locked to the topoctave, a single adjustment serves totune the whole instrument. This greatlysimplifies the setting up procedure andallows the instrument to be tuned easilyto other instruments. Secondly, the useof a crystal clock allows excellentstability of tuning.The master oscillator is necessarily morecomplex than the octave dividers whichfollow it. It is not possible to generate,say, topC and then divide by 1.0594631to obtain B, since digital frequencydividers can divide only by integralnumbers. The solution is to derive bothC and B (and all the other notes) bydividing down from a much higherclock frequency. For example, dividing1 MHz by 239 gives 4184.1 Hz or c5,.whilst dividing by 253 gives 3952.6 Hzor b5, and so on. The frequency ratiobetween these two notes is 1.05858,which is a reasonable approximation toa semitone. A more accurate approxi-mation could, of course, be obtained byusing longer divisors, but this would, ofcourse, entail raising the clock frequencyand using longer divider chains.For a more detailed discussion of thesepoints readers are referred to the article'Digital Master Oscillator' in Elektor 9and 10, January and February 1976.It is possible to design a master oscillator

Although primarily intended foruse in the Elektor piano describedelsewhere in this issue, the designof this master tone generator issufficiently universal to permit itsuse in a wide variety of electronickeyboard instruments.

L

using standard logic circuits, but fortu-nately ready-made master oscillatorsexist in the form of the General Instru-ments AV -1-0212 IC and the equivalentMotorola M 087 IC. These ICs operateon the principles described above.The complete circuit of the master tonegenerator is given in figure 1. A 1 MHzcrystal oscillator based on inverters N13and N14 provides the master clock fre-quency. The output of N14 is squaredup by a Schmitt trigger built aroundN15 and N16, the output of whichdrives the clock input of the masteroscillator. IC13. The clock frequency isdivided down by the ratios shown togive the twelve notes of the top octave,the twelve outputs being buffered byinverters NI to N12.The twelve buffered outputs are fed tothe top octave output connections ofthe master tone generator, and also totwelve, seven -stage CMOS binarycounters which provide the outputs forseven lower octaves, giving themaster tone generator a compass ofeght octaves, from C2 = 17.3 Hz toc = 4148.1 Hz, which should beadequate for even the most ambitiousinstrument. No vibrator is provided onthe master oscillator since the output ofthe master tone generator may he usedfor both the manuals and pedals of anorgan. If vibrato is added to the lowpedal notes of an organ the effect is

most unpleasant. A much better idea isto add vibrato and/or tremolo to themanual outputs using a Leslie speaker oran electronic vibrato system such as thePhasing and Vibrato' unit featured inElektor 20, December 1976.

ConstructionAlthough the design of the master tonegenerator may be unremarkable, theconstruction is worthy of note, sincethe entire circuit is mounted on a singlep.c. board only 12.5 cm x 16 cm. Thissmall size is achieved by the use of adouble -sided p.c.b., the layout of whichis shown in figure 2. To keep costsdown plated -through holes are notemployed, and all through connectionsare made using wire links. The padswhere such a through connection is

to be made are identified on the com-

9.10 - &dikter ieptember 1978 master tone generator

Figure 1. Complete ctrcurt of the master tonegenerator.

<

1

We Sc

0La Xj: Xi If-4 X.' V' "0'.* V V X.° t"

oa

0

0

090

o

I:0-60-..-.

no SC;) c°o ..® 0

V T0 0V V. 0 0

0V T

",

0VIe

C

r

i-1111111111D--0,9SON

...1-1171171

SS

MI*0111.

01. ems..

Figure 2. The complete tone generator cir-cuit is mounted on a single p.c. board only16 a 12.5 cm (EPS 9915)

Parts list

Resistors:

R1,R2 = 2k2R3 = 1 kR4 = 22 kR5 - 1 M

Capacitors:C1 = 27 pC2 - trimmer 451)C3,C4 = 47 nC5 = 47 p

Semiconductors.1C1 . IC12 CD4024IC13 - AV -1-0212 or M 087IC14 ...1C16 C04049

Miscellaneous:1 MHz crystal r= 30 pF

ponent layout by the symbol O. Theother components are soldered on thereverse side of the board only, i.e.no solder joints arc made on the com-ponent side of the board except to thewire links.An interesting feature of the board isthat the I? outputs for each octaveare brought out in the correct order sothat they can be connected direct tothe piano using 12 -way ribbon cable.Since the ICs used in the master tonegenerator are MOS or CMOS devicesspecial handling precautions should beobserved when constructing the unit.

TuningThe only tuning adjustment is to varytrimmer C2 until the oscillator fre-quency is exactly I MHz, when the tonegenerator will he tuned to internationalconcert pitch 1= 440 Hz). This can bedone by connecting a frequency counterto the output of NI6. Alternatively, if afrequency counter is not available thena1 can be adjusted to 440 Hz using atuning fork. This exact tuning procedureis not absolutely necessary for normaldomestic use. Since the outputs of thetone generator are locked to the clockoscillator the relative tuning is alwayscorrect, i.e. the tone generator is in tunewith itself, even though the overallabsolute tuning may he slightly sharp orflat

/11

000200000000 000000000000

lterat

.-29kkal

woouuogo-T

000000000000

Cu

00

0000

0

9.12 - slektor september 1978 electronic guano

«j1111 11IIThe principal difficulty in simulating theunique sound of a piano is caused bythe touch sensitivity of the keys. Whena key of an organ is depressed the loud-ness of the note produced is fixed,irrespective of how hard the key isstruck, and the note continues with thesame volume until the key is released.whereupon the note dies away more orless rapidly.The key action of a piano, on the otherhand, is much more complex. Thestrings of a piano are struck by felt -covered wooden hammers actuated bythe keys. The number of strings associ-ated with each note varies over thecompass of the keyboard. For example,at the bass end each hammer strikesonly a single string, whilst there may heas many as three strings per hammer inthe middle and treble registers. Thestrings for each note are equipped witha felt damper, which is raised when akey is pressed and falls back when thekey is released. The loudness of a noteplayed on a piano is determined by thefinal velocity of the hammer as it strikesthe strings, which is determined by howhard the key is struck. This also affectsthe harmonic content of the note, butthis is difficult to simulate electronicallyand only touch sensitive loudness isnormally provided on electronic pianos.When a key is struck on a piano thesequence of events is as follows: thedamper is lifted from the strings and thehammer strikes the strings with a vel-ocity dependent on how hard the keywas struck, then falls back. The notesounds with a rapid, percussive attackand then decays gradually over a periodof several seconds, unless the key is

released, in which case the damper fallsback and the note is terminated rapidly.A piano is also equipped with twopedals. The sustain pedal holds thedampers off all the strings, even whenthe key is released, and thus preventsrapid termination of a note with keyrelease. The soft pedal reduces the loud-ness of the notes, either by shifting thewhole keyboard sideways in the case ofa grand piano, so that the hammersstrike less strings, or by reducing thehammer travel in the case of an uprightpiano. The envelope which the outputof a piano follows is shown in figure I .

The amplitude dynamics of a piano canbe simulated electronically, though the

Although electronic organs haveexisted for many years, it is onlywith the advent of semi-conductors, and in particular,integrated circuits, that anelectronic simulation of a pianohas become possible. Over the pastfew years electronic pianos haverapidly grown in popularity,thanks largely to their compactsize and relatively low costcompared to a conventionalinstrument. The Elektor pianooffers all the facilities of aconventional piano, i.e. touchsensitive keying and pedals, plusa choice of three different voices,normal piano, honky-tonk pianoand harpsichord. A modulardesign allows the constructor tobuild a piano with as manyoctaves as required.

circuits to achieve this are considerablymore complicated than the equivalentkeying circuits of an electronic organ.However, the dynamic harmonic struc-ture of the piano sound is a differentproposition. As mentioned earlier theinitial harmonic content of a notedepends on the hammer velocity, andthe harmonic content also changes asthe note decays. In addition, due to themultiple stringing, a note may be pro-duced by up to three different strings,which do not vibrate exactly in unisonand so give rise to a complex pattern ofbeat notes.Sympathetic resonances may also occurbetween the strings for different notes.especially if the dampers are lifted byusing the sustain pedal, and resonancesof the frame and case of the piano alsocontribute to the tonal quality. It istherefore obviously not a feasible pro-position to attempt a complete simu-lation of the harmonic character of apiano, as this would be not only verydifficult but also rather expensive.Fortunately, it is possible to obtain afairly good approximation to a pianosound by using relatively simple tone -forming circuits, and although the elec-tronic piano is unlikely to replace theconventional instrument in the (classi-cal) concert hall, it nonetheless has anumber of advantages in other situ-ations.As mentioned earlier its cost is con-siderably less than that of a conven-tional piano, between a quarter and halfthe price of an upright type. The com-pact size and portability of an electronicpiano will appeal to itinerant musiciansand to those with limited dwellingspace, and finally, by using headphonesit is possible to practice on an electronicpiano without disturbing one's family orneighbours.

Block diagramA block diagram of the Elektor piano isshown in figure 2. As can he seen fromthis diagram the compass of the instru-ment is 5 octaves, as opposed to 7511octaves for a conventional upright pianoor 6% octaves for some compact,modern, upright instruments. The fre-quency range of the Elektor piano isfrom Clt = 69 Hz to c4 = 2092 Hz, i.e.the middle 5 octaves of a conventionalpiano. This somewhat restricted compass

I1° akietronie piano elektor septembar 1978 - 9-13

does not greatly detract from the versa-tility of the piano, since the top andbottom octaves of a piano are used onlytnfrequently, and it was felt that 5 oc-taves represented a reasonable compro-mise between performance, size andcost. Furthermore, 5 octave keyboardsare readily obtainable, whereas 6- or7 octave keyboards are not. However.since the construction of the piano ismodular it is a relatively simple matterto tailor the compass as required.The basic signals used in the Elektorpiano are squarewaves. These are ob-tained at the correct frequencies from adigital master oscillator, which generates

Figura 1. Envelope of a piano, showing howinitial loudness increases with key velocity,and the effect of the sustain pedal.

Figure 2. Block diagram of this Elektor piano.

Figure 3. Circuit of one envelope shaper wage.showing the internal arrangement of theAY -1-1320.

2X-talaseillaWr

&yids, per

mommallow

1111111111

11111111111111111111111111 1

3

I

pnstruinalt.loo Fore

IL

minion

GO°

°-]0114 I

ho

alm

9-14 - elktor septmber 1978electronic piano

loud pedal

07

100 200 300 400 500 600 700.61. Int* IMO

11,2;10.4444 4

the 12 notes of the top octave, and12 multi -stage frequency dividers, eachof which divides one output of themaster oscillator by 2, 4, 8, 16 etc. toobtain the lower octaves. This mastertone generator system is equally suit-able, not only for the Elektor piano, butalso for an organ or other keyboardinstrument, and has therefore beenmade the subject of another lseparate)article in this issue, see 'Master tonegenerator'.Each output of the master tone gener-ator is fed to a touch -sensitive keyingcircuit, one for each note of the piano.Each of these consists of a choppercircuit and an envelope shaper whoseoutput follows the attack -decay -releasecontour of a piano. The envelope shaperoutput is fed to the chopper circuitwhich is driven by the appropriate out-put of the master tone generator. Theresulting output of the chopper circuitis thus a squarewave whose amplitudefollows the output of the envelopeshaper.The touch -sensitive envelope shapers arebased on the General InstrumentsAY -1-1320 IC. Each of these ICs willprovide touch -dependent keying for 12notes, so one keying -circuit board con-taining an AY -1-1320 is required foreach octave of the piano.The outputs of the keying circuits foreach octave are summed and fed tofilters and voicing circuits which tailor

7

CD

tha!

0 -

111-111v-

1,14

-er

the harmonics of the output waveformto provide piano, honky-tonk or harp-sichord sound. Finally, the filter out-puts arc fed to a buffer preamplifierwhich allows the piano to he interfacedto a suitable power amplifier.

Envelope shapersFigure 3 shows the circuit of oneenvelope shaper. The section of thecircuit enclosed by the bold line is con-tained within the AY -1-1320 IC andeach IC contains 12 such circuits, so it iseasy to see why these ICs have revol-utionised electronic piano design.Each piano key is equipped with a set ofbreak -before -make changeover contacts,and the circuit senses key velocity bymeasuring the time taken for themoving contact to change over from itsrest position (normally closed) to con-nect with the normally open contact.The initial output voltage of the envel-ope shaper (before it begins to decay), isinversely related to the key travel timeand is therefore directly related to keyvelocity.Operation of the envelope shaper is asfollows: firstly, it should be noted thatthe AY -1-1320 is a MOS IC and oper-ates from a negative supply voltage. Theposition of the key contact is sensed bytwo comparators, kl and k2. When itis in the rest position the moving con-tact is connected to U2, the negativesupply voltage. This voltage is below the

thresholds of both comparators, so theoutput of kI is low (negative), whilstthe output of k2, which is an invertingcomparator, is high (zero volts). Ta andTd are turned on, whilst Tb is turnedoff. Assuming that Te is turned on viathe sustain input. the result Is that theoutput capacitor, Cb, is held in a dis-charged state via Rg. Td and Te (outputvoltage U is zero) whilst timing capaci-tor Ca is charged to U2 via Ta.When the key contact begins to movethe moving contact breaks its connec-tion with U2 and the input voltage tothe comparators now rises to Ub, whichis set by Ra, Rc and Rd. This voltage isabove the threshold of kI but below thethreshold of k2, so the output of klgoes high whilst that of k2 remains so.Td turns off and Ta also turns off, withthe result that Ca is no longer suppliedvia Ta and begins to discharge rapidlyvia Re to Ua, the voltage set at thejunction of Rc and Rd.When the moving contact closes, thecomparator inputs are connected tozero volts and the output of k2 goeslow, turning on Tb. The voltage on Cais thus applied via Tb to the gate ofsource -follower Tc. This voltage, slightlyattenuated due to the less than unitygain of the source -follower, appears atthe source of Tc and causes Ch tocharge rapidly.The voltage remaining on Ca by thetime the key contact closes, and hence

O electronic piano elektor september 1978 - 9.15

-IOc

NS*/

00

me so w$.

sly*CS0

wwt ma 4:),

0.° 0' Oe

.aat VD

00'III tow IT

CSOc

I Y

"D o 71 "1:1 "a "'a

Ye.qt °Ertl 'Olt 'Oct (ft111

enbe 11

C.

AY -1-1320

511 e11 IS .5 . .6jT T T

S.

TiL

T

- .77.1r -1 7.71: . Zvi 71 1wi 1:76I -1,7: ":4\ 771-

r. f. 1.11

.... : ....................

'

1

Cl i

T1 TS. TIPSI SS ICI CO MNS OS IQ CD NSID II 10 CD SDIci me de WMC11 C14 UMW V111 1111- Im111 1134411as an

the voltage transferred to Cb, is logar-ithmically related to the key velocity.The higher the key velocity the fasterthe key contact will change over and theless Ca will have discharged, so thegreater will be the voltage transferredto Cb.This is illustrated in figure 4, whichshows Ca discharging over about 60 mswhen the key is operated, and the out-put voltage U rising as Cb charges whenthe key contact closes.After the initial pulse, Cb receives nofurther charge from Tc, since the volt-age Uc decays very rapidly and Tc turnsoff. Cb now begins to discharge throughresistor Rh over a period of severalhundred milliseconds. The value of Rhis varied over the compass of the key-board, being smallest at the top end ofthe keyboards so that the high notesdecay more rapidly t han the bass notes

Figure 4. Showing the relationship betweenkey contact changeover and closing and thevoltages on Cc and Cb.

Figure 5. Illustrating the complete cycle of Ucand U from depressing the key to releasing it.

Figure 6. Chopper circuit used in the Elektorpiano

Figure 7. Chopper circuit suggested byGeneral Instrument. Although simple andcheap, this circuit has inadequate isolationin the 'off' state.

Figure 8. Complete keying circuit for oneoctave of the piano.

Table 1. This table gives the values of the dis-charge resistors R 1 to R 12 for each octave ofthe piano. In any one octave R I to R6 havethe same value and R7 to R12 have the samevalue.

Table 1.

lowest highest R1 ... RA R7... R12octave note (Uri note (Hz) WI 111)

1 c8i3 1109 c4 2092 150 k 180 k2 c/a 554 c 3 1046 220 k 270 k3 c41 277 c7 523 330k 390k4 c' 139 ct 262 470 k5 C'' 69 c 131 680 k 1320k

(see table I ).This relatively slow decay continuesuntil the key is released, when the out-put of comparator k I goes low and Tdturns on. Cb now discharges fairlyrapidly through Rg, Td, and Te, whichsimulates the action of the dampers in aconventional piano. However, if thesustain line is connected to zero volts,then Te is turned off and Ch continuesto discharge only through Rh. Thissimulates the lifting of the dampers bythe sustain pedal of a normal piano.The entire output waveform of theenvelope shaper is shown in figure 5, ona longer time scale than figure 4.

Chopper circuitThe output of each envelope shaper isfed to a chopper circuit, figure 6, whichconsists of an emitter -follower to act asa buffer and an electronic switch con-sisting of one quarter of a 4066 CMOSanalogue switch IC. The control inputof the switch is driven by the appropri-ate output of the master tone generatorso that the switch 'closes' and 'opens' atthe frequency of this signal. When theswitch is closed it has a resistance ofabout 80 ohms and the envelope volt-age is allowed through to the output.When the switch is open it has a resist-ance of several megohms and the envel-ope voltage is blocked. The output fromthe switch is thus a squarewave havingthe same frequency as the control volt-

9-16 - oloktor saptember 1978electronic piano

9

TOWHARP% -

PIANO A)C140#10

,41 12

age and an envelope the same as that ofthe envelope shaper output.This chopper circuit may seem com-plicated compared with the simple cir-cuit suggested in the General Instrumentsapplication notes, which is shown infigure 7. However, this circuit wastested in the Elektor laboratory and wasfound to have insufficient isolation inthe 'off' state, so that signal break-through occurred from all notes evenwith no key depressed, giving rise to anunpleasant background buzzing knownas 'beehiving'.This poorly designed chopper circuit hascaused many piano and organ manufac-turers to regard the AY -I-1320 withsuspicion, which is a pity, since theenvelope shaper IC performs its functionadmirably. Breakthrough in the Elektorcircuit, however, is minimal, and signalsuppression is further enhanced bymuting circuits in the filters, whichsuppress any signals or noise which arebelow a preset level. The complete envel-ope shaper circuit for one octave of thepiano is given in figure 8. The outputsof the individual envelope shapers aresummed by R25 to R36 and PI adjuststhe overall level for each octave.

Filter circuitsFiltering in the Elektor piano is carriedout in three stages. In a conventionalpiano the harmonic content of the notesvaries over the compass of the instru-

ment, low notes having a greater har-monic content than high notes. In theElektor piano preliminary filtering iscarried out by means of five lowpassfilters, one for each octave, as shown inthe block diagram of figure 9. Thefilters for the higher octaves have lowerturnover frequencies than the filters forthe lower octaves, with the result thatthe harmonic content of the higheroctaves is greatly reduced compared tothe lower octaves.The five filter outputs are then fed todiode muting circuits, which suppressany residual signal breakthough or noiseand pass only 'genuine' signals from thekeying circuits. The muting circuits arefollowed by four bandpass filters withdifferent centre frequencies. By feedingeach octave in different proportions totwo or more of these filters the har-monic structure of every octave can hetailored fairly accurately. For example,the low notes are to have a high har-monic content. This is achieved byfeeding a large proportion of the lowoctave output into the bandpass filterwith the highest centre frequency, andrelatively smaller proportions into thefilters with lower centre frequencies, theresult being that the higher harmonicsare boosted relative to the fundamentaland lower harmonics.The four bandpass filter outputs arethen fed to voicing filters, which pro-vide a choice of piano, honky-tonk or

harpsichord sound. Finally. the outputsof the voicing filters are mixed and fedto a preamplifier incorporating tone andvolume controls, the output of whichcan be fed direct to a power amplifier.A soft pedal or expression pedal mayalso be incorporated into this stage, aswill he described later.

Complete filter circuitThe complete circuit of the filters isgiven in figure 10, all the filters andtone controls being based on TexasTL 074 quad FET opamps.The five lowpass filters which performthe preliminary filtering are constructedaround Al to A5. The muting circuitsat the output of these filters consist ofDI to D5 and their associated resistors.The outputs of Al to A5, connected tothe anodes of these diodes, are at zerovolts with no input signal, whilst thecathodes of the diodes are biased toabout -0.16 V. This means that theoutput signals from Al to A5 mustexceed approximately 0.4 V before DIto D5 will become forward -biased andwill allow signals to pass. Any break-through or noise signals below this levelwill be blocked since they will beinsufficient to exceed the knee voltageof these diodes.The four bandpass filters each consist ofan opamp with a twin -T selective net-work in its feedback loop. Each of thefive octave outputs is split and fed in

' electronic pianoelektor september 1978 - 9.17

1101111.011

CAI

-

00.s

Do -T0t t±)I

10211 ---AF

.11 "13

4= -0- 3+. b tt

0

r-91b b0 -41 -

= 0-0

i±) it b b

.0 . i7D b -

'0

a, 00 Cy Ttr 1)46 66-.C1- ?LW L

AO All CI %OM 01. 0140I Oil 0114140C.42 UP 1.0 000

b tu b

1001.1001

b

./111=101 0E0

.0400 pa 111101 V* 4

different proportions to two or more ofthese filters to obtain the correct har-monic content. The outputs of the fourbandpass filters are then fed throughpassive voicing filters to a summingamplifier A6, and hence to a Baxandall-type tone control stage based on A7.The output of the tone control is finallyfed to an amplifier with presettablegain. A8. at the output of which is thevolume control, P8.Several possibilities exist for this con-trol. It may simply be a panel -mountedcontrol operated by a knob or it may heincorporated into an expression pedal.This latter arrangement is felt to offergreater versatility than a soft pedal,which merely gives a fixed degree ofattenuation by means of a foot operatedswitch. However, a soft pedal may beincorporated if desired by connecting apreset in series with P8, which can beswitched in and out by a footswitch.PS can then be retained as a panel -mounted master volume control. Pedalswitches for sustain and soft pedals areavailable in ready-made housings from.organ and piano component suppliers.A p board layout for the filter circuitsn green in figure I I.

Power supplyThe master oscillator used in the mastertome generator requires supply voltagesof -13 and -263 V, whilst the envel-ope shapers require a single, -26.5 V

Figure 9. Block diagram of the filter circuitsfor the complete piano.

Figure 10. Complete circuit of the filters,voicing circuits and tone control preamp.

supply. The opamps require both a

positive and negative supply, so they areoperated from the -13 V rail and a+14.5 V rail. The -13 V rail is also usedfor the chopper circuits. A powersupply circuit to provide these threevoltages is given in figure 12. No strin-gent demands are placed on the stabilityof these supplies, so simple zener/tran-sistor regulators are adequate. However,extensive decoupling is provided on thesupply lines to obviate the possibility ofinterference breakthrough. The powersupply p.c.b. is shown in figure 13.

ConstructionThe keying circuit for one octave isaccommodated on the printed circuitboard shown in figure 14, so five ofthese boards are required for the com-plete piano. The key contacts them-selves may be ready-made single -polechangeover contact blocks such as theKimber-Allen type G.1, or may be home-made. If ready-made contacts are usedthen the 'tail' of the moving contact ofeach switch should be soldered directto the pad provided on the p.c. board.The tail of the normally -open contactshould be soldered to a zero -volt busbarmade of stiff wire, whilst the normally -closed contact should be soldered to asimilar busbar connected to -26.5 V.For home-made contacts the movingcontact may be made of gold-platedphosphor -bronze wire, whilst the func-

9.18 - elaktor september 1978electronic piano

Parts list for figures 10 and 11.

Resistors:

R1,R3=68kR2,R5,R8,R11,R14,R17,

R20,R23,R26,R29 1k5R4,R10,R16,R22,R28 120kR6,R12,R30,R31,R32,R33,R36,

R37,R43,R50,R56,R57,R60.R61,R63,R65,R67,R7ELF182 = 10 k

R7,R9 - 82 kR13,R15,R25,R27,

R38,R39,R40,R42 = 39 kRI8,R24,R49,R51,

R52,R53,R55,R79 - 22 kR19,R21 - 33kR34 - 560 kR35,R58 - 8k2R41,R47 = 390 kR44,R45,846,R48 - 27 kR54,R71,R80 - 220 kR59,R62 6k8R64,1166- 15kR68,R69,R70 - 47 kR72,R73.R74 - 12 kR75,R76 3k9R77 - 820R81 - 100kPI,P3,P4 = 25 k 122 kl presetP2 = 50 k 147 k) presetP5 100 lin. pot.P6 - 500 k 1470 k) lin. pot.P7 - 250 k (220 k) presetP8 x. 10 k log. pot.'

Capacitors:Cl . C5,C35,C45,C46 = 100 nC6,C13,C33 - 15 nC7,C8,C11,C12 - 6n8C9,C41 4n7C10 - 2n2C14,C32 = 1 nC15,C36,C37,C39,C40 - 47 nC16,C17,C21.C22,C27,

C28,C29,C31 = 10 nC18,C26 = 560 PC19,C23,C24,C25,C30 = 22 nC20 = 68 nC34 - 470 nC38 - 1

C42,C43 = 1 si tantalumC44 = 220 n

SemiconductorsD1 ... D5= 1N41481C1,1C2,IC3 - TL 074 (Texas),

XR42121Exar)

Miscellaneous:

S1,52,S3 - single -pole on -off switch

If an 'expression' pedal is required, P8should be pedal controlled. Alterna-tively, if a 'soft' pedal is required, a 10 kPreset should be included between P8and C42, with a pedal -operatednormally -closed switch in parallel withthis preset.

Figure 11. Printed circuit board and com-ponent layout for the filter SectionIEPS 9981).

iiescaronie pianoslaktor septembor 1978 - 9-19

p 01 R121.0CHP/101-C) Cailaicja,

4,_9_, CH F4:80 ell 0 9"4"

33 aG." -.03 'Tr9_ Rn

11ia' is :4' 22

-6--i-6-4-a-HD 00_4F0

TQTI QT IY

it 21 0 -.....1.2.-

-6-' vAV

lions of fixed contacts and busbars maybe performed by rods of palladium orrhodium. All these materials are avail-able from organ component suppliers.The wiring to both types of contact isshown in figure 15.The actual mounting of the contactblocks and envelope -shaper boardsdepends on the type of keyboard used.The type of keyboard with actuatinglevers extending behind the keys maybe screwed directly to a baseboardand the contact blocks and p.c. boardsmounted behind it. The keying -circuitboards are designed to be mountedcopper side up to facilitate soldering tothe contact tails after the contact blocksand boards have both been fixed inposition, as shown in figure I6a. How-ever, since the envelope shapers areidentical with the exception of thedischarge resistors RI to RI2 it is alsopossible to turn the board over end -for -end so that the component side is upper-most, as shown in figure 16b. Input I

then becomes input 12, 12 becomes 1,2 becomes I I and so on. Resistors RIto R6 and R7 to R12 must also betransposed.If the keyboard has actuators under-neath the keys (e.g. Kimber-Allen SKAkeyboards) then the contact blocks andboards must be mounted beneath thekeyboard frame with the copper side ofthe keying boards facing downwards.The comment about transposition ofthe inputs and RI to RI2 then alsoapplies.Whichever way the keying -circuit boardsare mounted it must he rememberedthat the lowest note of each octave isC= and the highest note C, so theboards and contact blocks must be

mounted to align with the correct keys.The lowest note, C, of the five -octaveC -C keyboard is not used, as this wouldrequire an extra envelope shaper IC forone note, which is uneconomic.The master tone generator (see ac-

companying article) is built on a singlep.c. board which can easily be mountedclose to the keying -circuit boards andconnected to them using ribbon cables.The filter section and its associated con-trols are also mounted on a singleprinted circuit hoard which can easilybe fixed behind a panel on the instru-ment console. A complete wiringdiagram for the piano is given in fig-ure 17.

Tuning and voicing adjustmentTuning the piano involves nothingmore than trimming the clock fre-

quency of the master oscillator, whichis dealt with in the article on the mastertone generator. However, even thisprocedure is not absolutely necessaryunless the piano is to accompany otherinstruments or vocalists, since all notesof the master tone generator are cor-rectly tuned relative to one another,even though the overall tuning of thepiano may be slightly sharp or flat.More important is the adjustment of

9 20 - tilektor september 1978electronic piano

12'-fre.Cs

Noe 05*

CS

MIN MIMI000. 4IW*IV SOY

Figure 12. Power supply for the completepiano.

Figura 13. Printed circuit board and com-ponent layout for the power supply.IEPS 9979).

Figure 14. Printed circuit board and com-ponent layout for one octave of the keyingcircuit. IEPS 99141.

9979

u2C)25elir

Parts list for figures 12 and 13.

Resistors:

RI = 4k7R2,113 470 ft/2 W

Capacitors:Cl 220p/63 VC2,C4,C13 47 p/25 VC3,C5,C8,C10,C12,C14 100 nC6 = 4700 µ163 VC7C9,C11 - 47 kr/40 V

01 . 04 - 1N4002TI BC 14112. T3 BO 242

C cis 9111.in OW NM MOMIOC.. sr ICKS,

9914 IS

13V 0 U3 ® 141V

Semiconductors:DI . . D4 1N400205 , zone. 15 V/400 mWD6 zener 27 V; I W07 - zener 12 Vi400 mWD8,139 1N4148T1 BC 141T2,T3 = BD 242

Miscellaneous:

Tr mains transformer with2 x 30 V /500 mA secondaries or30-0-30/500 mA sec.

Si = double -pole on/off mains switch

efektronie piano

14

elektor usptember 1978 - 9.21

111:'.C541:3;9\0

;J1(

777/8

a r

IC1 IC r'

O»

4°5 " 14 i°4C34C1(11(144)))1"11W T -re

P TZI T r,

.r 7 T9 Tict rite.T.1.2

Ei r_ ...I.I

U 1

6UP

Parts list for figures 8 and 14.

Resistors:

RI ... R12 = see text and table 1R13 ... R24 = 47 kR25 ... R36 - 6k8R37 = 100 tiP1 - 10 k preset

Capacitors:Cl ... C12 - 470 n polycarbonate

or polyesterC13 . C24 2µ2/35 V

Semiconductors:ICI ... IC3 4066IC4. AY -1-1320T1 ... 712 - TUP

the piano voicing, which is carried outby ear. Preset PI on each envelopeshaper board allows the overall outputlevel from each octave to be adjusted,partly to compensate for variations inthe characteristics of the envelopeshaper ICs and partly to allow theloudness of each octave to be adjustedto that of a conventional piano.Presets PI to P4 in the bandpass filtersdetermine the tone of the piano, whichcan be varied from soft and muffled toa hard, steely sound. This, of course,is a matter of personal taste. The voicingadjustments should, of course, be carriedout with the tone controls in the flatposition. Preset P7 should be used to setthe gain of A8 to suit the power ampli-

fier used, e.g. so that with the volumecontrol at maximum the output is juston the point of clipping when a com-plex chord is played fortissimo. Finally,the soft pedal preset, if fitted, should beused to give the desired degree ofmuting when the soft pedal is pressed.As a final comment to avoid complaintsfrom the musical purists, it should benoted (no pun intended) that the pianodoes not give an accurate simulation ofthe amplitude dynamics of a harp-sichord. The strings of a harpsichord areplucked by quills rather than struck,and the loudness of a note is determinedlargely by the tension at which the quillreleases the string, rather than by keyvelocity.

9.22 - elektor september 1978 electronic piano

Bulk component list for 5 -octave piano preset 250 1/4 (220 k)(including master tone generator( potentiometer

Resist ors10 k log

potentiometervalue: number: 100 k in.

potentiometer100 5

820 I-1 1500Ir (470 k) lin.

1k1k5 10 Capacitors

2k2 2 value number:3k9 2 27 p 1

4k7 1 47 p 1

6k8 62 560 p 28k2 2 to 210k 19 2n2 1

12k 3 4n7 215k 2 6n8 422 k 9 10n 827 k 4 15n 333 k 2 nn 539 k B 47 n 747 k 63 68 n 1

68 k 2 100 n 1482 k 2 220 n 1

100k 1 470 n 61120 k 5 1µ 1

150 k 6 114 tantalum 2180k 6 2412/35 V 60220 k 9 47 u/25 V 3270 k 6 474/40 V 3330k 6 220 55/63 V 1

390 k 8 4700 µ/63 V 1

470 k 6 trimmer 45 p 1

560 k 7

680 k 6 Semiconductors:820 k 61 M 1

type: number

470 1112W 2 4024 12

preset 10 k 5 4049 3preset 25 k 122 k) 3 4066 15preset 50 k 147 k) 1 AY -1-0212, M 087 1

AY -1-1320TL 074, XR 4212TUPBC 141BD 2421N40021N4148zener 12 V/400 mWzener 15 V /400 mWzener 27 V /1 W

53

601

2

471

1

1

Miscellaneous:

crystal 1 MHzmains transformer 2 x 30 V!500 mA3 x SPST switchdouble pole on/off mains switch5 -octave CC keyboard

I Figure 15. The key contacts may be ready -assembled or homemade. In either case themoving contact is soldered to the keyingcircuit p.c board whilst the foxed contactsare connected to ground and -26.5 V bus -bars.

Figure 16. Depending on the type of key-board used, the keying circuit board may bemounted either wry up to facilitate solderingto the contact tails. Depending on which wayit is mounted it may be necessary to transposethe inputs and R1 to R12

Figure 17. Wiring diagram for the completepiano. Connections between the master tonegenerator and the keying circuits may bemade uung ribbon cable

electronic piano eiek for september 1978 - 9-23

17

tT

m

CN

C

9-24 - elektor soptomber 1978temperature -controlled soldenng iron

temperature-controlledsoldering iron

Since the days when soldering ironswere heated up on gas rings, the designof this virtually indispensable piece ofequipment has come a long way. Thereis now a wide variety of different typesof iron available, allowing power ratingas well as size, shape and composition ofthe bit to be selected to suit a particularapplication. Despite this plethora ofdifferent irons, it is nonetheless possibleto discern two basic categories, namelycontinuous heat and temperature -con-trolled soldering irons. With the formertype, the heating element is connectedcontinuously to the supply, with theresult that the iron tends to run veryhot when not being used.This means that the first joint madeafter the iron has been left standing maybe too hot, thereby incurring the risk ofa bad joint or of damage to delicatecomponents. If one attempts to combatthis problem by using a lower poweriron, there is the danger that, underheavy load conditions, it may be unableto supply sufficient heat and will makea dry joint. A further disadvantage ofcontinuous heat irons is that theirtendency to overheat shortens theeffective life of the bit and causes areduction in the heating capability ofthe iron.Temperature -controlled irons on theother hand suffer from none of thesedrawbacks. The only reason that theyhave not completely replaced continuousheat types is the fact that they costmuch more. However, with the currenttrend towards ever smaller and moresensitive components, the decision topurchase a temperature -controlled ironmay well prove a worthwhile long-terminvestment (particularly if one considerssaving the cost which results frombuilding the control unit oneself).Thermostatically controlled solderingirons must not only be able to maintaina constant bit temperature (to within afew degrees Centigrade), it must also bepossible to vary the soldering tempera-ture to suit different requirements.Designing a suitable control unit, whichboth meets the above conditions andyet is a reasonable financial propositionfor the amateur constructor, is no easymatter. However, the circuit describedin this article adequately fulfils all the

Electronic temperature -controlledsoldering irons offer a number ofadvantages over continuous heattypes: delicate components areprotected against thermal damage;they permit the use of higherwattages, thereby eliminating thedanger of dry joints when workingunder heavy load conditions; andfinally they increase the life ofboth heating element and bit.The following circuit is for athermostatic control unit, which isboth easy to build and usesstandard components.soldering irons containing abuilt-in heat sensor are readilyavailable from a number ofdifferent manufacturers.

desired design critena at a price which isroughly halfway between the cost of aconventional continuous heat iron andthat of a commercially available tem-perature -controlled model. The controlunit is designed for use with a readilyavailable soldering iron incorporating aheat sensor in the shaft adjacent to thetip of the bit.

Electronic control unitThe principle of the electronic thermo-static control unit is illustrated in theblock diagram shown in figure I .

A sensor mounted in the element asnear as possible to the bit tip provides avoltage which is proportional to the bittemperature. This voltage is thencompared with a (variable) referencevoltage on the other input of a com-parator, the output of which is used tocontrol a switch which regulates theflow of current to the heating elementin the iron. Thus, when the sensorvoltage is lower than the reference value,the switch is closed, current flows to theheating element and the bit temperaturerises; once the desired temperature isreached, the comparator output changesstate, opening the switch and therebycutting off the flow of current to theheating element. The bit temperaturethen falls until the threshold voltage ofthe comparator is again reached and thecontrol switch is opened. In this waythe temperature of the bit can bemaintained within a certain fixed range.The amount of hysteresis between achange in temperature and the corre-sponding change in sensor voltage isdetermined by the thermal inertia of thesensor itself and the thermal conduc-tivity of the bit (which in turn is deter-mined by the size and composition ofthe bit).The deviation from the nominal bittemperature as a result of the hysteresisof the control system is illustrated infigure 2. As can be seen, the bit tem-perature oscillates about a presetnominal value; the steepness of therising edge of the triangular waveform islargely determined by the output powerof the heating element, and that of thefalling edge by the rate at which heat islost to the atmosphere, solder, p. c. b.

1

temperature -controlled soldering iron islektor september 1978 - 9-25

etc. In practice however, the bit tem-perature only deviates very slightly fromthe desired nominal value, so that it is infact possible to speak of an averageworking temperature of the iron.As far as the choice of heat sensor isconcerned, various possibilities comeinto consideration. The firm Weller, forexample, manufacture a heat sensorwhich utilises an unusual property ofmagnetic materials. Above a certaintemperature, known as its Curie point, aferromagnetic material loses the prop-erty of magnetism. The bit of a Welleriron contains a slug of magnetic material,which, when the iron is cold, attracts amagnet. This in turn closes a switch andapplies power to the heating element.When the temperature of the bit reachesthe Cune point, the slug ceases toattract the magnet, causing the switchto be opened. The only disadvantage ofthis system is that a different bit con-taining a ferromagnetic slug with theappropriate Curie point is needed tochange the soldering temperatures.Other manufacturers employ heatsensors consisting of a thermocouple orof an NTC- or P IC thermistor, usuallyas part of a bridge circuit. One branchof the bridge is formed by a variableresistor with which the bridge is bal-anced. In practice this means that thetemperature range of the bit is deter-mined by the range of the resistor.Of the above -mentioned types of sensor,the thermocouple represents the bestchoice. The reasons for this are clearwhen one compares it with temperature -dependent resistors Firstly, the dimen-sions of the thermocouple are smallerthan those of an NTC or PTC thermistor,which means that is is easier to mountclose to the tip of the bit, and also that,because of its reduced mass, it respondsmore quickly to changes in temperature.

`The response of a thermocouple (volt-age as a function of temperature) is, asfigure 3 clearly shows, linear over a widerange of temperatures. NTC- and PTCthermistors, on the other hand, exhibita far less linear characteristic. Further-more, a thermocouple has no quiescentcurrent flow to speak of, and hence willnot generate any heat itself. The finalpoint in its favour is the lower cost ofthermocouples, a not insignificant factor

Figure 1. Block diagram of an electronicthermostatic control unit for soldering irons.The voltage from the heat sensor is comparedwith a variable reference voltage. The outputof the comparator controls a switch which IsUsed to turn the flow of current to theheating element on and off.

Figure 2. The response of a typical thermo-static control unit. The temperature firstclimbs to the desired (preset) value. However,once it reaches this value the temperaturecontinues to rise, due to the inherent hyster-esis of the system, before the interruption ofcurrent to the heating element begins to takeeffect. Similarly, when the temperature hasfallen to the nominal value, it will continue todrop slightly before the renewed flow ofcurrent to the heating element can begin topush the temperature back up. Thus theactual bit temperature of the iron tends tooscillate about the nominal 'controlled'temperature. However, in practice thesevariations in temperature are sufficientlysmall not to significantly affect the perform-ance of the iron, which remains at a more orless constant 'average' temperature.

when temperatures of the order of400°C are involved.

The Elektor control unitIn view of the above -mentioned points,an iron which was both readily avail-able and which incorporates a thermo-couple as heat sensor was taken as thestarting point of the Elektor control unit.Several manufacturers in fact distributesuitable soldering irons without the ac-companying control unit. For example,the firm Antex produce a 30 Wsoldering iron (the CTC) which includesa thermocouple, as well as a 50 W modelXTC) which should be available shortly.

Ersa are another company who have asuitable 50 W iron (TE 50).In order to ensure the complete re-liability of the Elektor control unit, itwas in fact sent to Antex for assessment.Their verdict was summarised as follows:'The performance of the sample testedshould be perfectly adequate for theHome Constructor'. Furthermore, thecontrol unit can also be used withsoldering irons from most of the othermanufacturers, even if they containNTC- or PTC thermistor sensors,although in that case certain changeswill have to be made to the circuit.Without entering into the theoreticaldetails, it should be noted that differentcombinations of materials can be usedto construct thermocouples, and thateach will deliver a different output volt-age for a given temperature. For theirCTC and XTC models, Antex use a K -type thermocouple, which is composedof nickel -chrome and nickel -aluminium.The response shown in figure 3 wasobtained using this type of thermo-couple.

Circuit diagramThe complete circuit diagram of thethermostatic control unit is shown infigure 4. Despite the small number ofcomponents used, the operation of thecircuit is somewhat involved, and forthis reason figure 5, which contains anoverview of the waveforms found at thetest points shown, is included to facili-tate explanation.The first problem which arises is the

9-26 - itlektor september 1978 temperature -controlled soldering Iron

50UirrAt)

45

40

35

30

25

20

15.

10.

5.

180 280 300 abo 500 600 700 800 900 1000

Tit9162.3

choice of switching element to regulatethe flow of current to the iron. The useof a relay involves several drawbacks(contact burning, contact bounce etc.)which can be avoided by employing anelectronic switch such as a triac. Anadditional advantage of a triac is thatthe switching point can be controlledwith a high degree of accuracy, i.e. in

order to reduce the switch -on surgecurrent and r.f. interference to a mini-mum, the triac can be triggered at thezero -crossing point of the AC waveform.This is in fact the arrangement adoptedin the circuit described here.

R4, D3, TI and the emitter resistors ofTI form an adjustable constant currentsource. D3 is a LED used to set the DCbase bias voltage of TI, but since itdraws very little current it will hardlylight up at all. The advantage of thissomewhat unusual approach is that theLED possesses the same temperaturecoefficient as T I , hence the stability ofthe current source is unaffected byvariations in temperature. This is onlytrue, however, if the ambient tempera-ture of the circuit does not rise too farabove normal room temperature, sincein that case the temperature coefficient

Figure 3. The temperature voltage character-atm of a nickel-chrome/nickel aluminiumthermocouple (the type used in e.g. Antestoldering irons/. As is apparent. the responseis virtually linear over the temperature rangeof this particular application.

Figure 4. The complete circuit diagram of theElektor thermostatic control unit. The modi-fications needed for the 40 V version are

listed in table 1.

Figure 5. Pulse diagram of the various signalsobtained at the test points shown in figure 4.

of the LED will cease to match that ofT I . Thus, if when the circuit and trans-former have been mounted in a case, thetemperature should rise by more than30°C, 133 should be replaced by an 8k2resistor. This step will obviously be

necessary if the soldering stand is to be

mounted on top of the control unit case.The current through P2 and R6 can bevaried by means of PI. P2 determinesthe amplitude of the reference voltageat the inverting input of ICI. Thethermocouple is connected across thenon -inverting input of ICI and thejunction of R3/R6. Thus the voltage

temperature -controlled soldering iron elaktor saptember 1978 - 9.27

e

e

e

,e

e

;e

1

62Mo.

OV

FF2J

FF2

n

/s, I

st

t- -

,I I

/OG.0

ol

1110.

It

II

111

e, -.--Y.IIII

Pty

;

!I '

it

triII

tPa

II

jiIl

81 t

t

9052 5

I

difference at the inputs of the compara-tor equals the difference between, onthe one hand, the voltage droppedacross R6 plus the resistance of P2, andon the other hand, the voltage devel-oped by the thermocouple. That is tosay, it virtually equals the thermocouplevoltage.If the soldering iron is cold, the thermo-couple voltage is very small, so that theoutput of ICI is low. When the tem-perature of the iron rises, the thermo-zouple voltage, and hence the voltagedifference at the comparator inputs,also rises, until the output of the

comparator swings high.ICI is followed by a Schmitt trigger, theoutput of which goes low when theinput exceeds approx. 3.2 V, and highwhen it falls below roughly 2.1 V. Thisarrangement could be used directly tocontrol the triac were it not that wehave to first ensure that the load isswitched at the zero -crossing points ofthe transformer voltage. To achieve this,one or two extra provisions are required.The transformer voltage (Utr in figure 5)is connected to the input of N3 via apotential divider, R9 and RIO, one endof which is connected to Ow stdbtir,ed

5.6 V supply rail. This means that thevoltage at point I (the input of N3)exactly tracks the transformer voltage,whilst remaining 2.8 V 'up' on the lattersee figure 5). The portion of waveform

above 6.2 V and below -0.6 V is shownas a dotted line, since CMOS Schmitttriggers contain clamping diodes whichprotect the inputs from voltages whichexceed these limits.The advantage of the 2.8 V positiveoffset is apparent from figure 5, since itmeans that when the transformervoltage is zero, the voltage at point 1 is2.8 V; since the Schmitt trigger changesstate at the threshold values of 2.1 Vand 3.1 V, we can say that, in spite ofthe hysteresis, it is only triggeredaround the zero -crossing point of thetransformer waveform (the small devi-ation from the ideal switching point ofexactly 0 V can be eliminated bymaking R9 variable and using a scope toadjust it to the correct value; in practice,however, this small error is of littlesignificance and does not materiallyaffect the operation of the circuit).When both inputs of N3 are high (i.e.greater than 3.1 V), the output is low,and since N4 is connected as an inverter,its output will be high, with the resultthat C2 will be discharged. If point 0then goes low, since C2 is still uncharged,point 0 will also go low, causing C2 tocharge up via R I I. The time constant ofRI I /C2 is 18 ms; shortly before thistime is reached the voltage at pin 12 ofN3 will have reached the logic 'I'threshold and since, at that moment,pin 13 has once more been taken high,the output of N4 is also returned high.Since capacitor C2 is already charged,the voltage across it would continue torise, but for the clamping diode in N3.The capacitor is rapidly discharged(figure 5 0), and a new cycle begins.The signals at points 0 and 0 form theclock signals for flip-flops FF I and FF2.The 3 -inputs of these flip-flops areconnected to points 0 and 0, where thevoltage is determined by the tempera-ture of the iron, whilst the K -inputs areconnected to ground. Only when theJ -inputs are high can the clock pulseshave any effect and change the state ofthe flip-flops. Since the voltage at point0 is an inverted version of that at point0, when the former goes low the firstpositive -going edge at point 0 will takethe 0 output of FF I (point 9) low,causing T2 to turn off and the triac tobe triggered. The soldering iron thenbegins to heat up, so that the voltage atpoint dD rises until it reaches the triggerthreshold of NI. When that happens NIchanges state, taking point 0 low andpoint 0 high; the next positive goingpulse at point 3 will take the Q outputof FF2 high and reset FF1, therebytaking point 0 high and resetting FF2.Thus T2 is turned on and the triacturned off, interrupting the flow ofcurrent to the heating element in theiron. The temperature of the iron willfall until the lower threshold value of

9.28 - eloktor sieptember 1978 temperature -controlled soldering Kan

Ports list

Resistors:

RI = 2k2 1 WR2 - 1 MR3 = 2k2R4 - 22 kR5 4k7R6 - 8211R7,R8,1112 - 10 kR9,R10 - 33 kR11 - 180 kR13 1 k 1 WR14 . 100 IIR15 - 100 kP1 - preset potentiometer 10 kP2 potentiometer 100 k linearP3 - preset potentiometer 100 k

Capacitors:Cl - 470 u/40 VC2 - 100 nC3- 10u/10 VC4 - 100 nC5 22 n

Semiconductors:DI - 1N4001D2 5V6/400 mWD3 LED redD4 - LEDT1 BC 559CT2 - BC 5476Tri 2 A/100 V (or 4 A/400 V/ICI - 3130IC2 - 4093IC3 - 4027Miscellaneous:

transformer 24 V/2 Afuse 0.25 A slow blowsoldering iron with built-in heat

sensor, e.g. Antex type CTC orXTC

soldering stand.

NI is reached, whereupon a new cyclewill begin.Those phases during which current isfed to the iron (i.e. when the triac isconducting) are indicated by LED D4lighting up.The waveforms shown in figure 5 do notexactly coincide with those obtained inpractice, since the noise at the inputs ofICI (which in no way affects theperformance of the circuit )has, for thepurpose of clarity, been omitted fromthe diagram.

ConstructionFigure 6 shows the track pattern andcomponent overlay of the p.c.b. for thecircuit of figure 4.Constructing the control unit shouldnot present any major problems. Con-nection points A ... E marked on theboard correspond to those shown infigure 4, and are in fact the holes forconnections to the soldering iron.Figure 7 shows the DIN -plug of theAntex CTC soldering iron with detailsof the correct pin connections andcolour of the leads.In principle the triac should require noheat sink; however if the circuit ismounted in a small case and the iron isoperating under heavy load conditions,then the use of a heat sink is stronglyrecommended I not to mention venti-lating the case). In fact every effortshould be made to prevent any rise inthe ambient temperature of the circuit,since, as was already mentioned, thiswill have an adverse effect upon thetemperature coefficient of the constantcurrent source.

F pure 6. Track pattern and componentlayout of the p.c.b. for the circuit of figure 4.(EPS 9952).

Figure 7. The Ante soldering iron is fittedwith a DIN -plug. Pins A . . E correspond toconnection points in the circuit of figure 4.Pon E is shown as a ground point and connectsto the metal body of the iron.On no account should the 0 V rail of thecontrol circuit be connected to mains earth,although it is permissible to earth a metal case.of one if used.

Table 1.

40 V -VersionUtr 40 V/1 AR1 4k7,3WCl 470 u/63 VR13 2k2, 3 W12 BC 546

temperature -controlled soldering iron alektor 'optimiser 1978 - 9.29

The photo shown on the first page ofthis article is a prototype model of theElektor control unit. For exhibitionpurposes the unit was housed in perspex.The soldering stand shown in this photois not particularly suited for low powerirons, since the contact between theiron and the metal rings leads to con-siderable heat loss and hence to the ironbeing switched on and off with excess-ive frequency. Soldering stands whichavoid direct metal -to -metal contact withthe iron should be given preference.These can be bought separately frommost electronics shops.

Adjustment procedureThe setting up procedure for the controlunit is as follows: -Firstly, with the soldering iron discon-nected, the inputs of ICI are shortedtogether. The offset voltage is thenreduced to a minimum by adjusting P3until D4 either just lights up or is just

extinguished (depending upon whichstate it assumes when power is applied).Next, the short is removed and thewiper of P2 is turned fully towards R6(anticlockwise). The soldering iron isthen plugged in and the tip is heldagainst a length of solder. Althoughsolder melts at approx. I89°C (60/40alloy), at around 185°C it exhibits a'plastic' consistency. By very graduallyadjusting PI , it is possible to set thetemperature of the iron such that thesolder is in this plastic state, just on thepoint of melting (185°C). P1 should beadjusted in small steps, always allowingthe temperature of the iron to stabilisebefore testing it against the solder andperforming another adjustment.By means of P2. it is then possible tovary the temperature of the iron be-tween 185°C and approx. 400°C. P2can be calibrated using the followingequation:

P2IT-v 185 + - x 85°C (P2 is in Si)

82

In conclusionAs was already mentioned, the proto-type model of the control unit wasdesigned for use with the CTC or XTCsoldenng iron from Antex. However itcan also be used with other types ofiron, particularly if they are providedwith a thermocouple heat sensor. If thisis the case, and if the iron operates off24 V, then it can be connected direct tothe Elektor control unit. In the case ofan iron with the same operating voltagebut which employs a different sort ofsensor, the situation is a little morecomplicated. With a PTC thermistor, D3and D4 should be omitted, TI replacedby a wire link between the emitter andcollector connections, and the value ofR2 altered accordingly. The sameprocedure holds good for irons incor-porating an NTC thermistor, with theexception that R2 and the NTC shouldbe transposed.In the case of an iron employing athermocouple and operating from a

40 V supply, the modifications shownin table I should be adopted.The above -described control unit is thussuitable for use with a wide variety ofdifferent types of soldering iron, andrepresents a considerable saving in costover commercially available models.The final point worth noting is that thecircuit can not only be used to regulatethe temperature of soldering irons, butcan be adapted for a number of otherapplications requiring a thermostaticcontrol unit, such as. eg. clothes irons,ovens, central heating etc.

9-30 - elektor september 1978 car start boost.

car startbooster

As all car owners know, during thewinter one has to be especially carefulto ensure that the battery voltage is notallowed to fall too low, otherwise thereis the very real danger of the car failingto start. The main reasons for this arefirstly, that the low temperature of theelectrolyte increase the effectiveinternal resistance and hence reducesthe capacity of the battery. At -20°Cthis can he as little as 40% of thecapacity at 25°C. This means a lowerdischarge voltage for the same dischargecurrent and therefore that it is easierto drain the battery completely flatunder low temperature conditions.Secondly, the low temperature increasesthe viscosity of the engine oil, makingit more difficult for the starter motorto turn the crankshaft. Thus, in winter,the starter motor draws more currentfrom the battery than in summer. Afurther obstacle is that the relativelylow temperature of the air -fuel mixturefrom the carburetter means that it is

more difficult to ignite.When the ignition key is turned to startthe car, the primary winding of theignition coil is connected via the contactbreaker to the battery. However, thelower discharge voltage of the batterywhen operating at low temperaturesmeans that the current flowing throughthe primary winding will likewise besmaller than normal. Thus, when thecontact breaker opens (at the end ofthe compression stroke), a smallervoltage is induced in the secondarywinding of the ignition coil. The resultis that a smaller voltage is applied to thespark plugs, which are being asked toignite a colder -than -normal mixture.Hardly surprising therefore that oneencounters problems when starting thecar under cold conditions.

The circuitIt should he clear from the aboveexplanation that one of the ways ofimproving the starting performance of

Thanks to present-day low pricesfor nickel -cadmium batteries, itis now an economic propositionto construct a car start booster- an accessory which can proveextremely handy on cold wintermornings.

Figure 1. A relay, two Ni-Cad cells and tworesistors are required for the 'car startbooster'.

a car would be to temporarily increasethe ignition voltage applied to the sparkplugs. This is the basic principle of thecar start booster, the circuit diagram ofwhich is shown in figure I.SI represents the ignition switch; dunnga normal 'cold start' when the ignitionkey is turned contact II is first connec-ted to the battery, so that with thecontact breaker closed, current owsthrough the primary winding of theignition coil. Turning the ignition keyfurther to the right energises the relay.thereby supplying current so the startermotor which turns the flywheel. At theend of the compression stroke thecontact breaker opens and the voltageinduced in the secondary winding of theignition coil causes the spark plug tofire. In the modified 'assisted -start'version of the circuit a second relaysituated between the ignition switchand ignition coil is energised at the sametime as the starter relay. This secondrelay is used to temporarily switch twoNi-Cad batteries in series with the carbattery, thereby increasing the voltagedropped across the primary winding ofthe ignition coil. The result is obviouslya higher current through the primaryof the ignition coil, more energy Is

stored in the resulting magnetic field.and that field collapses a greater volt-age is induced in the secondary.voltage induced in the secondary.Once the engine starts, the ignition keyis turned back to the left I normally it isspring -assisted), the relay drops out andnormal battery voltage appears acrossthe ignition coil. The Ni-Cad cells arethen recharged via RI and R2. The valueof these two resistors depends upon themaximum permissible charge current.Sintered -electrode Ni-Cad cells shouldhe used. since for short periods thedischarge current reaches around 4 A.

Figure 2. This sketch illustrates how thisvarious components are added to the existingcar electrical system.

Constructionthe relay, the Ni-Cad batteries and theresistors are best fitted near the onginal

car start booster alekior september 1978 - 931

9960 I

lee test

2111C/Cd

t2e/ tee

system. The various interconnectionsare illustrated in the drawing of figure 2.At this stage a few constructional tipswould not go amiss. First of all, the leadinto which the two relay contactsshould be inserted can easily be foundworking back from the ignition coil.The coil has three external connections;the middle connection, which isprotected by a rubber cap, goes to thedistributor, and since it carries ex-tremely high voltages, should be leftwell alone.Of the two other leads one goes via thecontact breaker to the chassis, and theother to the ignition switch in thedashboard. The latter lead is the onethat is wanted. The cable should be cut.and each of the ends connected toone of the changeover contacts of therelay. The normally -closed contacts ofthe relay should he joined together.The Ni-Cad batteries are then connectedbetween the two (normally -open)remaining contacts (taking care toensure they are the right way round!).

A tap can then he taken from the starterrelay lead to the new relay.Thus the coil of the relay is connectedbetween the tap and ground. Mountingthe resistors is no problem. R2 forexample, can be mounted in the fuse -box between the accessory fuse and aconnection to the anode of the Ni-Cadbatteries. R I can simply be mountednext to the relay and connectedbetween the appropriate contact of S2hand the earth connection of the relaycoil.If desired, the double -pole changeoverrelay can be replaced by a normalmanually -operated DPDT switch whichis mounted on the dashboard andswitched on when starting the engine.Although this starter aid should satis-factorily resolve some problems causedby cold weather, it should not beregarded as an excuse to forget aboutthe battery capacity altogether! That isto say, it will be of little help if, forexample, one leaves the lights on allnight.

2

C

h

d

e

It

i.

I.

Ir

-

d

ss

le

t.

Id

IC

nw I

'IC

.4

9-32 - elektor september 1978 epplikator

APPLIKATLIcomplex sound generator

The 'complex sound generator' recently intro-duced by Texas Instruments is intended forgenerating sound effects. This 28 -pin IC, theSN76477N, is one of the first linear 12L chips.A wide range of applications is suggested byTI: alarm indicators, timers, sound effects intoys, TV -games, etc.The internal block diagram of the SN76477Nis shown in figure 2, together with someexternal components. The 'complex sound'produced by the chip is determined by twoanalogue voltages (pitch and external VCOcontrol), eight logic levels and a handful offixed resistors and capacitors. Three basic signets are generated by a 'Super Low FrequencyOscillator' ISLFI, a Voltage Controlled Oscil-lator IVCO) and a noise generator.Supply voltage regulation is also incorporatedin the IC, although this is not always re-quired: the user has the option of applyingeither a stabilised 5 V supply to pin 15(Urn) or an unstabilised 7.5 to 9 V supply topin T4 (Ucc). In the latter case, the internalregulator will not only power the chip itselfbut can also deliver up to 10 mA (at 5 VIfrom pin 15.Logic levels are TTL- and CMOS-compatible,logic '1' being defined as more than 2 V(nominally 5 VI and logic '0' being zero volts.The various sections of the block diagram willnow be described in greater detail.

SLF (Super Low Frequency) oscillatorThis oscillator is intended for use in the verylow frequency range 10.1 to 30 HZ), but itwill operate at frequencies up to 20 kHz. Thismay prove useful in some specific appli-cations. The generator produces two outputsignals: a square -wave with a 50% duty -cycle(fed to the mixer) and a triangle -wave whichcan be used to sweep the VCO.The frequency (fs) of the SLF is determinedby the external components Rs and Cs:

0.64fs (approx.), where

Rs x Cs

f is in Hz, R in MS1 and C in u F.

VCO (Voltage Controlled Oscillator)The output frequency of the VCO can bevaried by means of a control voltage. Thiscontrol voltage can be derived from the SLFor, alternatively, an external 'Pitch Controlvoltage' (up) cars be applied to pin 16. Thelogic level at the 'VCO select' input (pin 221determines which one of these control voltages is applied to the VCO. Logic 0 at theVCO select input enables the Pitch Controlinput; a logic 1 causes the output of theSLF oscillator to be passed to the VCO con-trol input. When the Pitch Control input isenabled, increasing the control voltage upcauses the frequency of the VCO to decrease.In some cases it may prove useful to have twoexternal Pitch Control inputs. The secondinput may then be obtained by omitting Csand using pin 21 for Pitch Control - the SLFthen merely serves as an input buffer.The frequency of the VCO can be varied overa 1 : 10 range by means of the Pitch Controlvoltage(s). The low end of the range is set bythe values of Rv and Cv, as follows:

0fvmin

Rv6x4 Cv(approx.), when

f is in Hz, R in kit and C in siF.The 'External VCO Control voltage' (uv),applied to pin 19. determines the duty -Cycleof the square -wave output of the VCO. This,in turn, varies the harmonic content Ithe'sound'), producing an effect somewhat similar to that of a voltage controlled filter. Theduty -cycle can be calculated from

Upduty -cycle = 50-% (appme).

U,,

A 50% duty-cyle can therefore be obtained bysimply connecting pin 19 to pin 16. providedthe pitch control input is enabled Ipin 22 atlogic 0). However, a 50% duty -cycle can alsobe obtained by holding pin 19 at logic 1

(+5 V), even when the VCO is controlled bythe SLF.

Noise generatorA clock generator, 'Noise Clock', drives thenoise generator. The external resistor, Rc, setsan internal current level; the value of thisresistor should be approximately 39 to 47 k.The noise generator proper is a pseudo-random white noise generator (see ElektorE 21, January 19771. Its noise output is satis-factory for most audio applications, but insome cases a signal containing more low -frequency components may be desired. Thiscan be achieved by applying a TTL-compatible squarewave of a suitable frequency tothe 'External Noise Clock' input (pin 3).The output from the noise generator is fed toa low-pass filter Filter'). The cut-offfrequency If _3 dB) of this filter is deter-mined by the values of Rn and Cn

0-3 dB (approx.), where

RI x C n

f is in Hz, R in and C in µF.

MixerOne or more of the signals from the SLFoscillator, VCO and Noise generator areselected and mixed in the mixer stage. Thechoice of signals is determined by the logiclevels at the 'mixer select' inputs (pins 25, 26,27), as shown in table 1. Note that, althoughit would appear logical to have each of thethree 'mixer select' inputs correspond to oneof the three possible signals. this is not in factthe case.The output from the mixer is fed to an'Envelope Generator/Modulator', which willbe described furtheron.

System enable and one-shotThe audio output from the IC is only enabledwhen a logic 0 is applied to the 'SystemEnable' input 1pin 91. No audio output is

produced if a logic 1 is applied to this pin.The System Enable logic also triggers a monostable multivibrator (one-shotl on the nega-tive -going edge of the system enable signal atpin 9. The latter signal should remain at logic0 for the full duration of the one-shot period.The one-shot is intended mainly for obtainingmomentary sounds, such as gunshots, bells,etc. The maximum length of the one-shotperiod (TI is 10 seconds, and is determined bythe values of RI and Ct as follows

T = 0.8 x fit x Ct (approx 1, where

T is in seconds, R in Mil and C in µF.

It is also possible to connect an external one-shot (or timer), for instance if a longer periodis required. In this case Rt and C/ are omitted.Initially pin 23 is held at logic 0 and the pulseis started by applying a negative -going edge topM 9; the pulse is terminated by taking pin 23to logic 1.

EnvelopeThe output signal from the mixer is fed to the'Envelope Generator/Modulator'. Basically, ofcourse, envelope shaping is equivalent toamplitude modulation, and the modulatingsignal in this case is determined by the logiclevels at the two 'Envelope Select' inputs(pins 1 and 28) as shown in table 2.The wave -shapes shown in table 2 are given asexamples to illustrate the various types ofenvelope shaping. In these examples, themixer output is shown as a pseudo -randombinary noise signal as produced by the noisegenerator.The envelope shown for the one-shot requiresSome further explanation. In general, theaudio signal will not be turned on and offinstantaneously: the gradual rise to full out-put level ('attack') and gradual level reductionat the end of the pulse ('decay') are bothdetermined in part by the value of the capaci-tor (Cal connected to pin 8 In combinationwith this capacitor, Ra (pin 10) sets theattack time Ta and Rd (pin 71 the decay time:

Ta Ra x Ca (approx.) andTd Rd x Ce (approx.), where

T is in seconds, R in nom and C in µF.

Output amplifierThe gain of the output amplifier A is set bythe values of resistors Rf and Rg. Since thesignal levet applied to this amplifier is conslant, it is more practical to specify the peakaudio output level (uciaseats ) as a function ofthese resistors:

Rfuo,peak 3.4 -R laPProx.I, where

9

u is in volts and R in kaTo avoid clipping, the peak output levelshould be less than 1.2 V, which means thatRg should be at least three times the value ofRf.

Final noteIf some sections of the IC are not used in aparticular application, the correspondingexternal components may be omitted. Forinstance, if the noise generator is not required,Rc, Rn and Cn may be omitted and pins 4, 5and 6 are left floating.Table 3 lists the most important operatingcharacteristics. A few typical applications areshown in figure 2.

Texas Instruments preliminary data sheet.

Figure 1. Internet block diagram of the 'com-plex sound generator', type SN76477N. Theexternal resistors and capacitors shown willnot be required for all applications.

Figure 2. Typical applications of theSN76477N. The sound effects generated arethose for a train or propellor aircraft (2a),a gunshot or explosion 126) and various sirenand science fiction effects I2c).

Under the heading Applikator, recently introduced components and novel applications are described. The data and circuits given are based oninformation received from the manufacturer and,or distributors concerned. .Vormally, they will nor have been checked, built or tested by Elektor.

appirkator elektor septembar 1978 - 9.33

3

r

9

II

U

1,

C (pin 271 13 ,pin 261 A (pin 251 output

0 0 0 VCO0 0 1 noise0 1 0 SLF0 1 1 VCO/noise1 0 0 SLF/noiseI 0 1 SLF/VCO1 1 0 SLF/VCO/noise1 1 1 -

Table 2

mixer 111111 NIVCO F\I-11-1

one shot / \envelopeselect

Atynn B I 1,e, 28

modulation envelope modulator output

4/ VCO/AM IGH1111041),

o 1

nn:odutation 11 HEM1 O one slat -4.1 711./P-

I 1 VCO/product 11H110141

Table 3

Absolute maximum ratings

supply voltage U," (pin 151 6 Vsupply voltage Ucc (pan 14) 12 Vinput voltage at any other pin 6 V

Recommended operating conditions

min typ maxsupply voltage Urn (pin 151 4.5 5.0 5.5 Vsupply voltage Ucc (pin 141 7.5 9.0 V

Aerating temperature 0 25 70 C

Operating characteristics lUrag ' 5 V; Tamb - 25'C1

min typ max

supply current Icc (no ext. load) 15 40 mAUrog lUcc .. 8.25 V, (load 10 mA1 4.5 5.5 Vinput regulation for Ureg 150 mycurrant through resistorslogic '1' input current,

system enable inputother inputs

1

40

200

10052

µA

,u0,

µAlogic '1' input voltage 2.0 Vlogic 1;1' input voltage 0.8 VVCO cut-off (pin 161maximum output voltage

swing. peak -to -peak

2.5

2.5

V

Voutput impedance 100 II

&MAI= 11Nlr

Under the heading Applikator, recently introduced components and novel applications are described. The data and circuits given are based onotfonnarion received from the manufacturer and oor distributors concerned. Normally, they will not have been checked, built or tested by Elektor.

9-34 - elektor september 1978 24 dB-VCIF

In response to requests fromreaders who have built theFormant synthesiser the followingarticle presents a design for avoltage -controlled filter whoseslope is considerably steeper thanthat of the original VCF, in fact24 dB/octave as opposed to12 dB/octave. The filter offers achoice of highpass or lowpassmodes and slopes of 6, 12, 18 or24 dB/octave.

C. Chapman

VCFs with an extremely steep slopeseem to have a particular appeal formost synthesiser enthusiasts because ofthe greater range of tonal possibilitiesthat they offer. Formant users arcevidently no exception to this rulejudging by the number of requests for a24 dB/octave VCF. Of course, the filterdescribed here is not restricted to usewith the Formant synthesiser, but mayalso be used with other synthesiserdesigns.

New possibilitiesIt should be stated at the outset that the24 dB VCF does not render the existing12 dB design obsolete. On the contrary,the two filters are complementary toone another and can be used in combi-nation to provide greatly increasedpossibilities for tailoring the harmonicstructure of the sounds produced byFormant.For example, the 12 dB VCF can beused in the bandpass mode togetherwith the steep filtering of the 24 dBVCF to produce selective tone color-ation. The two filters can be controlledby the same envelope shaper or bydifferent envelope shapers, and may heconnected in cascade or in parallel.The latter arrangement offers severalinteresting possibilities. For example,hard, metallic sounds can he producedby applying a short, steep envelope volt-age to the 12 dB VCF and a longer,shallower contour to the 24 dB VCF.If the filter inputs are connected inparallel then interesting effects may beobtained by connecting one VCF out-put to one input of a stereo amplifier

and the other VCF output to the otherinput. This gives rise to a very distinc-tive dynamic amplitude characteristicand stereo imaging, particularly if thetwo VCFs are controlled by differentenvelope shapers.The audible differences between the12 dB VCF and the 24 dB VCF arequite prominent. The 12 dB VCFproduces sounds that are distinctly'electronic', which can have a slightlyfatiguing effect on the listener duringextended playing sessions. The soundsproduced by the 24 dB VCF, on theother hand, are much more 'natural',and can be listened to for extendedperiods without fatigue. This effect is

probably due to the more severe filteringof higher harmonics which the 24 dBVCF provides when used in the lowpassmode, since these harmonics tend tomake the sound of the 12 dB VCFmuch more shrill than that of the 24 dBVCF.The effect of the steeper filter slope ofthe 24 dB VCF is illustrated in figure 1.which shows the different outputs fromthe 12 dB VCF (dotted line) and 24 dBVCF (continuous line) when fed with asawtooth waveform. It is apparent that.due to the almost complete removal ofthe harmonics of the sawtooth, the out-put of the 24 dB VCF is practically asinewave, whereas the original waveformis still apparent at the output of the12 dB VCF since the harmonics are onlypartially removed.It is clear from the foregoing that a

24 dB VCF greatly extends the musicalpossibilities of a synthesiser and isvirtually a must for the serious user.

24 dB-VCF elektor september 1978 - 9.35

U.

12 dB Filter

/ s 24 dBFilter

99611

!ABC

0

WSJ-)

Design of the 24 dB VCFMost 24 dB VCF, are variations on theheavily -patented design by R.A. Moog,which has been around for a number ofyears. However, thanks to the advent ofinexpensive IC OTAs (OperationalTransconductance Amplifiers) a moreversatile design than Moog's is nowpossible, which can be operated inhighpass or lowpass modes with slopesof 6, 12, 18 or 24 dB/octave. Evengreater slopes than 24 dB would be poss-ible, but experiments have shown that agreater slope does not result in a corre-sponding increase in tonal quality.The design of the basic filter sectionshown in figure 2 is very similar to thatof the 12 dB VCF, which was describedin detail on page 12.29 of Elektor 31,December 1977. However, advantagehas been taken of recent developmentsin FET op -amp technology to simplifythe design. As described in that articlethe basic filter section is an integrator or6 dB/octave lowpass section consistingof an OTA driving a capacitor. The volt-age/current transconductance (gm) ofthe OTA can be varied by an externalcontrol current and hence, via an expo-nential voltage/current converter, froman external control voltage. This controlcurrent alters the time constant of theintegrator and hence the turnover fre-quency of the filter section.

The output current of the OTA must allflow into the capacitor, otherwise the21tegrator characteristic will be less thanzeal. This means that the output of theOTA must be buffered by an amplifierwith a high input impedance. In the

Figure 1. This illustrates the difference be-tween the outputs of 12 dB/octave VCF anda 24 dB/octave VCF having the same turnoverfrequency, when fed with sawtooth input.The 24 dB VCF removes practically all theharmonics giving a unaware output, whereasthe original weveshape is still distinguishableat the output of the 12 dB VCF.

Figure 2. The basic filter section of the 20 dBVCF n the same as that of the 12dB VCF, 1.o.en OTA integrator followed by a FET op -ampbuffer.

Figure 3. The hghpass function is obtainedby connecting the 6 dB lowpass section in thefeedback loop of an operational amplifier.

Figure 4. To obtain a 24 dBioctava filter, four6 dB/octave sections are cascaded.

4

00- FL1

- -r

LP

HP

FL2

LP,

-oHP

-r

LP : LP

F L3

HP HP

I2d8 18dB

))3 ,6d8 24 dB

9-36 - elektor september 1978 24 d8-VCF

5 Figure 5. Positive feedback around the entirefitter allows the response to be boosted aboutthe turnover frequency. The degree of boostcan be varied by '0' control.

Figure 6. Block diagram of the 24 dB/octevefilter, showing how the 0 control is incorpor-ated.

Figure 7. Complete circuit of the 24 dB VCF.The exponential voltage/current converter is

identical to that used in the 12 dB VCF.

12 dB VCF this was achieved by using adiscrete FET source follower and a 741op -amp. Fortunately, relatively inex-pensive quad FET op -amps such as theTexas TL084 are now available. The useof one of these ICs greatly simplifies thedesign and obviates the need to selectFETs, which is rather a chore when oneconsiders that the 24 dB VCF uses fourintegrator stages.

Highpass functionThe highpass mode of the filter is

achieved by connecting the 6 dB/octavelowpass section in the negative feedbackloop of an operational amplifier, Al, asshown in figure 3. A highpass filter re-sponse is then available at the output ofAl whilst a lowpass response is simul-taneously available at the output of A3.Of course, this arrangement gives only a6 dB/octave slope per section, and inorder to obtain a 24 dB/octave filterfour filter sections, built according tothe circuit of figure 3, must be cascadedas shown in figure 4. Switching at theoutput of each section allows selectionof highpass or lowpass mode, whilst a4 -position switch allows I, 2, 3, or4 filter sections to be switched in togive 6-, 12-, 18-, or 24 dB/octave slopes

respectively.It is apparent that this arrangement isdifferent from the two -integrator loopor state -variable filter which formed thebasis of the 12 dB/octave filter. In the12 dB/octave filter, lowpass, highpass,bandpass and notch modes were avail-able simultaneously at various points inthe circuit, though in fact only onefunction at a time could be selected atthe output.An interesting effect, shown in figure 5,can he obtained with the 24 dB VCF ifa feedback loop is connected from theoutput of the cascaded filters to thenon -inverting input of the first stage asillustrated in figure 6. Due to the phaseshift around the turnover frequency thiscauses positive feedback, which booststhe gain of the filter around theturnover frequency as shown in figure 5.The degree of boost is adjustable bymeans of a 'Q' control. The choice ofRx is important as too much feedbackwould cause the circuit to oscillate, sothe value of Rx is a compromisebetween stability and a reasonabledegree of boost.

Complete circuitThe complete circuit of the 24 dB VCF

is given in figure 7. The exponentialconverter, constructed around TI. T2and ICI, is identical to that used in the12 dB VCF and gives the same 1 octaveper volt characteristic to the turnoverfrequency of the filter. The control volt-age inputs are also the same as for the12 dB VCF, and are listed in table I.Since the 24 dB VCF must have theoption of being connected in parallel orin cascade with the 12 dB VCF, theinput switching arrangements are a

little complicated. A9 and A10 form anon -inverting summing amplifier for thethree VCO inputs. whilst the output ofthe 12 dB VCF is fed in via the IS con-nection. With S4 in position 2 the out-put of A10 is disconnected, so the VCOinputs are inhibited. The output of the12 dB VCF is fed to the input of the24 dB VCF via S4 and R51, so that thetwo VCFs are in cascade.With S4 in position 1 the output of A 10is connected to the inputs of the 24 dBVCF, whilst the output of -the 12 dBVCF is routed through A 11. The outputof Al 1 and the output of the 24 dBVCF are added together in the outputsumming amplifier Al2, i.e. the twoVCFs are connected in parallel.The four 6 dB/octave filter sections

24 dBNCF elak tor september 1979 - 9-37

76

ez =1

V 91. 911. c= c:L® cr. 100.1

IsV7- 116

1113*

ao

ww43.1 film

NO 0.111

7b

V C>

w

Al A2 A3 A4 IC2 TL 084AS + A7 A8 . ICS TL 084A9 + A10 + All + Al2- 1(311. IL 094

an

fa

ICI mA 741 MiNdipIC3 - CA 3090.IC4 CA 3080*I C,8 - CA 3080*IC7 CA 3080*

b.?

111638o

re tan

9-38 - elisktor saptember 1978 24 dB-VCF

13a Bb bottom view

bottom view

9913

CA3080 A

top view

- INPUTOUTPUT

INPUT

O

BFX 11BF X 362N3808 2N38011AD 820 822

ne63-112

0OUTPUT

'ABC

99519,

comprise Al to A8 and 1C3 to 1C7. Thefour poles of switch S2 select betweenhighpass and lowpass modes, while S3selects the filter output and hence theslope. The reason that 53 is a two -poleswitch may not be immediately appar-ent, hut is easily explained. Ignoring thephase shift introduced by the action ofthe filter, i.e. considering only signals inthe filter passband, each filter sectioninverts the signal fed to it, since Al, A3,AS and A7 are connected as invertingamplifiers. This means that the outputsof alternate filter sections are either inphase or inverted with respect to theinput signal. To ensure that the filteroutput is in the same phase relationshipto the input signal whatever filter slopeis selected, Sib is arranged to switchAl 2 between the inverting and non -inverting modes to cancel the inversionsproduced by the filter sections.Like the 12 dB VCF, the 24 dB VCFhas two outputs, a hardwire output con-nection 105 and an uncommitted out-put, EOS, which is connected to a frontpanel socket.

ConstructionAs far as the choice of components forthe 24 dB VCF goes, the same generalcomments apply that were made aboutthe 12 dB VCF and the Formantsynthesiser in general. All componentsshould be of the highest quality; re-sistors should he 51, carbon film typesexcept where metal oxide or metal filmtypes are specified; capacitors shouldpreferably he polyester, polystyrene orpolycarbonate, and must be these typeswhere specified. Semiconductors shouldbe from a reputable manufacturer.As with the 12 dB VCF the dual transis-tor may be any of the types specified in

Figure 8. Pinouts for the dual transistors andCA3080.

Figure 9. Printed circuit board and componentlayout for the 24 dB VCF. (EPS 9953-11

Table 1. Summary of the control functionsand input/output connections of the 24 dBVCF.

Table 1

a) hardwired inputs (not on the front panel)KOV - Keyboard Output Voltage

(from interface receiver)ENV - Envelope shaper Control

Voltage (from ADSR unit)VCO 1.2,3 Signals from VCOs 1, 2,3IS Internal signal from the 12 dB

VCF

bl external inputs (sockets on front parboilECV = External Control Voltage (for

exponential generator of theVCF)

TM = Tone Colour Modulationinput

ES = External Signal (from e.g.noise module)

c) outputsIOS - Internal Output Signal (from

VCF to VCAIEOS - External Output Signal

(socket on front panel)

dl controlsTM - P3; sets tone colour

modulation levelES - P5; sets external signal levelENV P2; sets envelope shaper

control voltageOCTAVES P1; coarse frequency

adjustmentO - P4; sets level of peak boost

around turnover frequencyOUT P6; sets lOS output level

el switchesECV/KOV Si; selects external or internal

control voltage input

Parts list to figures 8 and 10

Resistors:

R1 100 k metal oxideR2,R4 - 100 kR3 47 kR5 - 33 kR6 1k8R7,R9 - 330 kR8 = 2k2R10,R37,R39,R41,R43 = 12 kR11 .. F116,R19 . R22,R25 .. R213.R31 R34,R45,R46,R47,R50,R51.R52,R55,R56 = 39 kRI 7,R18.1323,R24.R29,R30.R35,R36 = 100 SZR38,R40,R42.R44 - 27 kR48 = 470 UR49 - 100 k (see text)R53,R54 - 10 kR57 - 82 k

Potentiometer:P1 P4 = 100 k linearP2,P3 = 47 k (50 k) linearP5 = 47 k 150 k) logarithmicP6 4k7 15 k I logarithmicP7 = 100 k presetP8 = 470 St (500 511 preset

Capacitors:Cl ,C8,C9 - 680 nC2 -1nC3 = 680 p (polystyrene, not

ceramic)C4,C5,C6,C7 150 p

(polystyrene, not ceramic)CIO ...C18 - 100 n

SemiconductorsIC1 = 741IC2,1C5 - TL 0134, TL 074IC8 = TL 084, TL 074, LM 3241C3 ..106 - CA 3080,

CA3080A IMINIOIP or TO;see text)

T1,T2 A0820 ...822,2N3808 ... 3811,8FX 11,BFX 36 (see text) or2 x BC 5578

Miscellaneous:31 -pin DIN 41617 connector or

terminal pinsSI SPOT$2 i= 4 -pole double throwS3 - 2 -pole 4 -way, index angle

approx. 30'$4 DPOT4 minature sockets, 3.5 mm dia.7 13 ... 15 mm collet knobswith pointer (to match existingsynthesiser modules).

24 dB-VCF eloktor september 1979 - 9-39

the parts list, or may be home-made bygluing together two normal transistors,though in this case thermal tracking willnot be quite so good. The CA3080should preferably he in a MINIDIP pack-age to fit the hole spacings on the p.c.board, though the metal can type can bemade to fit by splaying the leads. Thepinouts for the dual transistors and theCA3080 are given in figure 8.

Although not absolutely necessary, it is

a good idea to select OTA's with ap-proximately the same transconductance,

since the four sections of the filter willthen have almost the same turnover fre-quency. The CA3080 is available in twoversions, the standard version, in whichthe ratio between the maximum andminimum gm is 2:1, and the CA3080A,in which the spread in gm is only 1.6:1.A test circuit and test procedure forselecting ICs with similar gm are given atthe end of the article, and it is certainlyworthwhile buying a few extra OTAsand selecting the four with the mostsimilar gm. The 'reject' devices are per-

fectly acceptable for use in the 12 dBVCF or VCA, and need not be wasted.The other ICs in the circuit should allbe T1074 or TL084 quad BIFET op -amps, although for IC8 it is only permiss-ible to use an LM324. Thanks to the useof quad op -amps it is possible to accomo-date the 24 dB VCF on a standardEurocard-size (160 mm x 100 mm) p.c.hoard, although the control connectionsare not all on the front edge of theboard. The printed circuit pattern andcomponent layout for this board are

9

aP 6 0 Alm -y-0 °EOST

53b0.453 roe

84101 IIa bQ4054.1

9-40 - elxktor september 1978

given in figure 9, while a front panellayout is given in figure 10.

Test and adjustment

To enable the exponential converter andthe filter section to be tested separatelythey are joined by a wire link whichruns across the hoard from T2 to apoint adjacent to RI5. This link shouldbe omitted until the VCF has beentested.To test the filter section it is necessaryto provide a temporary control current.This is done by connecting a 100 k logpotentiometer between --15 V andground, with its wiper linked to thejunction of R39 and R4 via a multimeterset to the 100µA DC range. The testthen proceeds as follows:I. Turn the wiper of P4 fully towards

ground, select 24 dB slope with S3and adjust the control current to100 pA.

2. Feed a sinewave signal into the ESsocket and adjust either the sinewaveamplitude or P5 for 2.5 V peak -to -peak measured on a oscilloscope atthe wiper of P5.

3. Monitor the filter output on the'scope and check the operation ofthe filter by varying the sinewavefrequency and checking that thesignal is attenuated above the turn-over frequency in the lowpass modeand below the turnover frequency inthe highpass mode.

4. The function of S3 should now bechecked. Set S3 to the 6 dB positionand S2 to the LP position. Increasethe frequency of the input signaluntil the output of the filter is 6 dBdown on (i.e. 50% of) what it was inthe passband where the response waslevel. Now switch to 12 dB, 18 dBand 24 dB and check that the re-sponse is respectively 12, 18 and24 dB down, i.e. is reduced to 25%,12.5% and 6.25% of its original value.The exact results of this test willdepend upon the matching of theOTAs.

5. Set the Q control, P4, to its maxi-mum value, when the circuit shouldshow no sign of oscillation. If thecircuit does oscillate it will be necess-ary to increase the value R49. If itdoes not oscillate then the Q rangecan be increased by decreasing R49,taking care that instability does notoccur.

6. Finally, the linearity of the turnoverfrequency v. control current charac-teristic should be checked. Adjustthe input frequency until the re-sponse is a convenient number of dBdown (say 6 dB). Double the controlcurrent then double the input fre-quency and the response should stillbe 6 dB down.

7. To check the exponential converterconnect a 27 k resistor in series witha multimeter set to the 100 pA rangebetween the collector of T2 and the-15 V rail. Then follow the test

1024 1111-VCF I

3

24 418-VCF etektor teptornber 1978 - 9-41

/1

12

b el atsbol a /4ispHP/LP

(521 2a 2b 2c 20

del gi

VC0s/ES(541

OUT 0)a

IP41

ES())

ES(P5)

OUT(P61

OCTAVES it(P11

ENV(P21

TM r\Li -1 Thii-

(1.3)

ECV L)ECV/KOV

(511

9963 i2

Figure 10. Front panel layout for the VCF.(EPS 9953-21.

Figure 11. Showing the wiring between thep.c. board and the front -panel mounted cornponents.

Figure 12. The 24 dB VCF is connected intothe Formant system between the 12 dB VCFand the VCA.

9963,13

9-42 - elektor soptember 1978 24 dBVCF

Flour, 13. Test elreutt for the s4 Ilion ofOTAs.

procedure given on page 12-33 ofElektor 32, column 1 line 51. Theoffset and octaves per volt adjust-ments can also be carried out usingthe procedure given in this issue.During the offset adjustment P4should he set to minimum and S3should be set to the 24 dB position.During the octaves/volt adjustmentof PS the Q control, P4, should beset to maximum, as with the 12 dBVCF.

Using the 24 dB VCFBefore the 24 dB VCF can be put towork it must first he connected into theFormant system. Fortunately, as far asthe signal paths go this involves changingonly two connections and adding threemore. Ac can he seen from figure 12,the 24 dB VCF is connected betweenthe 12 dB VCF and the VCA, so thatthe IOS output of the 12 dB VCF nowgoes to the IS input of the 24 dB VCFinstead of to the VCA, whilst the VCAreceives its input from the IOS outputof the 24 dB VCF. The 24 dB VCF alsohas inputs from the three VCOs.In addition to the signal connections the24 dB VCF must also be provided withsupply to the VCF module in accordancewith the standard practice for Formant.Provision of control voltage inputs fromthe ADSR envelope shapers will hediscussed later.For satisfactory operation of the 24 dBVCF the correct setting of the inputlevel is important, even more than in thecase of the 12 dB VCF. On the onehand, the input level should not be solarge that distortion occurs, but on theother hand it should not be so smallthat the signal-to-noise ratio is degraded.The 24 dB VCF is designed so that theoptimum input level is obtained usingthree VCOs set to maximum output,

with one waveform selected per VCO. ifmore than three VCOs are in use ormore than one output waveform isselected from each VCO then the VCOoutput levels must be reduced. On theother hand, if only one VCO is usedthen the signal level may be too low. Inthis case it is best to patch the EOSsocket of the VCO to the ES input ofthe VCF, since this input has approxi-mately three times the sensitivity of thehardwired VCO inputs.The 24 dB VCF is capable of the samebasic functions as the 12 dB VCF;driven by the KOV control voltage itwill operate as a tracking filter, whilstthe ENV and TM inputs allow dynamicmodulation of the harmonic content ofthe VCF output. Due to the greaterslope of the 24 dB VCF the setting ofthe ENV level control is more criticalthan with the 12 dB VCF, but if cor-rectly adjusted then subtle nuances inthe tonal character of the output signalare possible.The question arises as to which ADSRenvelope shaper should be used tocontrol the 24 dB VCF, since only twoare built into the basic Formant system,and control the VCA and 12 dB VCFrespectively. Because of the modularconstruction of Formant it is, of course,perfectly feasible to build a thirdenvelope shaper, which is the mostversatile arrangement. The alternativesare to patch one of the other ADSRoutputs to the TM input of the 24 dBVCF, or to hardwire the ENV input ofthe 24 dB VCF to the output of theenvelope shaper that controls the 12 dBVCF. This latter arrangement is prob-ably preferable, as it allows the ADSRsignal to he fed to one or both VCFs bysuitable adjustment of their ENVcontrols and also allows the possibilityof patching the output of the otherenvelope shaper into the TM input ofeither VCF.

Appendix

OTA selection procedureAlthough not absolutely essential, it iswell worth selecting OTAs with closelymatched transconductance character-istics to ensure that the four filtersections track accurately.A test circuit for the OTAs is given infigure 13. This should be fed with asinewave signal of about 2 V peak -to -peak (or 0.7 V measured on an ACvoltmeter) from a signal generator orfrom one of the VCOs. The outputshould be monitored on a 'scope or ACvoltmeter. With a control current of10014A. measured on the multimrtrrin series with R5, the output voltageshould be between 0.7 V and 1.3 Vpeak -to -peak. Without changing theinput level or control current the OTAsto be tested should he plugged into thecircuit one at a time and the outputlevel for each OTA noted. The fourOTAs whose output levels are mostsimilar should he used in the VCF.The circuit can also be used to checkthe linearity of the transconductance v.control current characteristic of theOTAs, e.g. doubling the control currentshould double the output of the testcircuit and halving the control currentshould halve the output.

butferod/unbufiemd CMOS elektor september 1978 - 9-43

buffered/unbufferedCMOS

The two most popular logic families areTTL (Transistor -Transistor Logic) andCMOS (Complementary Metal -OxideSemiconductor). TTL is based onbipolar transistors, i.e. transistors whoseoperation depends upon two types ofcharge carriers, electrons and holes (e.g.normal NPN and PNP transistors).Other logic families based on bipolartechnology include DTL (Diode -Transis-tor Logic), RTL (Resistor -TransistorLogic), HLL (High -Level Logic), ECL(Emitter -Coupled Logic), DCTL (Direct -Coupled Transistor Logic) and Ill_(Integrated Injection Logic). With theexception of ECL for high-speed circuits,these other logic families are rarely usedby the home constructor, although 13Lis a developing new technology which isbeginning to offer serious competitionboth to TTL and CMOS.CMOS logic ICs, on the other hand, arebased on FET (Field Effect Transistor)technology, sometimes known as uni-polar devices because their operationdepends upon only one type of chargecarrier. The FETs used in CMOS are ofthe insulated -gate or MOSFET type, socalled because the gate connection isinsulated from the silicon substrate ofthe device by a layer of silicon dioxide.This means that MOSFETs have anextremely high input resistance, whichis something of a mixed blessing, as willbe seen later.

TTL v. CMOSAlthough a host of logic devices areavailable in both TTL and CMOS,ranging from the simple to the verycomplex, a look at the internal circuitof one device from both families willserve to illustrate the differences be-tween them. Figure la shows the circuitof a TTL two -input NOR gate, whilstfigure 1 b shows the internal circuit of aCMOS two -input NOR -gate (unbufferedtype).The TTL NOR gate operates in thefollowing manner: if eit'er one or bothof the inputs to the gate is high (orfloating) then base current will flowinto T2 or T3 (or both) through the 4kresistors and the forward -biased base -collector junction of TI or T4. T2 and/or T3 will thus be turned on and current

Many CMOS digital ICs are nowavailable in two versions, bufferedand unbuffered. The lack ofinformation available to theamateur on the differencesbetween them has given rise tosome confusion as to thecompatibility of the two types.This article examines both typesof CMOS, compares them with theother popular logic family, TTL,and indicates the applications inwhich buffered CMOS is superiorto unbuffered, and vice versa.

will flow through the lk resistor,turning on T6. Due to the forwardvoltage of DI and the saturation voltageof T6 the emitter of T5 is at a higherpotential than the base, so 15 is turnedoff. The output of the gate is thus low,the output voltage being equal to thesaturation voltage of .16.When both inputs to the gate are lowthen both T I and T4 will be turned onby current flowing through the 4kresistors into their bases. The bases ofT2 and T3 will be pulled low by T I andT4 respectively, so these transistors willbe turned off. T5 will thus be turned onby current flowing into its base via theI k6 resistor, while T6 will be turned off.The output of the gate will thus be high,the output voltage (with no load) beingsupply voltage minus the base -emittervoltage of T5 and the forward voltage ofDI.Whereas the TTL gate uses only onetype of bipolar transistor (NPN) theCMOS gate uses complementary pairs ofP -channel and N -channel MOSFETs,hence the term Complementary MetalOxide Semiconductor.The basic configuration of the CMOStwo -input NOR gate shown in figure I bis simply two complementary pairs ofMOSFETs. The diodes and resistors atthe two inputs are protection circuits,which are required only at externalinputs to the gate. In more complexlogic devices where some elements ofthe circuit have no external connectionsthese protection circuits would be

omitted, only being included in sectionsof the circuit connected to the pins ofthe IC. This simple gate configurationallows CMOS IC chips to have a muchgreater packing density than TTLdevices, since the resistors used in TTLcircuits occupy a large proportion of thechip area.Operation of the CMOS NOR gate is

quite easy to understand. When bothinputs are low then the P -channel FETsTI and T2, which are connected inseries, are both turned on, whilst theN -channel FETs T3 and T4 are bothturned off.The output of the gate is therefore high,being connected to positive supply viathe drain -source resistances of TI andT2 Under no-load conditions the high

9-44 - elektor september 1978 buttered/unbuffered CMOS

la

v cc

Ic

output voltage is virtually' equal tosupply voltage (VDD).If either input to the gate is low then T1or T2 (or both) will be turned off,effectively disconnecting the outputfrom positive supply. T3 or T4 will beturned on, so the output will be pulledlow via the drain -source resistance ofone (or both) of these transistors. Underno-load conditions the low outputvoltage will be virtually zero (Vss).The characteristics of CMOS circuitsapproach those of an Ideal' logic familymuch more closely than TTL, for anumber of reasons.I. An ideal logic family should be

capable of operating over a widerange of supply voltages. CMOS canoperate over a wide supply range, butTTL is limited to a small operatingrange around 5 V, due to the use ofresistors and semiconductor junctionvoltage drops to define operatingconditions within the circuit.

2. The transfer characteristics (i.e.output voltage plotted against inputvoltage) of a logic gate shouldapproach that of an ideal electronicswitch, i.e. for input voltages up tohalf supply the output should remainin one state, above half supply theoutput should be in the oppositestate, the transition from one state tothe other being as abrupt as possible.This is illustrated in figure 2, whichcompares an ideal transfer character-istic with those of TTL and CMOS. Itcan be seen that CMOS approachesthe ideal much more closely thanTTL, which has a decidedly asym-metric transfer characteristic.

3. The output of an ideal logic deviceshould be capable of dnving theinputs of a large number of similardevices (fanout capability) withoutthe load causing the output voltageto fall below its permissible high levelor rise above its permissible low level.The low output of a TTL deviceconnected to the input of anotherTTL device must be able to sink thecurrent of around 1.6 mA which

buffered/unbuffered CMOS stook tor september 1978 - 9-45

flows out of the emitter of the inputtransistor. In the high state the base -emitter junction of the input transis-tor is reverse -biased, so a high outputneeds to supply very little current.Most TTL devices, except buffers,can drive 10 other TTL devices(quoted as a fanout of 10).CMOS, on the other hand, has a vir-tually unlimited fanout, at any ratefor low operating speeds. Since theinput resistance of a CMOS gate ispractically infinite it imposes no DCloading on the output of the gatewhich is driving it. As the operatingspeed increases, however, the picturechanges. The input of a CMOS gatehas a capacitance of typically 5 pFdue to the capacitor formed by thegate electrodes, oxide layer andsubstrate. In addition the circuitryexternal to the device will have itsown stray capacitances. The outputof a CMOS gate has a resistance ofseveral hundred ohms which forms alowpass filter with these stray capaci-tances. This limits the maximumoperating frequency so that a trade-off must be made between outputdrive capability and operating speed.Increased load capacitance also in-creases the power dissipation ofCMOS.

4. The power consumption of an ideallogic family should be as low espossible. TTL devices rely on thevarious resistors in the circuit tocharge and discharge the transistorcapacitances. These RC time con-stants determine the maximum oper-ating speed of TTL, so the resistorvalues cannot be too large or speedperformance will suffer. Furthermore,since several of the transistors in aTTL gate are turned on at any time,providing a path to ground via theseresistors, TTL circuits dissipate a

considerable amount of power. Inaddition to the standard (74XXseries) TTL circuits there are othervariants of TTL which reflect thespeed/dissipation trade-off. Low -

Nuns 1a. Internal circuit of a two inputTTL NOR gate.b. Internal circuit of an unbuffered CMOStwo input NOR gate.c. Internal circuit of a buffered CMOS two -Input NOR gate.

Figure 2. Comparison of the transfer characwishes of TTL and unbuffered CMOS.

Figure 3. Power dissipation Iper gate) versusfrequency for standard TTL, Schottky TTL,lowpoww Schottky TTL and CMOS.

Figure 4. Illustrating that the output resist-ance of buffered CMOS remains constant,whilst that of unbuffered CMOS depends onhow many Inputs of a gate are used.

3 wee

soo

so

a

Schott*, 11

CMCNI iVoc. IS VI

PaColl Moo 50 511

4

CMOSIVDD-Svi I_".or

II0' 1o' we 50'

(1.4

sus 3

4 a

w es

b

gear a eeie 4 09.111 4.1

power TTL, for example, uses higherresistance values, giving lower dissi-pation at the expense of speed. High-speed TTL uses lower resistancevalues, increasing operating speed atthe expense of dissipation. SchottkyTTL utilises Schottky barrier transis-tors to obtain high operating speedswith dissipation similar to that ofstandard TTL, while low -powerSchottky TTL uses Schottky transis-tors and higher resistance values toobtain the same speed as standardTTL but with reduced dissipation. Itseems fairly likely that low -powerSchottky will eventually become thestandard TTL logic family.

However, returning to CMOS circuits,it is fairly apparent that the staticdissipation of these devices is vir-tually zero. Since the upper andlower transistors of a complementarypair cannot be turned on simul-taneously there is never a currentpath to ground, and since the inputresistance of a CMOS gate is ex-tremely large no current is taken bythe inputs of other gates which arebeing driven.The situation is different for ACoperation, however. As the output ofa CMOS gate switches between itshigh and low states it passes througha transitional region where bothFETs of a complementary pair areturned on. This causes a current ofseveral milliamps to flow from thesupply rail to ground through thetwo FETs, causing power to bedissipated in them. At low operatingspeeds, provided the input pulses fedto the gate have short risetimes, theaverage power dissipation will be

small since both FETs are on foronly a very small proportion of thetotal time. As the input frequencyincreases, however, the transitionregion will occupy a greater pro-portion of the total cycle and thepower dissipation will increase. Asimilar effect occurs in TTL due toboth output transistors being turnedon simultaneously, but at lowoperating speeds it is the static powerconsumption which is predominant.This is illustrated in figure 3, whichshows power consumption per gateversus frequency for CMOS, standardTTL, Schottky TTL and low -powerSchottky TTL. It can be seen thatthe power consumption of CMOSrises steadily with frequency, whereasthat of TTL stays fairly constant upto about 5 MHz.

5, An ideal logic family should havegood noise immunity. The definitionsof noise immunity are quite compli-cated, hut basically it is the ability ofa logic device to resist false switchingby noise pulses. The so-called DCnoise immunity of CMOS is typically45' of supply voltage, with 20%being guaranteed. TTL, on the otherhand, has a DC noise immunity ofonly 400 mV or so. The one par-

9-46 - elektor september 1978 buffered/unbuffered CMOS

5a

0

10 15

v,n20

9920 5.

b

t

'WO

ab

S

S 0 20VI -an

0938 Sn

ameter in which TTL scores overCMOS is operating speed, the propa-gation delay of a TTL gate being ofthe 10 ns, whilst that of unbufferedCMOS is an order of 10 greater.Buffered CMOS is even slower.

Unbuffered v. buffered CMOSAl.! the foregoing comments apply tounbuffered CMOS devices, and thedifferences between these and buffereddevices will now be considered. Figurelc shows the internal circuit of a

buffered, two -input CMOS NOR gate.In fact, this NOR gate actually consistsof a NAND gate with inverters on its in-puts and output, which (by DeMorgan's)theorem, is logically equivalent to a

NOR function. An alternative approach,adopted by some manufacturers, is to usea NOR gate whose output is buffered bytwo cascaded inverters. The logicdiagrams for both these arrangementsare shown in figure lc. Both are logi-cally equivalent to a single NOR gate.The buffered NOR gate is obvi-ously much more complex than theunbuffered gate, so what advantage doesit offer, and what are its disadvantages?

To begin with, since the output of abuffer gate consists of a single, comp-lementary pair, a buffered gate has aconstant output resistance, equal to thedrain -source resistance of whicheverFET is switched on. This means that therise and fall times of the output signalare fixed for given load conditions,which can be an important factor insome applications. This is illustrated infigure 4a, in which the output FETs of abuffered gate are represented as twoswitches and two series resistors.The output resistance of an unbufferedgate, on the other hand, depends on thestate of the inputs. This is illustrated infigures 4b to 4d, in which a two inputunbuffered NOR gate is represented byswitches and resistors. If one input ofthe gate is high the output is low, theoutput resistance being the drain -sourceresistance of one of the lower FETs. Ifboth inputs are low then the output ishigh and the output resistance is that ofthe two upper FETs connected in series.However, if both inputs are high thenboth lower FETs are turned on and theoutput resistance is the parallel connec-tion of the drain source resistances ofthe two FETs, i.e. R/2. With multi -inputgates the variation in output resistance

can be even greater. This variation inoutput resistance also affects thetransfer characteristics as will be seenlater.

GainSince a buffered CMOS gate has twoextra stages compared with anunbuffered gate it has a much highergain, which is reflected in the transfercharacteristics. Figure 5a shows thetransfer characteristics of an unbufferedtwo -input NOR gate whilst figure 5bshows the transfer characteristics of abuffered gate. Not only are the transfercharacteristics of the buffered gatemuch closer to the ideal, due to the highgain, but it makes little differencewhether one or both inputs arc used,due to a combination of the higher gainand constant output resistance. With theunbuffered gate, on the other hand,there is a marked difference betweenusing one input and using both inputs.The gain of buffered CMOS is alsopractically independent of supply volt-age, whereas that of unbuffered CMOSis extremely voltage dependent. This isillustrated in figures 6a and 6b whichshow the gain of unbuffered and buffered

lastfered/unbutfered CMOS elektor september 1978 - 9-47

Table 1

buttered unbufferedpropagation delay 150 60 nsnoise immunity 30 20% of VDDnoise marginoutput impedance(four- input gate)

1

400

0.5 V

100.. 400 SItransition time 100 50 .. 100 nsAC gain 68 23 dBbandwidth 280 885 kHzoutput oscillation can occur undetectable(determined exper- with input forimentally) signals ..x inputs <

input capacitanceaverage

1 ms

1 2

100 ms

2 ... 3 pFmasimum 2 4 5 . 10 pF

measured for a 5 V supply and CL = 50 pF

Table 2

Application Buffered Unbuffered

High-speed

low -speed in high noise environmentslow -speed signals with slow edges

applications requiring constant outputresistance. e.g. D/A, A/D conversionlinear amplification, medium gainat high frequencieslinear amplification, high gain atlow frequencies

.

.

.

CMOS, versus frequency, at threedifferent supply voltages. A furtheradvantage of the greater gain of bufferedCMOS is that noise immunity is im-proved due to the better transfercharacteristics. The final advantage ofbuffered CMOS is that it has a lowerinput capacitance than unbufferedCMOS.

DisadvantagesButtered CN1OS is not without itsdisadvantages, however. Since a

buffered gate has more stages than anunbuffered gate is has an inherentlygreater propagation delay, which meansthat its maximum operating speed is

lower. Secondly, due to their highergain, buffered CMOS gates have a

tendency to oscillate as the outputpasses through the transition region ifthe risetime of the input signal is fairlyslow. This means that buffered gates areless suitable for low -speed systemswhere the pulses have slow edges. It alsomeans that buffered gates are lesssuitable for linear applications, wherethe input is biased so that the output isat half supply. Here again oscillation ishkely to occur since the output of thepie is in its transition region.

Figure 5. Comparison of the transfer characterrines of unbuffered CMOS and bufferedCMOS.

Figure 6. Comparison of the gain of unbufferedCMOS with that of buffered CMOS.

Figure 7. Interfacing CMOS to standard TTLand low -power (Schottky) TTL.

Table 1. Comparison of the principal specifi-cations of buffered and unbuffered CMOS.

Table 2. Applications preference table forbuffered and unbuffered CMOS.

Table 3. Main points of the JEDEC specifycation for B series CMOS.

Table 3

JEDEC-B

Absolute Maximum Ratings (Voltages referenced to VSS):DC Supply Voltage -0.5 to +18 VInput Voltage -0.5 to vDD +0.5 VDC Input Current)any one input) 10 mAStorage Temperature Range -65 to +150'CTotal dissipation 400 mVV

Recommended Operating Conditions:

DC Supply Voltage +3 to +15 VOperating -Temperature Range:Military Range Devices -55 to +125"CCommercial -Range Devices -40 to +85`C

Clearly, buffered and unbuffered CMOSdevices are not always interchangeable.so in order to help readers choose thebest device for a particular applicationtables I and 2 are given. Table 1 lists theprincipal differences between bufferedand unbuffered devices, whilst table 2indicates which type of device is pre-ferred for particular applications.

CMOS-TT L-CMOSIt is sometimes necessary to interfaceCMOS circuits to TTL circuits, and thiscan be done in a number of ways. Todrive standard TTL a current sinkcapability of I mA per TTL input isrequired. A normal CMOS output cansink only 0.5 mA, so to drive standardTTL a CMOS buffer such as the 4049(inverting) or 4050 (non -inverting) mustbe used. Two TTL loads can be drivenwith one of these devices. DrivingCMOS from TTL presents little diffi-culty due to the greater source/sinkcapability of TTL. The only minorproblem is that the 'high' output voltageof TTL may be inadequate to driveCMOS, so a pullup resistor is connectedfrom the output of the TTL gate topositive supply. This ensures that the

high output of the TTL gate is equal tosupply voltage and therefore adequateto drive CMOS. Interfacing CMOS tostandard TTL is illustrated in figure 7a.CMOS can be interfaced to low -powerTTL and low -power Schottky TTLwithout difficulty, since all CMOS ICswhich conform to the JEDEC norm candirectly drive one low -power (Schottky)TTL load. Low -power (Schottky) TTLcan, of course, drive CMOS directly if apullup resistor is included. This is illus-trated in figure 7b.In all these cases the CMOS circuitsmust of course. operate from the samesupply voltage as the TTL circuits (5 V).

B -series CMOSTo add another confusion factor to theCMOS scene, not only are CMOS ICsavailable in buffered and unbufferedtypes, but there are also two differentseries of CMOS. Prior to 1976 manymanufacturers produced 'A' seriesCMOS, a principal parameter of whichwas an absolute maximum supply volt-age of 15 V, although some manufac-turers produced devices capable ofwithstanding 18 V.A -series CMOS ICs were availablemainly in unbuffered versions, though

9-48 - elektor september 1978

8 10buffered/unbuffered CMOS

12

RCACD4307AE

4 9

Figure 8. Markings on CMOS ICs from RCA,who prefix their 4000 type numbers with theletters 'CD'. The suffix is either 'A' for A -series (unbuffered) 'UB' for B -series un-buffered and 'B' for B -series buffered, fol-lowed by 'E' to indicate a plastic package or13' to indicate a ceramic package.

9

Figure 9. Solid State Scientific CMOS IC.Prefix SCL', suffix 'A' for A -series unbuffered,'A/B' for A -series buffered, '08' for 8 woesunbuffered and '8' for B -series buffered. Thisis followed by 'E' for plastic package or 'D'for ceramic package. Maximum supply voltageof all SSS CMOS ICs is 18 V.

some buffered versions were produced.A -series CMOS has now largely beenreplaced by 'B' series, though there arestill considerable numbers of A -seriesdevices around, particularly on theamateur market.In 1976 the major CMOS manufacturersmet to draw up a standard for B -seriesCMOS, a major feature of which is anabsolute maximum supply voltage of18 V (though some manufacturers quote20 V for their devices) and a maximumrecommended operating voltage of 15 V.

M MC14011F

CP 7723

.MC1404 9 BCPM 77 -25

Figure 10. Motorola CMOS ICs. Prefix 'MC'.Note that Motorola use '14000' type numbersinstead of 4000, but apart from the extra '1'their type numbers correspond to those ofother manufacturers. Pre JEDEC B MotorolaICs carry the suffix 'A' for devices with amaximum rating of 18 V and 'C' for deviceswith a maximum rating of 16 V. Package auffix is 'P for plastic and 'L' for ceramic.

11

Co4040AE/TP4u4(0AN9/P767e

Figure 11. Texas Instruments CMOS IC. ThisIC carries two type numbers, One typenumber prefixed 'CD' is identical to that usedby RCA, the other is prefixed 'TP'. Suffix 'A'for A -series unbuffered or '8' for 8 -seriesbuffered. Package suffix is 'N' for plastic or'J' for ceramic.

The principal specifications of theJEDEC standard for B -series CMOS aregiven in table 3. B -series devices are avail-able in both buffered and unbufferedversions, except where the internalcircuit of the device makes buffered orunbuffered operation mandatory.

Identification marksThe problem faced by the average

Figure 12. National Semiconductors CMOSICs. Note the two different vignettes used bythis company. The 4000 type number is pre-fixed 'CD'. Suffix is '8' for B -series buffered,'U8' for B -series unbuffered. A -series deviceshave no special suffix. This is followed by aletter to indicate the temperature rang* of thedevee I'C' for standard, 'M' for extended),and the last letter of the suffix is the packagecode, 'N' for plastic and 'D' or 'J' for ceramic.National devices also carry an 'MM 5000' typenumber which is peculiar to National.In addition to 4000 series CMOS Nationalalso make a series of CMOS which is pincompatible with 74XX-series TTL and is

called the 74C -series.

13

F 4511PCNye 7704

INOONESIA

Figure 13. A Fairchild CMOS IC. Type num-bers of Fairchild CMOS devices are prefixedby 'F' and sometimes by '3'. All Fairchild ICsare buffered except where the circuit con-figuration makes this impossible. Packagesuffix for plastic or 'D' for ceramic,followed by 'C' for commercial temperaturerange or 'M' for military temperature range.

constructor, going into a shop to buy aCMOS IC, is finding out if a device is A -series or B -series, buffered or unbuffered.This is not helped by the fact that manyretailers know even less than the pur-chaser, added to which different manu-facturers each have their own code formarking devices.The usual procedure is that the packageis printed with the CMOS 4000 -seriestype number of the device, which has aprefix peculiar to the manufacturer anda suffix which indicates if the device is

buffered/unbuffered CMOS elektor soptembor 1978 - 9.49

14

HU. 45 32 HP

h 1/14

Figure 14. A Philips/Volvo CMOS IC. PhilipsICs are prefixed 'FIEF' and suffixed '13' forbuffered or 118' for unbuffered. Some PhilipsICI have the suffix 'V' which denotes that therecommended maximum operating voltage isonly 12.5 V. and that the device is buffered.The last letter of the suffix is P. for plasticpackage or 'D' for ceramic package.

A- or B -series, buffered or unbuffered.The suffix is usually A for A-senesunbuffered, A/B for A -series buffered,UB for B -series unbuffered and B forB -series buffered. The suffix may alsocontain other letters which give ad-ditional information such as the type ofpackage.However, this is by no means a generalrule, and the only way to be certain isto be familiar with the codes used bydifferent manufacturers. Identifying themanufacturer of an IC is in itself nomean feat, as manufacturers do notpnnt their full name but just the initialletters, or else some form of symbol orvignette. Figures 8 to 15 show sometypical CMOS ICs from the majormanutacturers. An explanation of themarkings on each IC is given in thecaption to each figure.

Using CMOS ICsFor low and medium speed applicationsCMOS is an almost ideal logic family, asit offers the advantages already dis-cussed, i.e. low static power consump-tion, high noise immunity, wide rangeof operating voltages and large fanout.However, it does suffer from one majordisadvantage, susceptibility to damageby static charge. The oxide layer whichseparates the gate electrodes from thesubstrate is extremely thin (typically1000 A) and has a breakdown voltagebetween 80 and 120 volts. The humanbody can become charged to several

15

kilovolts simply by walking across anylon carpet, and though the energystored is very low due to the smallcapacitance (around 300 pF) of thebody, it is still sufficient to break downthe oxide layer, which has an extremelyhigh resistance and very small capaci-tance.Unlike the breakdown of a reverse -biased PN junction, breakdown of aMOSFET gate is irreversible, since a

small hole is actually punched throughthe oxide layer by the discharge.Fortunately the manufacturers ofCMOS ICs do incorporate input protec-tion circuits. These usually take theform of pairs of diodes connectedbetween the inputs and supply lines,together with series resistors to limit thecurrent. In the event of a voltageexceeding positive supply being appliedto the input the upper diode is forwardbiased and the current is shunted awayto the low impedance of the positivesupply rail. For negative input voltagesthe lower diode is reverse -biased and thecurrent is shunted away to ground. Ofcourse, this protection is really effectiveonly if the device is in circuit with thepower applied. When handling CMOSdevices suitable precautions must still betaken and even when a device is incircuit signals should never be applied tothe inputs with the power switched off.Unlike TTL, the unused inputs of aCMOS device must not be left floating,as the output of the device may then bein its transition state (around half

Figure 15. All the S's. The foregoing list ofmanufacturers is by no means comprehensive,and Signetics, Silos, SGS ATES. &lurk ndSolitron could be added, to name just thosemanufacturers whose names begin with S.However, the examples given should help thereader to know what to look for in IC

markings. Note that the remarks made aboutpackage suffixes apply only to dualip line(DILI packages. Flatpacks, rarely used by thehome constructor, have different suffixes.

As a final note, different manufacturers havedifferent ways of indicating the orientation ofan IC. Most use a notch in the end of the ICadjacent to pin 1; some use a dot, bump orfigure 1 on the top of the package next to pin1.

supply voltage) in which a large currentis drawn. Unused inputs must always beconnected to positive supply (VDD) orthe zero volt rail (VSS).To summarise, the following precautionsshould be taken when using CMOS:I. Store CMOS ICs with their pins

embedded in conductive plastic foamor aluminium foil - not in expandedpolystyrene (which practice is notunknown).

2. Don't handle CMOS ICs any morethan necessary.

3. Work on a metal surface, such as atin tray or aluminium foil, earthedthrough a 1M resistor (A I M resistorwill leak away static charges, but inthe event of simultaneous contactwith mains live and the work surfacethe current flow will be insufficientto give the constructor a shock).

4. Before handling CMOS ICs, alwaysearth yourself to the work surface. Ifyou leave the workbench for anyreason, earth yourself upon returning.

5. Use an earthed soldering iron.6. Connect unused inputs of CMOS

devices to VDD or VSS.7. Don't apply signals to the inputs of

CMOS gates if power to the circuit isswitched off.

LiteratureRCA Application note ICAN 6558.Data books from the major CMOSmanufacturers.

9-50 - istoktor saptinnbor 1978

1.

New calculatorsA new series of five financial andscientific calculators featuring anadvanced degree of humanengineering was announced todayby the Hewlett-Packard Company.Prices for the new Series F.calculators fall comfortably withinreach of students and youngerprofessional and business people.Calculators in the new range arethe HP -37E business management,HP -38F advanced financial

programmable, the HP -31Escientific, HP -32E advancedscientific and 11P -33Eprogrammable scientific models.All feature more powerful. larger,easier -to -read displays, diagnosticerror code systems, accuraciespreviously available only in themost advanced calculators, non-slip ruhber base pads, and positiveclick -action keys.

ltd.. King Street Lane.Winnersh, Wokingham, Berkshire,RG1 I 5AR England

1827 MI

Compact lab. supplyA new low-cost, three -in -onebench Power Supply fromHewlett-Packan1 is designed forengineers who design and testbreadboards and prototypes usingintegrated circuits. The compactHP Model 6235A Triple -outputPower Supply delivers threeadjustable DC output voltages:0 to 6 V at 1 A, 0 to +18 V at0.2 A and 0 to - 18 Vat 0.2 A. Asingle 0 to 36 V output at 0.2 A

.00

can also be obtained byconnecting across the 18 V and+18 V terminals.The controls, meter, and bindingposts are all arranged convenientlyon the front panel. One voltagecontrol simultaneously adjusts the+18 V and I8 V outputs, whichtrack one another to poweroperational amplifiers and othercircuits requiring balanced positiveand negative voltages. Thesupply's dual outputs have addedversatility with an adjustabletracking ratio control that can setthe negative output to a lowervoltage than that of the positiveoutput. Once the tracking ratiocontrol has been set by the user,the voltage ratio between thepositive and negative outputsremains constant as the +18 Vvoltage control varies bothoutputs. A separate voltagecontrol sets the 0 to +6 V output.The supply is of the constantvoltage/current limit type witheach output voltage beingcontinuously adjustable over itsrange, while the maximumcurrent available is limitedautomatically to preventoverloading. The unit's outputsshare a common output terminaland are isolated from chassisground so that any outputterminal can be grounded ifdesired. Voltage or current can beselected and monitored quicklyfor each output with pushhutton

meter switches. With dimensionsof only 90 mm high, 155 mmwide, and 190 mm deep, the newsupply is small enough to bepicked up with one hand andtakes up a minimum of benchspace. The HP6235A weighs only2.3 kg. It can be powered from115 V or 230 V.47-63 HzAC input.The Hewlett-Packard 6235ATriple -Output Power Supply ispnced at E 119.lIewlert-Packard Limited,King Street Lane, Winnersh,Wokingham, Berkshire.RGI I 5AR England. (828 MI

20 MHz scopeThe Leader LBO -508 is a dual -trace oscilloscope with bandwidthof 20 MHz and sensitivity of10 mV/cm. Stabilized powersupplies ensure a measuringaccuracy oft 37+, and the displayscreen is 8 x 10 ems. Fixed andvariable controls cover a sensitivityof 10 mV/cm to 50 V/cm, andwith an add function and CH.2trace invert, inputs may be addedor subtracted. Triggering may heselected from C11.1 or 011.2 and

the circuit will extract syncsignals from both T.V. line andframe signals. Timebase from0.5 ps/cm-0.2 sec/cm also has avariable control and X 5 magnifi-cation. AUTO/NORMALtriggering is pmvided, as is anexternal trigger input. AnX -Y function switches one inputto the horizontal axis forX -Y display.

Martmn Ltd., 20 Park St.,Princes Risborough,Bucks. England.

(830 M)

1 W audio ampRecently added to the range ofSprague low to medium poweraudio devices, is the ULN-228313integrated audio power amplifier.Its predominant features are thewide operating voltage range andlow quiescent current drain. Aminimum number of externalparts. together with 'on -chip'short-circuit protection, enhancethe system's reliability.The wide operating voltage range

makes this device eminentlysuitable for use in hand-heldbattery -operated equipment.In an environment, where space isat a premium, an importantfeature is the fact that this circuitdots not require a heat -sink, asthe copper -alloy lead frameconducts the hem into the printedwiring board.Sprague Benelux, Industrie:one,P.O. Box 104, WOO Ronse,Belgium.

(832 A41

Low power IF/AF circuitfor fm receiversPlessey Semiconductors newSL1664 circuit offers the entireIF and audio sections of anFM receiver, including a 250 mWoutput stage, in a singleintegrated circuit. It may be usedwith IF% between 455 KHz and21.4 MHz and will give supersignal/noise ratio with lowdeviations and high II' s. as wellas low distortion with highdeviation and low IF.The SLI664 has a sensitivity of10 microvolts or better, and astandby consumption of 20 mW,making it ideal for both low -pow ei pocket receivers forbroadcast FM and portablenarrow FM transceivers.The device incorporates a stablesquelch system and operates fromsupplies of between +6 and+9 volts. Its operatingtemperature range is - 30' to+70 'C and it is supplied in an18 lead plastic DII. package.The SL1664 Au designed to useexternal LC quadrature circuitrywhich means that even in highperformance receivers for narrowdeviation signals of 1.5 KHz orless, simple quadrature circuitrywill give excellent S/N ratio andexpensive crystal quadraturecircuitry is not necessary.Plessey Semiconductors,Cheney Manor, Swindon.Wiltshire S.V2 2Q W, England.

1837 M1

TTL * *A14,5.or14 -

.14 ,

AAA110AleoAAAAIleAMeMe,....AAMe,,,PAP

40,A641414Oft

114114IPAM.15.,a,IleIN110VIA.....Dv ,'StA o

rAANoA 'Up tAONto

'1/.7404,Ile 'Ple ,Aall, r

MO 7

NAA '41"/M

.. A. pb

,- IIIP. AA1400... op

1,1 A 13 sas.24 mosIs MI1 jip 1,1,I,' 14,,,I, la.IA A129 sp.I1' 14.,,. ,,,,'I' pft'41 2/4-.43 Pb,.. TAA A.4 nor.4 1644.90 IA, es.'1.1 siriv lAIsA pp,

op.Pp,

Pp 1110

3' 519I; ..'el 14,so 124At IA' 314.., mos.. I IA4 A. .3 A..0 ops.

' IA.., A*... ....

libI. Ato ripou nepr. A,gm WmPG RepIC MN

10s2 Ma5"..so mem44 alIS AI, Agol IAes IA

m MAO6354 n.

11114

"'LINEAR Y3.4.533." rs'ea: 105) A OMIT A. , .. A AAA Mem) I 'IP SA ..ca A

Me 114,02P-0 AmKIM Me sea Iiiallo. ISANN 10114. i101.21,1% 101,woo 44. sums INN

C47/14 MN 5.510111014 5LAS 3146. .55 A(0110111 AU A IPAPPO IACA), 10 INA r 5160 Alm61'00 IA NASH M.CAP 731 ale I aadel 1464LAO, ,. IA AA KO 3110. IA TM 1/11. 'ppm 36400.11" A 1°110 lab.9.391 A rApro IAOAT 19 16P mobs A,Venom 1144 rs"01 .34teen , . 114 ro644101 IAL1"004 '4.0 1ASJO0 nag 4 1' ^ ''' 1644 7.46300 PN.... 3' ' 1.1P 1911. 35140 230.43". A '646400 11141.

15 644100 oft;1"" MA 'AAA A.P180. A 1564., ? A. VI.' IA 5A VI KOOLIPS% A OA,* PA1150/ A s .260 AP.P.V0 01 No sa somo mms.

,11.12 I.. A 51011 lifeL107)1 151 A .1410 11001.1"/3 LPL A '11611.10 IA.10/II IA 44.0000 100.16`4' A 'NM A.34 4 46. 130))10 /10,. KIPS 7,04 4 42 KM1 AP.room A r 5 PALOA A TA op, 124...111009 A 120 Taotm /,-14 11541 A 5 :Al 11.4 ' I' 24 Mg 120' 4144V. 4 A AA 044..,..

.310 .1122.6 A,ou's 1'4 Mall ...r./ au.14455 Mb 545 15NI 643 Ms .54201WA IA P.141'. IA014,1 MO, ALL ,4 110,

1411 A11 1

"TRANSISTORS I14 14 163.1.. AA 111 121 "15 4! ` " II, Of 111 A 11114 14 'I' A 511. MA Al MO PIP 1111 n.4 'M Alp at 'el A11121 Or 4. MlNT''', ' 2 lap or ,4) A 0617) 44. AV ROI 11,A. '1' "' Alp 'Al 34 11 111 A ' 01 AAt )4 lb w- ;90 lb 111)1 'A II* INA, 3' Ja. .a,,,, A 11 1 0 elm lbAt".1 A ou N. lb OS IPA to EPP. A

1151611. COVI1113) ClOrk II TIMEII 535 Op 11561A. A 111 SAC. 4917064C. NA 115T. A 24 `5. 1019NSA 110p AI SWIG 114 30 14C. MAP12 92.2, 00, L4 %Pg. Oft ).SAG 664IS WIG 5 40 She. 7164 11'.44 1110

A. '.. A P, 146 ja, IP MI 10ap AA. "A MO PC /114 lb 111(1 i

Al 1 /Op PM 104 lb 01 11 AU IP, AA. 64 11 11? IA III 11 A 1 420 IA4 ' A 11, 1,2 11, 11 IP. A 1420 IAA. . . 34 /V PP 11 60 /PO A re I* Me0L .2 os. 10113 10, G01101 A

NA I. 2, 1, 60 113 595 lb

GI

ROTARY 04141TCONSOVLOGLINI As). I 12 5.111..t 6 AA T . 'A KM 3 n I AI I at alp 0445

4. 4. AAc.,,CC A I "1 pa, 51 151 Pe liolVtd. 16.

. 64 2,4. Ilk II IN lam "Al, 54. ' Al 1414 / ' ' ma 10111 Mo. 1" Pep1" 4 Isms . :I'll lb 10 A .... ?WA A4 "" IA Gm 5 A 10 Mb ....; ,,,ap 'A "I A A. ZO ma no OS 44, 1.10.4 A

'1 Um a No MIL ..o.101. NA .4 Flo II gm Ng ' ' "" Ms ..lAl "11 .. I" A IP A A 1, A

A' ' A 1, P 11. 1014 32. : / ' .14.

A' ' ''' A 4 5 14 ..IM A -," AA' ' ,.. a IT 163 0. /P. A , "' Ow

,..3.2 606.P A G "1 I* N. A" 14.,A sho A. 54

OPTO ELECTRONICCORNER'APEG11. SCOOP 04 filon 1" 0, O) or. RID LIDSIte mmt. '0 'L. CI 20 ICO atefa 00 . 'PSI "., 110 00

; 0 1.I I -ow. 'apon, C000tora -26. '2, it, . S.1.5C1A

1 12 00 mako .29.42 Loo, '..5.151

2.4 5 gsAlen PI,AP 2 )9 44'!: SOO

AL .: IAP"' ''' lbP.'"0'14 A°A ' . Ile

14

25 A1144 6 no

11 A 0' /11 To AIA A .h Pao "Ow10 5 5 0 pj. MIP . .44' Ple .I. 2, 5 , A 04., '4 A MD usA 11 lb XSA 11.14 kenA."' leeA 16. ''^"" A

.1 r I" ., ...Milt 0 .112441. .." A.., ...No OS"CI 30 NO 4 5 11.011.".1, ; 1 Pt,. lb DV ....conAA

a 0 7 001.5 AM'0 IP*Goo , i ns 2646166 1011 to./10000111117 7101"1I"' ' * ... A aft Mog040 140 M. I* A ll

IAA mallwaed 11 2Sgml.

4105" *.A4" lib11 IN"' " me.

1K '01 A6 726 IN14( OK AMs '00 lasN'00 A

i a.r :

544 A so 5 A%AP 14, .01.141 A

44' APe.5 P.P..% Pk PP

11' 110 5.10, aft ..) 14 pp.500,9. M illb So A OP

g. 4 Mb Mae . 56.24 64et 42 A OA 34 le4/ 3 14 MO A 5 TPA" No6. 99 A NA A 4..- .1,0 lip mo OM A 1.5-4P 110

DECODER BOARDCONTAINING15. 74196 2. 24--I. 341101. MIS2.60 SAY LOGEFA* only so6 01 OA - .

13 1041 14m Magog' Jo. '"'' IDIM' At 0 PO A " ' AP ar

IK 111 A s. '2 A mo IA a% /1414G IA29,321

er.564160615.o OA111 MIAL 1.61C ImPts Tip mgdoo6- ANNA SSW 15

0 --- -POWERSUPPLY CAPSMA 16 Ile2 Mb Al) 15IND ILI MO 4100 PO 114DIM 103 Mb ,mm 53 IA1110 MI A Keil r, MNSST 63 A WM IA4100 ft A 251 ls A4KO 11. bbI 52,01'. len A"' dimPLI .1,

A

04.1. 2 op II, A PoTe 34. A25 13It," I* ?III, A 5. 64 A A

BOO 14 4.1 I' Me Do 4 A '.. 261.

KIT 14 toil Op so in IA "' 1041,415' 1/4 IP,'" NO Do 113 A 29 SO mas

1S1,4 A G'16 310 a ... ' "hp 1% n1 1.A

0a 1'4 II, SO' A me a. Te PT.' AA A10111 14 111' l' A M.. my 1y'" ASOP 14 1101, A 4. .. n A 'A 7.4GC IN 20 sou., AI. /1)441 10,04.140 IN 10," IA a 9 At0/1,1 A 6 c 4 moo so , las .^./0" ,p,

jp, 0015/ IA AI . A. .. NA . Illpet 14.2 Pesc I I .., swot .14 40 S2 1 .-.. i,P1 %14,

IN 14, 5 1402,, ' OA so .ss 15 'NI.'" ...'K 1441 10, 11011< Ole 61401 To inp.,.64 A 14, 111):27 ole aa 6 14AC TN me 1n21) sag 466 A .14 1101111 A 2,41 'X lb

T.5 NO I C111.0114m654455..... ..... 60- 1004 oa

SIR or p64

0.. .

P.

It* 44,"'- - 16144441

I 5104 St 102 Pro 151.0.5.14 Doi 14615 U.4.6 LA .44.0. IA PA CoN IS 00

41

416Ais.IA

50

IBA OOP9'2 or'4,1 Wile

"" .....%..., , -

A

11 1

ELEC CAPACITORS

1, II" 16 orse

0It 6' orPp 4,21 Pp aP

tin 1, 4, SO llPo OD '0 6P

1 I A IP PM ' 44 ' 'A It lb " Or

N 'CO SO am

IP KO 61 Aa So N 2341 If lb41A 'PP 7/0 IS A'0' 0 4 MN lb'0 4 TO ?I IN1029 Is 1343 16 OA15643 4 330 90 A23 111 N 10 10 A13 0 41. AAA21 111 N 53 Al A2211 Pt 410 110 A22 A ha IS. 14 26022 SO AIMO 76 341361? Pp 1000 lb A

11 ,°0040 A33 PS Op 1000 63 A.4301 PP , Pr. 01 A11 IP 1, .1 ' Mb

gc 61 SA A lb6(196

O1101 11.

KI61114 CIA/ .M1,91

24A 1616 1,1*14MOM 2915 AOr 0ke ,.. suen , ele o INA 149. 'I.'''. 1401.

.a.a 504109 is. mini sas '''.' '' Ao6. '11, lap 110404 POP mom Isom /nom mg. 44 IN BOA 0 110 MLA 224 "11°1 5

Pee 161 1.011.

,......... .... moo 'I/ OA. 11,11011

6.

MICRO BLOCK

2102 250 543.4 SpAA., PAY 11024aA' T 170 101

TTL OFFERS

72 '63OAP letON .04ON 10 4.53 10 OM 11)4NM MapA 104Ism as.IS 1010'00' ,...

a

14aAAA5OAlbNo444AA06.

4

a on 4. DO 4 12 11114 s toe 1.. :AOKI" ,p,.. , oss ors. 1 moo 4124 IN "4 4 4 14sA

A

rITALIA 104COMB.A3 WA PI

4 .p. L140a no 11600 */WI 4'1. 4., AP.S. 2,31.1 '724. '*

1104141

22 2.010 1 111 11.IPAS100 5 1110 PAN 04'A 5 NM mom essa 4.

.2

0.43111110161 ON

' 0.9 vole.L.' 04 'CA LOG'Oa ov. lit LOG

* it-LOOTFOOFFIRDUI/noo os.ES.00

A.A4 10 15.7Mbtat 16 5,3Ma1.41alagp,Mb

44,1 ll 10 564 5 CII 0015 (II MI *2 17 450 ..44.3-54Sot./ PAY .766 4

a Ord.`1 PM .103 16.09sox or. 'OM * *,,_ a .1. l A

A4 c'(II

irsaLUSIPIOV14 III I 10 51210OM 5 En 00

VA I .111.14to so. S21 50

LOG10* 114. 11111 1151)A 1. Ken LOGum0Los sa Lot.A Lie TO LOG/V 1 04 ... 38. e...

S

ILIEni00SADO

As o A 313AM pans .4 flea

7IBMSoa amtFm s4501...., r 31. 14 15

la I04 1:22 SO 7(saw

2x][4"4.1.13(P1P101,101 um*,5 (7 Mirth *

CMOS

4011 210414, AA IA14 AI Mob

AO AP140 4M0* A50 A1611 MIN 13041Me 4063 IA60. 435 1214rs 1 AC A

.1 94- el A

, C A. 14

, o oss

..., v..011/ A41.5 IA

. 41,11 IA4620 1104. asm ISIcy

glo.NAIA

11 .1 IP A.5 *10 to C2 00

43561.C. POO

1. 1.6 ..14.. I I ,C

0 26 0142 Comom.141664 .161Oilen Wt. . AO'MI5 As 5

AAmgconegos

1rlev Mb

261) CnrrleFLAnporagro "noom raw

11 OS am**446310,E Rao.

POLY CAPS544 4

' 4Ppn100 10 l /I .1A a. a II Sp

a /90 4 047 -I OA41410 *001 _Pr 4 2. 2 .1 no027 .1 14 r .# ISO0 036 ...0 4 All a300 AI * my

'1

WHAM MODULESecm . rom LO A.0.45 ale 3 . r LUGso. 144 LO 0 LOG

Ayhaws dgmaagssom.6 mil, KAN TA

Axe AN. A II 00..11? t,5211 =

.

1 P 1167 10101,1111.3 4000LP1116 4/4141.11116 40106111169 5012LP I ISO 4006

IA 15 .6 A 3 OGA 7.301. Liao r LOG9C0a KOK i oo. a a 0674 OP' 04 0 "IGno No 34 A LOG

4444 lop 15

41114 AN

MI6 PIOMOM .5. ool) 15l v123 IA

&II olin MaMe 0. IAMar MON AMN PT elAMP 15411, AAN Prn. 114

.4

412 Ill, op..a_'12.11131 TT Stele Looli

D. me. 12 00 amok

all T 01019 15663111 77 411066301181 MOO611402 41106111000 211061410)1 nibGRIM 2104

,41

Om 10C4114564 lag14 Po. 14IG Ns *So

GorlA .14'OA I*.a moo OAar Pa 504MEN.a IP re Am Aot.44 Ma11111,1/ ,144ANT 1 5

M C A TIOAC1

4COs I AVM(1 /0

6110r '4 AV,II IC

T.4445e.120056

I., Lome.TI140144n ea ..,,,

AV'S- 1013 U l, '

A 11100

TANT. BEADSal %v to, 1 1 MA 14.10,111111 I. 4 PAN my0 rralv 14. 4 , 26V 14.pulpy 34 4 3 Tv MAo a/ Mb 14 553 A0 4 /TV 14 1111161/ 140605 A 1 9TV A1 5104

flap 116111al.I Was lap 16 111 34'GA A. 435 los

,).111v 14 al119v ba22 ma lap vastrn IN

1

111110311111*So DAT GAIN! OR 5.01,0 eled A ENKAisms 31 01320IA aw..1, 1AMWM/ CAMIA IA2 bp. _ep2 AM AnSW 130 race 204/ 010 -Agog100; NO *AIN NedFERRITE KAM6644 leop 00 MAID 115bar ioo 444 1200

. PON306 ST PAULS ROAD HIGHBURY CORNER

LONDON N1 01.226 1489 STOP mOuRSMON - 4404_ spasmSAT 9 4)O 1410

,101 RATA ULTMASON I-- 4 AWAKE PV2401.114 S32 10 Each E.00 Pam

11111511.14 trAl 430. A%NG 160.1316 0 A...11, .OA COMIS I 0 519534 P 0

Raw 011.11 100V AC CAMAAP CI A116615150/3 OM makAmmo.] lormovm 1 4

. r PSI AK 473114711,3 S RI k I l ALWAYSA16, 4111. a ..._r i OM(

441.1 16 1 14011SPECIAL OFFER

.461., ro . mar Ism

551462.64n 20 4.4 Cl 26

A.

WALK."." 1110,05180 C01164 PI

TUN

ALL PRICES INCLUDE POST AND VAT