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Wouter Faelens
D-mode GaN HEMTsDesign of a multi-MHz resonant driver chip for high-voltage
Academic year 2017-2018Faculty of Engineering and ArchitectureChair: Prof. dr. ir. Koen De BosschereDepartment of Electronics and Information Systems
Master of Science in Electrical EngineeringMaster's dissertation submitted in order to obtain the academic degree of
Supervisors: Prof. dr. ir. Jan Doutreloigne, Dr. ir. Pieter Bauwens
Wouter Faelens
D-mode GaN HEMTsDesign of a multi-MHz resonant driver chip for high-voltage
Academic year 2017-2018Faculty of Engineering and ArchitectureChair: Prof. dr. ir. Koen De BosschereDepartment of Electronics and Information Systems
Master of Science in Electrical EngineeringMaster's dissertation submitted in order to obtain the academic degree of
Supervisors: Prof. dr. ir. Jan Doutreloigne, Dr. ir. Pieter Bauwens
Permission of use on loan
”The author gives permission to make this master dissertation available for consultation and to
copy parts of this master dissertation for personal use.
In the case of any other use, the copyright terms have to be respected, in particular with regard to
the obligation to state expressly the source when quoting results from this master dissertation.”
January 2018
Preface
First of all I would like to thank my supervisor Jan Doutreloigne for advising me throughout
this year. He challenged me to really dig into the matter and understand it at a fundamental
level. He also helped me along when I was stuck on certain problems, and provided helpful
suggestions where needed. Lastly, I enjoyed his work ethic greatly: doing what needs to be
done, with simple down to earth methods and explanations, without the need for rigid official
systems.
Secondly, I would like to thank all my friends for giving me joy and energy. First off my
flatmates Jasper and Lukas, for giving spice to my life and putting up with my lesser sides.
Secondly, thank you Wouter and Rori for inspiring me to work hard both on my thesis as in
general, as well as thinking critically about life and society. Lastly, I would like to thank ’de
duivenmelkers’ for giving me so many good times in life, both as gaming buddies as well as in
real life.
Finally, I would like to thank my family in supporting me for all this time. Thank you
mama for showing me the joys in life, like travelling, and supporting me no matter what. Thank
you papa for driving me to not settle for good enough but strive for perfection. Challenging,
encouraging and helping me to give the best of myself, while still keeping in mind what is really
important in life. Thank you Femke for giving me joy and being awesome, you are the one I
would turn to in hard times. Thank you Ruben for being my ’big brother’: a living example of
things to achieve in life. Lastly, thank you Linde for reminding me why we strive everyday to
make this world a better place. Special thanks to my dad and Ruben for giving suggestions on
this thesis, improving it so many times over, and elevating it to something I am proud of!
Wouter Faelens, janauri 2018
Design of a multi-MHz resonant driver chip
for high-voltage D-mode GaN HEMTsWouter Faelens
Master’s dissertation submitted in order to obtain the academic degree of
Master of Science in Electrical Engineering
Academic year 2017-2018
Supervisors: Prof. dr. ir. Jan Doutreloigne, Dr. ir. Pieter Bauwens
Faculty of Engineering and Architecture
Ghent University
Department of Electronics and Information Systems
Chair: Prof. dr. ir. Koen De Bosschere
Abstract
Depletion mode (D-mode) gallium nitride (GaN) High Electron Mobility Transistors (HEMT)
can be used as the switching element in boost converters, a type of switched-mode power supply
(SMPS). Charging the gate capacitance of the switching element at high frequency provides
an important contribution to the power losses in this application. By using an inductor and
recovery path to recycle this energy, the power efficiency can be improved.
To evaluate the power efficiency, a resonant and a direct driver are designed that drive the
HEMT directly at its gate. We also incorporate an internal negative voltage generator so the
driver can be controlled via positive signals and a single positive voltage supply.
Through simulation, the power loss and efficiency of the direct and resonant driver are compared
under different parameter values: duty cycle (0.2 - 0.9), switching period (200 - 1000 ns), resonant
inductance (1 - 10 µH ), diode types (Schottky vs normal diodes), booster supply voltage (1 -50 V) and load impedance (5 - 500 Ω).
By driving resonantly, driver power losses can be lowered significantly, especially at higher
frequencies. At 5MHZ switching frequency, the resonant driver consumes 89.6mW while the
direct driver consumes 197.4mW. The boost converter with the highest efficiency has an efficiency
of 84%, 2% more than in direct drive with identical parameters. Typical efficiency increases for
most parameters are around 1% - 2%. The largest efficiency increase comparing resonant and
direct drive is an improvement of 17% (59.5% versus 42.5%).
The simulation results show that a resonant driver is more efficient than a direct driver. This
result is robust under varying parameter values. However, a resonant driver requires extra off-
chip components, making it more costly to produce. It will depend on the application if the
benefits outweigh the costs.
Design of a multi-MHz resonant driver chip forhigh-voltage D-mode GaN HEMTs
Wouter Faelens
Supervisor(s): Prof. dr. ir. Jan Doutreloigne, Dr. ir. Pieter Bauwens
Abstract—This research tests the efficiency improvement of driving a D-mode GaN HEMT in a resonant way. It does this by designing a resonantdriver and direct driver and comparing their efficiency under different pa-rameters via simulations. The proposed resonant driver is predicted to havea higher power efficiency than a direct driver for any switching frequency,booster voltage, load, and for duty cycles ¡ 0.5. Efficiency increases of upto 17% were shown (42.5% to 59.5%), although typical efficiency increaseswere around 1% - 2%. Driver power losses can be reduced up to 50%, espe-cially at higher frequencies: 89.6mW resonant driver loss versus 197.4mWin the direct driver, this at 5MHz switching frequency. However, a resonantdriver requires extra off-chip components, making it more costly to pro-duce. It will depend on the application if the benefits outweigh the costs.
Keywords—Depletion mode, GaN HEMT, Switched-mode power supply(SMPS), resonant driver
I. INTRODUCTION
EVERY electronic circuit needs power. This power can bedirectly drawn from the source like batteries or a poweroutlet, but more often than not a special power circuit is usedto convert the incoming power to a more suitable power for therest of the circuit. In an electric car for example, this could betransforming a low battery voltage towards a high voltage usableby the motor.
One of the most common power converters are switched modepower supplies (SMPS). These power supplies utilize a switch-ing element to convert power efficiently. Instead of using a lin-ear amplifier, the pass transistor of a switching- mode supplycontinually switches between low-dissipation full-on and full-off states, and spends very little time in the high dissipationtransitions, which minimizes wasted energy. Voltage regulationis achieved by varying the ratio of on-to-off time, called dutycycle. In contrast, a linear power supply regulates the outputvoltage by continually dissipating power in the pass transistor.
In this abstract, we will use a GaN HEMT as the switchingelement. This switching element needs a driver to turn it on andoff. This driver consumes some power, lowering the efficiencyof the total system. In this paper, we aim to lower the power con-sumption of this driver by recycling some energy via a resonantcircuit.
II. CONCEPTS
A. Boost converter
One of the main power converters in use is a boost converter.This is a DC/DC power converter that boosts a low voltage to-wards a higher voltage. A circuit of a typical boost convertercan be seen in figure 1. When the transistor is on, the inductorexperiences a positive voltage and current is built up. After ashort while we turn the transistor off, and the inductor current isforced through the diode towards the load. This current charges
the output capacitor to a high voltage. Because of this high volt-age, the inductor experiences a negative voltage and the currentgoes down again. Charges are pushed towards the high-voltageload until either the transistor turns on again (continuous mode)or the inductor current falls to zero (discontinuous mode). Thecharging cycle can repeat again.
While the transistor is off and the inductor current builds up,the output load is powered by the output capacitor. The outputvoltage goes down during this time. This means the output volt-age goes down slightly when the transistor is on, and the voltagegoes up again when it is off. The larger the output capacitance,the less this voltage will bounce up and down.
The faster the transistor switches, the smaller the inductor andoutput capacitor can be. A faster switching time means the in-ductor has less time to build up its current, with a lower averagecurrent as a result. This means the booster circuit will lose lesspower due to conduction losses. On the other hand, if we doublethe switching frequency, we can halve the inductor value with-out increasing our average current. The output capacitance canbe lower as well, because it needs to power the load for a shorteramount of time. Lower inductance and capacitor values have theadvantage of being cheaper to produce on an integrated circuit(IC), and having lower parasitics.
In order to switch this transistor on and off, we need a driver.This driver consumes some power that is wasted, reducing theefficiency of our system. In this article, we try to minimizethis driver power loss. We do this in two ways. First, we usea GaN HEMT (Gallium-Nitride high-electron-mobility transis-tor). This is a type of transistor with very low parasitic capac-itance. This means we do not need a lot of charge to turn thetransistor on or off, and hence can switch it faster with an equalamount of current. This also means we can improve the switch-ing frequency. Secondly, we use a resonant circuit inside thedriver. This is a type of circuit using an inductor to recycle someof the energy used to charge and discharge the gate voltage ofthe transistor.
Fig. 1. Circuit boost converter
B. GaN HEMT
GaN is a wide band gap semiconductor (band gap energyEg=3.44eV). This wide band gap translates into an ability tohandle high internal electric fields before electronic breakdownoccurs.
GaN is particularly interesting because of its ability to formheterojunctions with wider band gap semiconductors such asaluminium gallium nitride(up to 6.2 eV). These heterojunctionsresult in the forming of a 2-dimensional-electron-gas (2DEG) atthe interface due to large polarisation in the material which pro-vides a highly dense, majority carrier channel with a large elec-tron [1]. Due to the 2DEG, GaN offers a much larger currentdensity than other materials. This makes it so the gate can besmaller, resulting in lower gate capacitance while still maintain-ing a great on-resistance. This also means there is less chargeneeded for switching, resulting in lower switching losses in thedriver and faster switching.
Due to the presence of this 2DEG, we have a normally-ondevice, or a depletion mode FET (D-mode FET). This meanswe have to apply a negative gate-source voltage in order toturn the HEMT off. Enhancement-mode GaN HEMTs havebeen demonstrated, but their performance is worse than D-modeHEMTS.
There are two ways to generate a negative gate-source voltagein order to turn the HEMT off. First, we can tie the gate toground and raise the voltage of the source. We do this by placinga MOSFET at the source of the HEMT. When we now switchthis MOSFET, we can raise or lower the voltage at the source ofthe HEMT, effectively turning it off and on. The MOSFET doesnot need to be able to handle high voltages, since it only raisesits drain (the HEMT source) high enough until the HEMT stopsconducting and is off. The advantage of driving it this way is thatwe can use conventional gate drivers to drive this MOSFET. Thedisadvantage is that we need a large MOSFET with low enoughon-resistance in order to have good efficiency. This also meanslarger parasitic capacitances and bigger switching losses.
The second way to drive the HEMT is to tie the source of theHEMT to ground and put a negative voltage on its gate. This ismuch more efficient, but the driver will need to be able to useand generate negative voltages to be able to do this. Because itis more efficient, we choose to use this driving scheme.
In this thesis, we use a MGG1T0617T. This D-mode GaNHEMT has a threshold voltage of -3V and low parasitic capac-itances. It can operate at drain-source voltages of up to 600V,meaning we can use this HEMT in high-voltage applications.
C. Resonant driver
The second way we try to improve efficiency is by using aresonant driver [2]. The circuit of the used resonant driver canbe seen in figure 2. When we charge the gate voltage by turningMDR1 on, we also send a current through the inductor. This cur-rent keeps building up, until the gate voltage reaches the uppervoltage. The voltage will not go higher because it will drainaway via the upper diode. The current will now free-wheelthrough MDR1-L-DDR1. When we turn MDR1 off, the currentin the inductor wants to keep flowing. The only way this currentcan flow, is via MDR2-L-DDR1. This is effectively pumping
current from a low voltage towards a high voltage, recoveringthe energy stored in the inductor and destroying its built-up mag-netic fields, lowering the current through it. After some time, thecurrent will have died out and the driving circuit is in rest.
Once we want to shut the transistor off again, we set MDR2on and a current starts flowing from the gate, through L towardsthe low voltage. This charges L again, transferring the energystored on the gate capacitance to the inductor. Once the gate ca-pacitance has discharged completely, the inductor current free-wheels again via DDR2-L-MDR2. When we now shut MDR2off, all the energy stored in the inductor will be used to pushcharge from a low voltage to a high (recuperating energy) viaDDR2-L-MDR1.
This way, we can recover a lot of charges otherwise dissipatedin the circuit, and improve the driver efficiency.
Fig. 2. Circuit resonant driver
III. MAKING THE CIRCUIT
In order to drive the GaN HEMT directly at its gate in a res-onant way, we need a few things. First, we need a negativevoltage to put on the gate. Secondly, we need to translate the ex-ternal positive control signals into negative control signals thatwe can use to drive the HEMT. Thirdly, we need to design theresonant circuit itself. Lastly, we build a boost converter thatuses the HEMT driver in order to test its working and efficiency.
We start with generating negative voltages. We need two neg-ative voltages: a small one to drive the first level shifter of -3V,and a second one to drive a second level shifter and provide thenegative voltage for the gate itself at -5V. This second generatorneeds to be much stronger. We construct two Dickson chargepumps [3], which generate output voltages of -3V and -5V.
The negative charge pumps need a clock signal in order towork. We construct an astable multivibrator [4] to generatethis clock signal on-circuit. Because the dickson charge pump’svoltage output depends on the load, we design some additionalfunctionality. We put OR gates at the input of the charge pumpclocks with as input the original clock signal and an OK sig-nal. This way, when the OK signal is low the clock just passesthrough. When the OK signal is high however, the output of theOR gate will remain high and no pumping action will occur.
The OK signals are generated by two simple opamp compara-tors. When the -3V voltage is low enough, we output a highV3 OK signal, and when the -5V is low enough, we output a
high V5 OK signal. This way, the generators are shut downwhen their output is low enough, and they will stop consumingpower.
Next come the level shifters. These shift the positive controlsignals coming into the driver towards negative voltage signalsusable to drive the DMOS (double-diffused MOS) in the reso-nant circuit. DMOS are a type of MOSFET that has an increasedtolerance for high voltages on the drain. The gate-source volt-age on the other hand has to remain at lower voltage in order forthe DMOS to not break. This is why we use two level shifters:one to drive the upper pDMOS going from -3V to ground and asecond one driving the lower nDMOS from -5V to -2V.
These level shifters power the DMOS in the resonant cir-cuit. The size of these DMOS should be chosen large enoughso that they can provide enough current to charge the HEMTgate quickly. Sizing them too large however enlarges their par-asitics, making it slower and less power efficient to drive theseDMOS.
Then comes the energy recovering element itself: the induc-tor. This also needs to be sized according to the desired switch-ing frequency. The larger the inductor, the more energy it canrecover but the slower the HEMT can switch. This not onlylowers the possible switching frequencies, but can also lowerthe efficiency of the total system by increasing the power lossover the HEMT. When the HEMT is switched slowly, its hardswitching losses increase. This means it dissipates a lot of powerin the time where the drain-source voltage is increasing but the(large) current still flows through the HEMT. This happens whenthe gate voltage is slightly above the switching voltage of theHEMT, or slightly above -3V for the used MGG1T0617T. Wecan conclude that a balance needs to be found in the sizing ofthis inductor: big enough to have enough energy recovery, butnot too large so we maintain sufficient switching speed.
As recovering elements we use schottky diodes. These have alow voltage drop and have very low recovery charges, meaningthey can react very quickly to changing currents.
IV. SIMULATION RESULTS
The efficiency improvement of driving the HEMT in a reso-nant way will now be tested. We do this by comparing the powerconsumption and efficiency of the resonant driver with a conven-tional direct driver. This direct driver is built by removing theresonant circuit in the previously built driver.
We will vary different design parameters and test their influ-ence on the power consumption and efficiency. The parameterswe will vary are the duty cycle D (0.2 - 0.9), switching periodTin (200 - 1000 ns), resonant inductance L (1 - 10 µH ), diodetypes (Schottky vs normal diodes), booster supply voltage Vb(1 - 50 V) and load impedance RL(5 - 500 Ω). Unless other-wise mentioned, the base parameters used in the simulations aregiven in table I
A. Base case
We start by simulating the driver with default parameters toestablish a base performance for our system. A waveform us-ing the default parameters can be seen in figure 3. We noticethat the real duty cycle is higher in both drivers: the HEMTstays on 50ns longer than we expect from the input duty cycle.
TABLE IDEFAULT VALUES FOR THE MAIN DESIGN PARAMETERS
Width clock buffer 200 µmCpump -3V generator 100 pFCpump -5V generator 400 pFSwitching period 500 ns
Pulse width to turn on 80 nsPulse width to turn off 85 ns
Duty Cycle 0.4nDMOS transistor width 200 µmpDMOS transistor width 400 µm
Resonant inductor 2 µ HInput voltage boost converter 3.3 V
output Rload 100 Ω
This is due to the non-linear parasitic capacitance of the usedHEMT. Once the HEMT is on, the parasitic capacitances aremuch larger. This results in most of the switching time beingspent above threshold voltage, for both turn-on and turn-off.
We see that output voltage, booster input power and boosteroutput power are about the same for both drivers : 5.9V, 400mWand 350mW respectively. The driver power consumption how-ever is lower in the resonant case: 68mW versus 78mW in thedirect driver. This results in an efficiency increase of 0.8%, from73.5% in direct drive to 74.3% with resonant drive.
Fig. 3. Waveform of a typical simulation with default parameter values.
B. Duty cyckle
The first parameter we change is the duty cycle, since thishas a big influence on the boost converter performance. We no-tice a few effects for both drivers. First, as expected, the outputvoltage goes up with higher duty cycles. Secondly, the effectiveduty cycle is larger than the theoretical, due to the HEMT stay-ing longer above threshold voltage when switching due to itsnon-linear parasitic capacitance. Lastly, varying the duty cycledoes not change the driver power consumption.
For input duty cycles below 0.5, the resonant driver is moreefficient: 70.5% versus 73% at a duty cycle of 0.3. At low dutycycles below 0.5 the output power is relatively low meaning thepower saving in the resonant driver account for a substantial ef-ficiency increase. For duty cycle above 0.5, the direct driver is
more efficient. This is because the direct driver switches faster,which results in less hard switching losses in the HEMT.
C. Input period
The second thing we vary is the input period. The result ofvarying this from 200ns - 1µs can be seen in figure 4. Twoclear trends are found. First off, the driver power consumptiongoes up with faster switching periods. This is normal, since theHEMT gate will have to be charged and discharged more of-ten, generating higher losses in the circuit. At an input periodof 200ns, a possible reduction of more than 50% (89.6mW ver-sus 197.4mW) was found. Secondly, we confirm that a higherresonance inductor leads to more energy recovery, especially atfaster switching speeds. At an input period of 600ns, an effi-ciency increase up to 5.5% was found: 74.8% efficiency in di-rect drive and 80.3% in resonant drive.
Fig. 4. Driver power versus switching period for different inductance values.L=0 is the direct drive scheme. duty cycle=0,5.
D. Resonant inductance
Third, we vary the value of the resonance inductor. Very lowvalues (≈ 1 µH) resemble the direct drive, both in behaviourand in power consumption. Higher inductance (≈ 5 µH) val-ues work as intended, and too high inductor values (≈ 10 µH)pose problems. These problems include the hard switching ef-fect discussed earlier. The higher the inductor, the better thedriver power consumption. This does not mean total efficiencyhowever, since very high inductor values lead to slower switch-ing speeds, and thus the GaN HEMT consuming more powerdue to this hard switching.
E. diode types
In the design we used schottky diodes, which are not availablein every technology. We now test the influence if we switchout the schottky diodes for standard pn-junctions, which have ahigher voltage drop of 0.6V.
The first schottky diode is found in the resonant circuit. Ifwe switch out this diode to a normal diode with a higher volt-age drop, the driver power consumption only increases by 1mW (56.36mW versus 57.34). This does not seem like a bigdeal. This is however simulated with a very simple diode model.When we use a more complicated model which also modelstransient diode behaviours like reverse recovery charge and -time, we see that a normal diode is not fast enough to work as a
recovery diode. In conclusion, the voltage drop of the diode isnot that important, but its transient behaviour is.
Secondly, we change out the flywheel schottky diode in theboost converter. Doing this leads to an efficiency decrease of5%: from 74.4% down to 69.2%. Since all the output current hasto go through this flywheel diode, the voltage drop has a largeinfluence on the total power loss and thus efficiency. In realitythe driver efficiency will be even worse since the reverse recov-ery time of normal diodes is much higher than those of schottkydiodes, resulting in even bigger losses. Some boost convertershave been shown that do not use flywheel diodes anymore toeliminate these losses.
F. boost converter supply voltage
The driver was designed for and needs a supply voltage of3.3V to work. The boost converter on the other hand can useany voltage it wants. We vary the voltage from 1 to 50V and testthe working of the driver.
We notice that the higher the input voltage, the higher theefficiency. This is because higher input voltages lead to higheroutput voltages, and thus more output power. This also meansthat the driver losses are a bigger portion of the total loss whenthe boost converter input voltage is low. This is why the resonantdriver is more efficient than the direct driver for input voltagesbelow 5V: the power saving in the resonant driver account fora substantial efficiency increase at low voltage. At 2V input,the efficiencies for direct and resonant drive are 53.3% and 50%respectively. At higher voltages, the booster power consumptioninfluences the efficiency the most. Due to the slower switchingspeed of the resonant driver, it has more hard switching lossesin the boost converter and its efficiency is lower. At 10V, theresonant driver has an efficiency of 86.29% and the direct driverone of 88.3%.
The second thing we note is that the driver works up to verylow voltages, but breaks down at very high voltages. At an in-put voltage of 50V for example, the boost converter only pumpsthe output to 60V. Some modifications to the circuit can be doneto let the driver work at these high voltages, albeit at some effi-ciency costs.
G. load impedance
Lastly, the default driver can effectively power loads from20Ω to 500Ω. Loads above 500Ω provide problems becausethey require very large output voltages. The same type of driveradjustments as for high input voltages can be made, since theorigin of the problem is the same: a large output voltage.
When we compare a resonant and direct driver for differentoutput loads, we see that the resonant driver is more efficient athigher output loads, while the direct driver is more efficient atlower load resistances. This once again boils down to the res-onant driver consuming less power but being more susceptibleto losses in the HEMT due to hard switching behaviour. Whenthe booster current is high for low output loads, the direct driveris more efficient due to the HEMT dissipating less power. Oncethe output load is 100Ω or higher, the output power becomeslower and the power driver consumption becomes more impor-tant,resulting in the resonant driver being more efficient, up to11% at an output load of 500Ω.
H. Best achieved results
H.1 Best absolute efficiency
The boost converter with the highest efficiency at an inputvoltage of 3.3V has an efficiency of 84% with a resonant driver,versus 82% in direct drive. This was in a boost converter switch-ing at 1MHz and a duty cycle of 0.5. The resonant inductor usedwas 5µH and the output load 100Ω. At this efficiency, we havean output voltage of 7V, more than two times the input voltage.
When this driver would be designed into real commercialproducts, the efficiency could be much higher. The goal of thisthesis was to test the efficiency increase of resonantly driving aGaN HEMT. We did not focus our efforts on creating a driverwith the best possible efficiency, since this was not the goal ofthis research.
H.2 Best efficiency increase
The strength of resonant drivers lies in their low driver powerconsumption. We tested the best efficiency increase of the res-onant driver in ideal circumstances. We found this to be at aninput voltage of 3.3V, switching period of 400ns, duty cycle of0.25, output load of 500Ω and inductance value of 5µH.
These parameters result in a direct driver with an efficiency of42.5% and a resonant driver with an efficiency of 59.5%: an ef-ficiency increase of 17%! The output voltage with the direct andresonant driver is respectively 6.25V and 7.85V. This proves thatthe resonant driver can deliver a significant efficiency improve-ment in certain applications.
V. CONCLUSION
The goal of this research was to test the possibility and/or ef-ficiency increase of driving a GaN D-mode HEMT in a resonantway. The simulation results show that a resonant driver is moreefficient than a direct driver. This result is robust under varyingparameter values. Resonant driving scheme is possible and im-proves efficiency by typically 1%, although improvements of upto 17% were demonstrated. However, a resonant driver sacri-fices driver simplicity and flexibility, and requires extra off-chipcomponents, making it more costly to produce. It will dependon the application if the benefits outweigh the costs.
The designed driver can be used for a very wide variety ofboost converter input voltages, but the driver itself can only bepowered by 3,3V. The load resistance can vary from 10Ω to500Ω, depending on the application. The designed driver canswitch frequencies up to 5 MHz. Bigger resonance inductor val-ues lead to more energy recovery, and thus lower driver powerconsumption.
ACKNOWLEDGMENTS
I would like to acknowledge the help of my supervisors, giv-ing great suggestions and coaching. I would also like to thankmy family for the never ending support, and my friends for giv-ing me energy and happiness in my life. Special thanks goesout to my dad and brother for the continued interest in this workthroughout the year, as well as reviewing it and giving sugges-tions.
REFERENCES[1] Gan O. Ambacher, J. Smart, J. R. Shealy, N. G. Weimann, K. Chu, M.
Murphy, W. J. Schaff, L. F. Eastman, R. Dimitrov, L. Wittmer, and etal., Two-dimensional electron gases induced by spontaneous and piezo-electric polarization charges in n- and ga-face algan/gan heterostructures,Journal of Applied Physics , vol. 85, no. 6, pp. 3222-3233, 1999. doi:10.1063/1.369664.
[2] Y. Chen, F. C. Lee, L. Amoroso, and H.-P. Wu, A resonant mosfet gatedriver with efficient energy recovery, IEEE Transactions on Power Elec-tronics, vol. 19, no. 2, Mar. 2004
[3] D. Matousek and L. Beran, Comparison of positive and negative dicksoncharge pump and fibonacci charge pump, 2017 International Conference onApplied Electronics (AE), 2017. doi: 10.23919/ae.2017.8053595.
[4] Astable multivibrator.[Online] http://www.electronics-tutorials.ws/ wave-forms/astable.html,
CONTENTS i
Contents
1 Introduction 1
2 Literature study 3
2.1 Power converters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
2.1.1 Types of power converters . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
2.1.2 Example power converter . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
2.2 HEMTs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
2.2.1 GaN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
2.2.2 AlGaN/GaN HEMTs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
2.2.3 HEMT usages . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
2.2.4 HEMT simulation model . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
2.3 HEMT drivers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
2.3.1 Cascode driven . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
2.3.2 Direct drive . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
2.4 Resonant drivers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
2.4.1 Circuit Explanation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
2.4.2 Circuit discussion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
2.5 Negative voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
2.5.1 Negative voltage generator . . . . . . . . . . . . . . . . . . . . . . . . . . 19
3 Driver components 22
3.1 Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
3.2 Clock generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
3.2.1 Ring Oscillator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
3.2.2 Schmit trigger oscillator . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
CONTENTS ii
3.2.3 Astable multivibrator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
3.2.4 Efficiency measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
3.3 Negative voltage generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
3.3.1 -3V generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
3.3.2 -5V generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
3.4 Negative level detector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
3.5 Level shifters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
3.6 Resonance circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
3.7 HEMT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
3.8 Boost converter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
3.9 Complete system . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42
4 Simulation results 46
4.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
4.2 Base case . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
4.3 Duty Cycle . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48
4.4 Switching period . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51
4.5 Resonant inductance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54
4.6 Diode types . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 55
4.7 Boost converter input voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 56
4.8 Load impedance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57
5 Discussion 61
5.1 Best possible boost converter efficiency . . . . . . . . . . . . . . . . . . . . . . . . 61
5.2 Best possible efficiency increase . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62
5.3 Possible improvements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63
5.4 Comparison direct drive versus resonant drive . . . . . . . . . . . . . . . . . . . . 64
5.5 Possible changes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65
6 Conclusion 67
6.1 Conclusions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 67
6.2 Suggestions for future work . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68
Appendix A Datasheet HEMT 72
CONTENTS iii
Appendix B Verilog-A code 78
B.1 Diode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 78
B.2 Shottky diode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 78
B.3 HEMT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 82
B.3.1 Drain-source current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 82
B.3.2 Capacitances . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 83
LIST OF FIGURES iv
List of Figures
2.1 DC/DC boost converter circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
2.2 The waveforms for continuous and discontinuous driving modes. The current in
continuous mode never drops to zero, while in discontinuous mode it does. . . . . 7
2.3 Typical AlGaN/GaN HEMT structure (not drawn to scale). . . . . . . . . . . . . 10
2.4 Typical band diagram for an AlGaN/GaN HEMT (not drawn to scale). . . . . . 11
2.5 Ideal characteristics Ids-Vds for an AlGaN/GaN at different gate voltages [6]. . . 11
2.6 Two different driving schemes for a D-mode HEMT device. . . . . . . . . . . . . 14
2.7 Resonant circuit for designed for high frequency PWM applications [18] . . . . . 16
2.8 Example implementation of 7-stage negative dickson charge pump. . . . . . . . . 20
3.1 Overview of all the building blocks of the driver. The blue arrows are external
control signals. Almost all blocks are connected to ground and the positive voltage
rail, but these connections aren’t shown. . . . . . . . . . . . . . . . . . . . . . . . 23
3.2 Circuit ring oscillator for 20 MHz clock. It contains 27 stages and delay-time
capacitors of 10pF between every stage. . . . . . . . . . . . . . . . . . . . . . . . 25
3.3 Circuits Schmitt trigger oscillator . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
3.4 Astable multivibrator circuit for 20 MHz clock. It also buffers both (complemen-
tary) clock outputs. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
3.5 -3V negative voltage generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
3.6 -5V negative voltage generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
3.7 Negative level detector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
3.8 Circuit differential opamp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
3.9 DC response opamp. Vdd is 3,3V. Vss and Vin+ are tied to ground and Vneg is
varied. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
3.10 Level shifter shifting +3,3V to -5V and 0V to -2V.Vdd=3,3V;Vss=-5V . . . . . . 35
LIST OF FIGURES v
3.11 Level shifter shifting +3,3V to 0V and 0V to -3V. Vdd=3,3V;Vss=-3V . . . . . . 36
3.12 The used resonant circuit. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
3.13 The average and standard deviation of critical parameters . . . . . . . . . . . . 40
3.14 The circuit of the boost converter . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
3.15 The complete circuit of the boost converter, including the driver. . . . . . . . . . 43
3.16 Waveform of the driver and boost converter with default parameters. . . . . . . . 45
4.1 Waveform of the default simulation. . . . . . . . . . . . . . . . . . . . . . . . . . 47
4.2 Driver power and efficiency versus input duty cycle. L=2µH. . . . . . . . . . . . 49
4.3 Driver power versus switching period for different inductance values. L=0 is the
direct drive scheme. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
4.4 Efficiency versus switching period for different inductance values. L=0 is the
direct drive scheme. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
4.5 Efficiency at different booster input voltages. . . . . . . . . . . . . . . . . . . . . 57
4.6 Segment of the driver waveform when Rload=1kΩ. . . . . . . . . . . . . . . . . . 59
LIST OF TABLES vi
List of Tables
3.1 Power consumption of the three different clock generators at 20 MHz clock frequency. 28
3.2 OR gate thruth table . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
3.3 Default values for all the design parameters . . . . . . . . . . . . . . . . . . . . . 44
4.1 Power consumptions for resonant and direct driver at default configuration. . . . 47
4.2 Power consumptions for varying duty cycle of the resonant driver. Tin=500ns . . 48
4.3 Power consumptions for varying duty cycle of the direct driver. Tin=500ns . . . 48
4.4 Power consumption for varying duty cycle of the Resonant driver. Tin=1µs . . . 49
4.5 Power consumptions for varying duty cycle of the direct driver. Tin=1 µs . . . . 49
4.6 Power consumptions for varying switching periods of the resonant driver. L=2
µH, duty cycle=0.5 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51
4.7 Power consumptions for varying switching periods of the resonant driver. L=5
µH, duty cycle=0.5 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51
4.8 Power consumptions for varying switching periods of the direct driver. duty
cycle=0.5 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51
4.9 Power consumptions for varying switching periods of the resonant driver. L=2
µH, duty cycle=0.3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52
4.10 Power consumptions for varying switching periods of the resonant driver. L=5
µH, duty cycle=0.3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52
4.11 Power consumptions for varying switching periods of the direct driver. duty
cycle=0.3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52
4.12 Power consumption for varying inductance values in the resonance driver. . . . . 54
4.13 Power consumption of the resonance driver with different diode configurations.
L=5µH . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 55
4.14 Power consumptions of the resonant driver for varying booster input voltages. . . 56
LIST OF TABLES vii
4.15 Power consumptions of the direct driver for varying booster input voltages. . . . 56
4.16 Power consumptions of the resonant driver for varying load impedance. . . . . . 58
4.17 Power consumptions of the direct driver for varying load impedance. . . . . . . . 58
5.1 Parameter values to achieve the highest possible efficiency. The parameters not
mentioned are default. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62
5.2 Parameter values to achieve the highest possible efficiency increase. The param-
eters not mentioned are default. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63
GLOSSARY viii
Glossary
AC/DC Alternating current and direct current.
MOSFET Metal-oxide-semiconductor field-effect transistor
GaN Gallium nitride
D-mode Depletion mode, or normally-on
HEMT High electron mobility transistor
SMPS Switched mode power supply
CMOS Complementary metal-oxide-semiconductor
2DEG Two-dimensional-electron-gas
HV/LV High voltage and low Voltage
DMOS Double-diffused metal-oxide-semiconductor
nDMOS N-type DMOS
pDMOS P-type DMOS
PWM Pulse-width modulation
Vcc Supply rail voltage
Vg (HEMT) gate voltage
IC Integrated Circuit
EMC Electromagnetic compatibility
INTRODUCTION 1
Chapter 1
Introduction
In modern day life, electronics is something you see everywhere. You wake up with an electrical
alarm clock, work on a computer, watch TV in the evening... All these electronic devices need
power, and this is what the little black boxes are for that we plug into the wall socket. Welcome
to the field of power electronics. It controls power coming from the input (e.g. power socket
or batteries) and converts it into something the electronic circuit can use. One of the most
important metrics in power electronics is the efficiency: the ratio of input power versus useful
output power. A device with bad efficiency uses extra energy which can lead to bad battery life
and excess heat generation.
In the field of power electronics, a big topic is power converters. These convert an input
voltage into a different output voltage. Some example of different outputs are a DC voltage, a
higher or lower voltage or a voltage that can draw a lot of current. Most modern day power
converters use some type of switching element. These types are called switched-mode power
supplies. Most often, the switching element is a power MOSFET, an insulated-gate bipolar
transistor (IGBT) or a thyristor.
In recent years, GaN D-mode HEMTs (Gallium Nitride depletion-mode High Electron Mo-
bility Transistors) have started receiving attention in research and products as the switching
element in power converters. They have high voltage capabilities, high switching speeds and
high electron carrier mobility, resulting in a low ON resistance. The high voltage capability lets
them be used in high-voltage environments and converters with high voltage outputs. The high
switching speeds and high electron carrier mobility on the other hand improve the efficiency
when it is used in a power converter.
The major disadvantage of these GaN HEMTs is that they are depletion mode devices. This
INTRODUCTION 2
means that they are normally-on at zero gate–source voltage. To turn this type of transistor off,
we have to apply a negative gate–source voltage. We can do this by either putting a negative
voltage on the gate of the HEMT, called direct drive, or we can raise the voltage at the source
and hence create a negative source-gate voltage, called cascode drive.
A different method used to improve the efficiency of power converters is driving the switching
element in a resonant way. This means using some sort of component (most often an inductor)
to recover some of the energy used for switching the switch element on and off. This method
has already been proven to be effective in silicium technology, but has not yet been tested for
GaN HEMTs.
This thesis aims to test the efficiency improvement of power converters when we combine
both technologies: driving a GaN HEMT with a resonant driver. We will test the efficiency of
a power converter by first driving it in the conventional way, and then we will test the resonant
driver. The efficiency will be tested for different switching frequencies (1-5 MHz), different load
resistances and different input voltages. A GaN HEMT can be driven in two ways: via a cascode
circuit or directly at its gate. Here, we will drive the HEMT directly at its gate.
This thesis is divided into six chapters. The first chapter is this introduction, stating the
goal and used technologies. Next, we explore what research has already been done, and what we
can use to build upon in chapter 2. We design and build the circuits in simulations in chapter
3. In chapter 4, the results of our simulations are given and explained. In chapter 5, we will
discuss the results in a broader scope. We will finish the thesis by summarizing our results and
giving a final conclusion as well as some directions for further research in chapter 6.
LITERATURE STUDY 3
Chapter 2
Literature study
The first step in designing and testing a new system is understanding what already exists and
what is already known. We do this by making a literature study. First we will discuss in
what type of applications this research can be used: switched-mode power supplies. Next, we
will discuss the used switching element, a GaN HEMT (Gallium nitride high electron mobility
transistor) and its main advantages and disadvantages. We also discuss how we will model
this component. Following, we take a look at the possible driving schemes, and its driving
circuits. Lastly, we take a look at possible negative voltage generators. This is because we want
to generate the required negative voltages ourself, instead of relying on an external negative
voltage power rail.
2.1 Power converters
A big topic in power electronics are power converters. These transform an initial AC or DC
voltage into a different AC or DC voltage, with the difference being in voltage level, frequency
or other properties. A brief discussion of some power converters is given below.
2.1.1 Types of power converters
DC to DC converters are primarily used in portable electronic devices such as cellular phones
and laptop computers, which are supplied with power from batteries primarily. Such electronic
devices often contain several sub-circuits, each with its own voltage level requirement different
from that supplied by the battery or an external supply (sometimes higher or lower than the
supply voltage). Additionally, the battery voltage declines as its stored energy is drained.
2.1 Power converters 4
Switched DC to DC converters offer a method to increase voltage from a partially lowered
battery voltage thereby saving space instead of using multiple batteries to accomplish the same
thing.
AC to AC transformers used for voltage conversion at mains frequencies of 50–60 Hz must
be large and heavy for powers exceeding a few watts. This makes them expensive, and they are
subject to energy losses in their windings and due to eddy currents in their cores. DC-to-DC
techniques that use transformers or inductors work at much higher frequencies, requiring only
much smaller, lighter, and cheaper wound components. Consequently these techniques are used
even where a mains transformer could be used. For example, for domestic electronic appliances
it is preferable to rectify mains voltage to DC, use switch-mode techniques to convert it to high-
frequency AC at the desired voltage, then, usually, rectify to DC. The entire complex circuit is
cheaper and more efficient than a simple mains transformer circuit of the same output.
A switched-mode power supply (SMPS) is an electronic power supply that incorporates a
switching regulator to convert electrical power efficiently. Like other power supplies, a SMPS
transfers power from a DC or AC source (often mains power) to DC loads while converting volt-
age and current characteristics. Unlike a linear power supply, the pass transistor of a switching-
mode supply continually switches between low-dissipation full-on and full-off states, and spends
very little time in the high-dissipation transitions, which minimizes wasted energy. Ideally, a
switched-mode power supply dissipates no power. Voltage regulation is achieved by varying the
ratio of on-to-off time, called duty cycle. In contrast, a linear power supply regulates the output
voltage by continually dissipating power in the pass transistor. The higher power conversion
efficiency is an important advantage of a switched-mode power supply. Switched-mode power
supplies may also be substantially smaller and lighter than a linear supply due to the smaller
transformer size and weight.
Fast rise and fall times of the semiconductor devices are required for efficiency. The shorter
the switch stays in its power-inefficient linear region, the higher the efficiency of the driver will
be. However, a downside of fast switching is that these fast transitions combine with layout
parasitic effects to make circuit design challenging. They can also give rise to EMC problems.
Faster devices also permit higher switching frequencies. This means the requirements for the
low-pass filter at the output of the power converter are less strict and hence can be made more
power-efficient and smaller/cheaper. The filter can use smaller components, which have less
parasitics and can be cheaper. If the components are small enough, we can even incorporate
2.1 Power converters 5
them on-chip. The low-pass filter at the output is necessary to transform the high-frequency
output to a DC voltage. A (small) voltage ripple will always be present on this ’DC’ output.
The higher efficiency of a switched-mode converter reduces the heatsinking needed, and in-
creases battery endurance of portable equipment. Efficiency has improved since the late 1980s
due to the use of power FETs, which are able to switch more efficiently with lower switching
losses at higher frequencies than power bipolar transistors, and use less complex drive cir-
cuitry. Another important improvement in DC-DC converters is replacing the flywheel diode
by synchronous rectification using a power FET, whose ”on resistance” is much lower, reducing
conduction losses. This improvement however is not used in this project, because it increases
the complexity of the circuit while not providing more information on the topic of a direct driver
versus resonant driver.
We can summarize that power converters are used in almost every piece of modern electronic
equipment. They use switching power semiconductor devices for both DC as well as AC con-
versions. These devices need to be able to handle large voltages and/or currents. Because of
efficiency reasons, they need to be able to switch fast between their on and off state. A last re-
quirement is that their on-resistance should be as low as possible to minimize power dissipation
in the device itself.
2.1.2 Example power converter
One of the simplest power converters is a dc-dc boost converter. Battery power systems often
stack cells in series to achieve higher voltage. However, sufficient stacking of cells is not possible
in many high voltage applications due to lack of space. Boost converters can increase the voltage
and reduce the number of cells. Two battery-powered applications that use boost converters are
electric vehicles and lighting systems.
In this thesis, we test the efficiency increase of driving a power converter with a GaN HEMT
in a resonant way, by incorporating it in a dc-dc boost converter. The circuit of this boost
converter can be seen in figure 2.1. Its advantages are that an (arbitrary) higher output voltage
than the input voltage can be obtained. Secondly, the switch can be driven with respect to
ground as compared to high side or isolated drive required for buck or buck-boost converters.
Lastly, the input current is continuous (no discontinuities) which means it is easy to filter and
meet EMC requirements.
Of course, the circuit has some disadvantages. A large output capacitor is required to reduce
2.1 Power converters 6
the voltage ripple as the feeding current is pulsating. There is also a slower transient response
and lastly the need for feedback loop compensation in order to achieve a steady voltage.
Figure 2.1: DC/DC boost converter circuit
The key principle that drives the boost converter is the tendency of an inductor to resist
changes in current by creating and destroying a magnetic field. We make use of this tendency
by constantly switching it ON and OFF. When the switch is closed, current flows through the
inductor L in clockwise direction and the inductor stores some energy by generating a magnetic
field. The voltage over the inductor is positive and hence the amount of current flowing through
it will increase.
When the switch is opened, the current running through the inductor cannot go via the
HEMT anymore. This current will now be forced to go through the diode into the output
capacitor and load. The magnetic field previously created will be destroyed to maintain the
current towards the load. Because the output voltage is higher than the input voltage, the
voltage over the inductor will be negative and the current will decrease. Since there is now a
current flowing into the output capacitor Cout, the output voltage will increase.
When we now close the switch again, the voltage over the diode will become negative again
and it will stop conducting. The process of charging the inductor starts anew.
Meanwhile, the output capacitor powers the load, but loses charge while doing so. In order
for the output voltage Vo to remain steady over this period, we need the output capacitor to be
rather large. The load impedance experiences a DC voltage over the complete cycle, with some
ripple present due to the constant charging and discharging of the output capacitor.
2.1 Power converters 7
(a) Continuous mode (b) Discontinuous mode
Figure 2.2: The waveforms for continuous and discontinuous driving modes. The current in
continuous mode never drops to zero, while in discontinuous mode it does.
There are two modes the boost converter can operate in: continuous and discontinuous
mode. Their respective waveforms can be seen in figure 2.2. When a boost converter operates in
continuous mode, the current through the inductor never falls to zero. On the other hand, if the
ripple amplitude of the current is too high, the inductor may be completely discharged before
the end of a whole cycle. This commonly occurs under light loads. In this case, the current
through the inductor falls to zero during part of the period. Although the difference is slight, it
has a strong effect on the output voltage equation.
In continuous mode, the output voltage is affected by the duty cycle. The output voltage
can be calculated via equation (2.1). In this equation, D is the duty cycle and ranges from
0 (S is never on) to 1 (S is always on). In our example waveforms the duty cycle is equal to
0,5. This equation supposes that all component are ideal. The output voltage equation can
theoretically go to infinite. This is ofcourse not possible in practice, since our components are
not ideal (power loss in switch, power loss over diode, limited current in the diode/FET...).
Discontinuous mode on the other hand has a more complicated output voltage equation
(2.2). The output voltage gain not only depends on the duty cycle (D), but also on the inductor
value (L), the input voltage (Vi), the commutation period (T) and the output current (Io). Once
again this equation supposes ideal components.
Which of the two modes we have depends on a few factors, like output current (in other words
the load impedance), duty cycle and maximum currents through the elements. Continuous mode
2.2 HEMTs 8
has a steadier output voltage and input current, but discontinuous mode is more power efficient,
since less current flows through the system and as a result less power is dissipated in the different
components.
VoVi
=1
1 −D (2.1)
VoVi
= 1 +ViD
2T
2LI0(2.2)
2.2 HEMTs
A HEMT is a high-electron-mobility transistor [1] , also known as heterostructure FET (HFET)
or modulation-doped FET (MODFET). It is a field-effect transistor incorporating a junction
between two materials with different band gaps (i.e. a heterojunction) as the channel instead of
a doped region (as is generally the case for MOSFET).
2.2.1 GaN
Since the invention of the metal-oxide-semiconductor field-effect-transistor (MOS-FET) in 1959,
the semiconductor industry for electronics has been dominated by silicon (Si). This is because
of the ease and cost of growing Silicium-oxide which enables ’easy’ production of complementary
metal-oxide-semiconductor (CMOS) process which has driven electronics forward to where we are
today. Si is however a low band gap material (1.1 eV) and hence not very suitable (although used)
in power electronics. Because of this, new materials are investigated as potential replacements,
such as gallium nitride (GaN), silicon carbide (SiC) and diamond.
GaN based transistors were developed in the early to mid 1990’s and have been extensively
researched and used for high-power high-frequency applications. In more recent years, it has
been rediscovered for use in power-electronics as high-voltage power switches.
GaN is a wide band gap semiconductor (band gap energy Eg=3.44eV) similar to diamond
and SiC. This wide band gap generally translates into an ability to handle high internal electric
fields before electronic breakdown occurs [2]. GaN is particularly interesting because of its
ability to form heterojunctions with wider band gap semiconductors such as aluminium gallium
nitride or aluminium nitride (up to 6.2 eV). This heterojunctions results in the forming of a 2-
dimensional-electron-gas (2DEG) at the interface due to large polarisation in the material which
provides a highly dense, majority carrier channel with a large electron mobility. This results in
2.2 HEMTs 9
a device which is capable of operating as a high-voltage (HV) power switch or high-frequency
power amplifier.
Due to the 2DEG, GaN offers a much larger current density than other materials. This
makes it so the gate can be smaller, resulting in lower gate capacitance. This also means there
is less charge needed for switching, resulting in lower switching losses. Of course, since the GaN
material is more expensive than Si, and the chip production process is much more optimized for
Si wafers, the price of GaN HEMTs is higher than that of Si power FETs.
Because the conduction happens via the 2DEG located in the undoped GaN layer, there is
a very low source-drain resistance. There are few to no lattice deformations and no impurities
to hinder the flow of electrons, creating a very low ON-resistance.
The advantages of using GaN as material can be summarized as follows [2]:
1. GaN has a higher breakdown voltage. The critical electric field is around 300V/µm (vs
Si: 3V/µm [3]), meaning that for electrodes on GaN with a spacing of 1 µm, a theoretical
bias voltage of around 300V could be applied without material breakdown.
2. They have a lower on-state resistance. AlGaN/GaN HEMTs display on resistances of
1mΩcm2 compared to an on-resistance 100mΩcm2 for Si. This leads to lower conduction
losses in the power transistor, improving the efficiency of power converters and reducing
the heat generation.
3. A higher current density due to the 2DEG lets the gate be smaller, resulting in lower gate
capacitance. This means the HEMT can be switched with less gate charges and hence
faster and more efficiently.
4. GaN can withstand higher temperatures. Devices have been shown to work beyond 300°C,
which leads to a lower need for cooling systems and large heat sinks.
2.2.2 AlGaN/GaN HEMTs
Many high power applications use gallium nitride based electronic devices in the form of high
electron mobility transistors (HEMTs). HEMTs will be the type of device used in this thesis
and this section will provide some details on their structure and properties.
A typical GaN based HEMT structure is shown in figure 2.3. This structure is typically grown
by metal organic chemical vapour deposition (MOCVD) or molecular beam epitaxy (MBE).
2.2 HEMTs 10
If a wider band gap material is grown on top of this layer, a heterojunction forms where
electrons can be confined into a quantum well forming a two dimensional electron gas (2DEG).
In this quantum well, electrons are able to move very easily and their mobility can be up to
2000cm2/V s [4]. The wider band gap materials commonly used with the GaN are the semi-
conductor alloy aluminium gallium nitride (AlxGa1−xN), aluminium nitride (AlN) and indium
aluminium nitride (InxAl1−xN) [5]. The most popular of these options is the AlxGa1−xN , and
uses an Al content (x value) of around 20-30%.
The structure shown in figure 2.3 includes some extra layers, such as a thick foreign substrate,
a bonding layer and a cap layer. The substrate layer is used due to the difficulty and cost in
growing native GaN. The typical materials are SiC, sapphire or Si. To reduce thermal stress
and latice mismatch with the GaN layer, a thin nucleation layer of AlN is grown on top of the
substrate. Then the GaN layer comes, on which a wide band-gap layer is put (AlGaN). On
top of the AlGaN barrier a GaN cap layer is grown which helps to reduce gate leakage currents
compared to devices without this cap layer. This layer works by increasing the effective Schottky
barrier height. This structure, in contrast to other HEMTs, does not require impurity doping.
Figure 2.3: Typical AlGaN/GaN HEMT structure (not drawn to scale).
In figure 2.4, the band diagram of the AlGaN/GaN heterostructure is shown. It shows the
wider band gap AlGaN to the left hand side and the narrower band gap GaN on the right. The
difference in conduction band energies at the interface of the materials results in a conduction
band offset ∆EC and a quantum well is formed at the material interface where the electrons
which make up the 2DEG will occupy the conduction band due to preferable (lower) energy.
2.2 HEMTs 11
The 2DEG is typically very narrow, and this is the reason why it is called two dimensional.
Figure 2.4: Typical band diagram for an AlGaN/GaN HEMT (not drawn to scale).
Figure 2.5: Ideal characteristics Ids-Vds for an AlGaN/GaN at different gate voltages [6].
Due to the presence of this 2DEG, we have a normally-on device, or a depletion mode FET
(D-mode FET). This means we have to apply a negative gate-source voltage in order to turn
the HEMT off. An example of the I-V characteristic of a GaN HEMT is given in figure 2.5.
This D-mode device is more difficult to work with than a standard enhancment mode device
(e-mode), such as conventional CMOS technology [7]. Because of this, there has been a push to
devellop e-mode GaN HEMTs. While high-voltage (HV) enhancement mode (normally OFF)
GaN devices have been demonstrated, depletion-mode GaN HEMTs are typically superior in
intrinsic performance. Additional process steps such as the recessed-gate technique and fluorine-
2.2 HEMTs 12
based plasma treatment are required to obtain enhancement-mode GaN HEMTs, which increase
the cost of fabrication and impact the device performance.
2.2.3 HEMT usages
HEMTs have long been used in radio-frequency application because they have high gain, which
makes them useful as amplifiers at high switching speeds. This is achieved because the main
charge carriers in HEMTs are majority carriers, and minority carriers are not significantly in-
volved. Extremely low noise values are achieved because the current variation in these devices
is low compared to other FETs, as a result of the current travelling through undoped material.
However, in recent years the HEMT has been rediscovered in the field of power electronics.
Here, the switched-mode power supplies utilize a fast switching HV FET. Thus a need arises
for more power efficient switching drivers. In DC-DC converters, this switching FET is followed
by a lowpass filter in order to transform the output to DC. The higher the switching frequency,
the more leniency you have on this filter and the smaller you can size your components. Smaller
components are often more efficient as well, with less losses and heat generation. This results
in smaller, cheaper and more powerful electronics.
Because a HEMT has such exceptional carrier mobility and switching speed, we can minimize
the gate size for the same current density compared to a normal FET, and hence minimize the
gate capacitance. This means we have less switching losses, since this loss is equal to equation
(2.3). This also means we can switch faster with the same amount of gate current.
Pgate = QgVGfS (2.3)
2.2.4 HEMT simulation model
This thesis uses simulations for the development of the driver, as well as the efficiency measure-
ments of the different drivers. There is a lot of research being done to accurately simulate GaN
HEMTs, modelling all the charges moving and accumulating inside the material [8], [9], [10],
[11], [12]. However, this research is still ongoing, and an accurate model needs both an accurate
measurements of the device itself, as well as significant computation time.
Because of these two downsides, we prefer to make a simulation model specific for one
HEMT. We choose the GaN HEMT MGG1T0617T, because it suits the needs for this project.
We model most of the properties of this GaN HEMT given in its datasheet, see appendix A
2.3 HEMT drivers 13
[13]. Specifically, we model the drain-source current which is dependant on the gate-source and
drain-source voltage. We also model the voltage-dependant Cds, Cgd and Cgs, since these have
a very big influence on the switching performance of our HEMT. In this model, we neglect the
gate current leakage, which is very small anyway. We also neglect the gate resistance. This
could have an effect, but since the current needed to charge/discharge the gate is quite small,
this effect will be minimal.
Guidelines how to model a voltage-dependant capacitor is given in [14]. Extra care needs to
be taken that you conserve the charge on the component. In other words, the charge function
is continuous, and the current through the component is dqdt .
To model this component, we use the Verilog-A language. Verilog-A is an industry standard
modeling language for analog circuits. It is the continuous-time subset of Verilog-AMS.
2.3 HEMT drivers
In this thesis, we want to be able to drive a depletion-mode HEMT. This means the device is
normally on, and needs a negative gate-source voltage to turn off. As a result, we cannot use the
conventional drivers we use for silicium. There are currently two major ways to drive a HEMT:
either directly via its gate or via a cascode configuration.
2.3.1 Cascode driven
The most popular way of driving a normally ON device is via the cascode-drive structure, shown
in figure 2.6a. A LV (Low Voltage), high current Si MOSFET is connected in series with the HV
(High Voltage) GaN HEMT, and the gate of the HEMT is tied to the source of the MOSFET.
This way, the high voltage is positioned over the HEMT instead of the Si MOSFET, and a
MOSFET with low on-resistance can be used instead of the HV MOSFETs, which have a higher
on-resistance.
The cascode works by elevating the source voltage of the HEMT, so that the gate-source
voltage becomes negative, and the HEMT turns off. The cascode device now has a normally-off
characteristic. A conventional MOSFET driver can be used to drive the LV MOSFET at high
speeds.
The cascade mode has a major disadvantage however. It has higher switching losses due to
the need to charge both the Cgs of the HEMT as well as the Coss of the LV MOSFET, and the
Coss of the LV MOSFET is quite large, because its needs a large gate. Since it is made in Si
2.3 HEMT drivers 14
technology, the gate area has to be big because the large current flows through this MOSFET,
so to achieve a low RON and have efficient operation we need a big gate. Since the goal of this
thesis is to minimize driving power consumption, we will not use this driving scheme.
(a) Cascode HEMT driver (b) Direct HEMT driver
Figure 2.6: Two different driving schemes for a D-mode HEMT device.
2.3.2 Direct drive
The other commonly used way to drive a HEMT is by driving its gate directly with a negative
voltage, shown in figure 2.6b. The major disadvantage of this way is that we need to apply
a negative voltage to the gate. This voltage can be supplied via an external negative supply
rail or generated on-chip. The proposed driver has negative voltage generators, implemented as
dickson charge pumps.
Because the direct-drive scheme is always on, we also add an extra LV MOSFET in cascode
with the HEMT. This MOSFET is there to be able to shut the HEMT off for longer times
without consuming any power, and let the driver do a startup sequence before the HEMT starts
conducting. An UVLO (under voltage lockout) can also be incorporated here. This MOSFET
does not need to switch quickly or often, since its only function is to turn the HEMT switching
operation on or off. This means there is no power loss due to switching. Some small conduction
losses due to the (small) Ron introduced by this MOSFET still remain present.
The cascode and direct drive schemes were previously demonstrated for SiC junction field-
effect transistors (JFETs), which are also normally ON devices. It was concluded that the
cascode driving scheme has larger switching losses, due to the driving of the cascode LV MOSFET
[15], [16]. The cascode driving is also an indirect driving scheme, which offers less controllability
2.4 Resonant drivers 15
over the device switching behaviour.
2.4 Resonant drivers
High power density applications require converters operating at high frequencies. Therefore, the
size of the energy storage elements can be reduced and the losses associated with the equivalent
series resistance of the capacitor and inductor decrease. At higher frequencies however, gate
driver power consumption also increases considerably. The power consumption for a conventional
gate driver is dissipated almost completely via the gate resistance.
With the increase in operating frequency, the gate drive power loss becomes significant
enough to affect the efficiency of the converter. This loss can be reduced by introducing an
inductor which helps in either recovery or recycling of energy. Many authors have proposed
different techniques of energy recovery and energy recycling using the principle of resonance. A
detailed study of different resonant drivers for MOSFETS is given in [17].
Since our goal is low power consumption, we want a circuit that minimizes this as much as
possible. The second thing to take into account is the circuit complexity. At the input of our
circuit we only have positive voltages, but in order to turn the HEMT off we need (switching)
negative voltages. Because of this, we need a level shifter for each switching voltage. This means
we need to add a lot of extra complexity for each extra MOSFET in the original circuit.
With this caveat in mind, we chose the circuit depicted in figure 2.7a. This circuit only uses
two MOSFETS, two diodes and one inductor. It has the advantage of recovering the energy and
the removal of cross conduction losses. It has the disadvantage of only clamping the gate during
energy recovery. In some circuits, it can also cause unreliable ringing. To accomodate for this
ringing, we will generate a negative voltage sufficiently below the cutoff point of our HEMT.
The working of this circuit will now be discussed in more detail.
2.4 Resonant drivers 16
2.4.1 Circuit Explanation
(a) Circuit(b) Ideal waveforms when diode forward drop is ne-
glected and M1’s gate capacitance is approximated
as linear
Figure 2.7: Resonant circuit for designed for high frequency PWM applications [18]
In this circuit, a complementary driving pair (MDR1 and MDR2) is inherited from the conven-
tional driver, an inductor (LR) is inserted as the basic resonant element, and two more diodes
(DDR1 and DDR2 ) are designed to clamp and recover the driving energy. The timing of the
circuit is set to cycle the inductor current within each driving transition so that the circuit per-
formance will have no dependence on the operational duty cycle. In addition, when the diodes
recover the driving energy an equivalent low impedance path is provided.
The working of the circuit can be best explained with the help of the ideal waveforms given
in figure 2.7b. We begin with VGS M1 = 0 (t < t1) where both mosfets are off and the inductor
current is zero. At time t1, MDR1 is turned on and a voltage step appears at the source of
MDR1. Responding to this voltage, the inductor current ILR and the capacitor voltage VGS M1
both start to rise, until a time t2 when VGS M1 = VDD and ILR = IPeak. If the quality factor
Q of the resonant circuit is high enough, IPeak and the rise time tr (= t2 − t1) can be easilyestimated.
IPeak =VDDZ0
= VDD ·√CG M1LR
(2.4)
2.4 Resonant drivers 17
tr =π
2ω0=π
2·√LR · CG M1 (2.5)
where CG M1 is the equivalent gate capacitance of M1, Z0 the characteristic impedance of
the resonant circuit and ω0 the resonant frequency.
During the period between t2 and t3, VGS M1 is clamped at VDD by diode DDR1 and iLR flows
freewheeling along MDR1, LR and DDR1. This freewheeling phase should be as short as possible,
since conduction losses exist over all these elements and thus enlarges the power consumption
of the driver.
Once we arrive at time t3, MDR1 is turned off and the energy recovery process starts: the
inductor current turns on the body diode of MDR2 and flows through the path of MDR2 - LR -
DDR1 - VDD. With a constant voltage VDD across LR, the inductor current diminishes linearly
and the recovery time trec (=t4 - t3) = is simply
trec =LR · IPeakVDD
=√LR · CG M1 (2.6)
Because the gate voltage VGS M1 is at the right value, the driving circuit does nothing from
t4 to t5. When we now want to turn the switch off again, we need to discharge this gate voltage.
We do this by turning on MDR2. This imposes a ’negative’ voltage over LR and a current starts
flowing from CG M1 through LR and MDR2 to ground. Once the voltage is back at ground level
(at time t5, the current freewheels along DDR2, LRandMDR2. In reality, the inductor current iLR
will diminish a bit because of voltage drops over DDR2 and MDR2. Lastly, the recovery process
starts again when we shut off MDR2. From t7 - t8, charge is pumped from the low-voltage ground
to the high-voltage source VDD. Now the cycle can start anew
This description explains the operation of the circuit in terms of voltages and currents. We
can also explain the operation in terms of energy processing. From time t1 to t2, energy is
transferred from the power source VDD to the resonant inductor (0.5LRI2Peak) and the gate
capacitor (0.5CG M1V2DD). The subsequent stage (t2 to t3) freewheels the inductor energy, or in
other words takes power from VDD and moves it back to VDD. Lastly, from t3 to t4 the energy
stored in the inductor is returned to the power source.
In the discharging state, the potential energy present on the gate capacitor (0.5CG M1V2DD)
is now transferred to the inductor (discharging the gate capacitor and shutting off the switch)
at time t5 to t6. When we now start the recovery process this energy stored in the inductor is
finally returned to the power source.
2.5 Negative voltage 18
2.4.2 Circuit discussion
Theoretically, this circuit has zero power usage compared to the original power usage shown in
equation (2.3). In practice however, dissipation exists in every component and the power usage
is greater than zero. The dissipation is calculated theoretically in [18]. In this thesis, we will
adapt the circuit so it can drive a D-mode HEMT and simulate this so we can measure the
power usage/efficiency increase quantitatively.
A second advantage of this circuit is that the voltages are clamped via the diodes. This
means we have immunity to false triggering and we do not need extra ’safety circuitry’ to
protect against voltage spikes. The gate voltage will always reside between ground and source
(plus diode voltage).
A third advantage is that this circuit can handle (almost) any duty cycle, and a wide range
of frequencies, depending on the values of LR and CG M1 (because of tr and trec). A higher
inductance value has lower conduction losses (lower IPeak => lower I·V power consumption)but a lower possible operating frequency as well.
Of course the circuit has disadvantages too. The first being that an (ideally large) inductance
is needed. Inductors take up a lot of space (and are hence costly) to incorporate in an IC. An
external inductor will probably be needed. However, this external inductor takes up a lot of
space (landing pads + external connection) and brings a lot of extra parasitics with it, and
hence adds to the gate resistance. Secondly, the forward voltage drop of the clamping diodes
dissipate a lot of power in the freewheeling and recovery stage. We can lower these losses with
the use of Schottky diodes, if their leakage current does not cause a substantial Vgs drift at the
operating frequency. Lastly, the gate voltage Vgs is floating from t4 to t5 and t8 to t1. It is
clamped between VDD and ground but not connected to a specific voltage source. When there
is some leakage current, the exact value can change during this period of time. If this period is
too long, or in other words the switching frequency is too low, the switch could shut off earlier
than expected.
2.5 Negative voltage
As discussed earlier in section 2.3, we need a negative voltage to shut the HEMT off. We can
either bring this negative voltage in externally via a negative voltage supply rail, or generate
this voltage ourself. In this thesis, we choose to generate this voltage on-chip.
2.5 Negative voltage 19
A big problem in working with negative voltages on a chip is the possibility of charge leaking
from ground to the negative voltage via the body of our MOSFETs in the driving circuit. If
the source (or drain) has a low enough voltage, the parasitic source-bulk (or drain-bulk) p-n
junction can be forward biased and conduct a lot of charge. In order to prevent these losses,
we have to put the bulk of our chip at the lowest potential, so the lowest voltage. This also
means special care has to be taken in order to not destroy the MOSFETs due to large voltage
differences between its terminals.
In this thesis we work with the Intelligent Interface Technology I3T80 process technology
[19]. This technology has a maximum voltage of 3.6V over the terminals of the LV MOSFETs.
However, MOSFETs with floating body are also present, so we can still provide positive logic if
we float the body of the respective MOSFETs at e.g. ground. In short, care has to be taken that
a technology is chosen in which the MOSFETs can either handle big enough voltages between
their terminals without breaking, or have the option of floating the body of the MOSFETs and
taking special care to never exceed the maximum allowed voltage between terminals.
The I3T80 technology also has DMOS (double-diffused metal–oxide–semiconductor) present,
which has a higher allowed voltage between drain-gate, drain-source and drain-body. They can
be used when we want to sweep the voltage (for example at the HEMT gate) over more than
3,6V. The downside of these DMOS is a larger Ron. This results in either more conduction losses
or a larger gate, and thus more switching losses and more chip area needed.
2.5.1 Negative voltage generator
The classical DC/DC converters are based on inductors or transformers. These converters are
suitable for a high output power, but are too large for our application. Charge pumps are
a sufficient alternative to classical DC/DC converters for a lower output power and a smaller
dimension. A well-known variant of a charge pump is the Dickson charge pump. The output
to input voltage ratio is directly proportional to the number of Dickson charge pump stages.
The Fibonacci charge pump is a suitable alternative to the Dickson charge pump especially for
a higher output to input voltage ratio [20].
2.5 Negative voltage 20
Figure 2.8: Example implementation of 7-stage negative dickson charge pump.
An example negative dickson charge pump is given in figure 2.8. The circuit works as
follows: it first inverses and buffers the clock, creating signals clka and clkb. Now we will look
at the voltage over C1 during the clock transition. Suppose the voltage is initially 0V above
C1, and the clock clka transitions from 3V to 0V. Since the voltage over the capacitor cannot
change drastically, the voltage above C1 lowers as well by 3V, so to -3V. Diode D2 will start
conducting, until the voltage is distributed equally between C1 and C2, or a voltage of -1.5V is
at both terminals of D2 (assuming C1=C2 and ideal components).
In the next step, clka goes up again from 0V to 3V, and the voltage ’above’ C1 goes to 1.5V.
This activates D1, discharging the voltage until it is back to 0V, the same as before. However,
in this step clkb goes from 3V to 0V, lowering the voltage above C2 from -1.5V to -4.5V. This
activates D3, and again charge flows until both C2 and C3 are at the same voltage.
This process keeps repeating until eventually the node above C1 switches between 0 and -3V
and cannot pump charge anymore. The final stage is diode D8 with a load capacitor, changing
the switching negative voltage into a continuous negative DC voltage.
By constantly switching the clocks on and off, charge gets pumped towards the ground, and
a negative voltage is generated. Some voltage drops over the diodes is unavoidable, but we can
minimize this by using shottky diodes with a low voltage drop.
The biggest design parameter these voltage generators have are the pumping capacitor size,
the amount of stages and the clock frequency at which they pump their charge. The larger the
capacitors are, the more current the output can draw, or the more current loss is possible in the
following circuit, being the HEMT and the resonance circuit. Adding more stages to the pump
results in a lower voltage being achievable, and at higher voltages there is more current possible
at the output. Adding more stages has the downside of more steps where power can be lost, so a
2.5 Negative voltage 21
lower efficiency as a result. Lastly, increasing the clock frequency lets more charges be pumped
towards the load, so a higher output current is possible.
A second design factor is the output storage capacitor. Increasing this makes it so the output
voltage is steadier, but the start-up time will be larger as well. A larger output capacitor has
the downside that it takes up more space on a chip. It can also lead to a larger leakage current.
DRIVER COMPONENTS 22
Chapter 3
Driver components
We test the impact of the resonance driving on the performance by making use of simulations.
The used simulation program is Cadence. The used circuit simulator is Spectre: a SPICE-class
circuit simulator. It provides the basic SPICE analyses and component models. It also s
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