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8/20/2019 A Dual-Polarized Planar-Array Antenna for S-Band and X-Band Airborne Applications-5aa
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A Dual-Polarized Planar-Array Antenna for
S-Band and X-Band
Airborne
Applications
S
H
Y R
2
, a K C
1
Department of Electrical and Computer Engineering, Texas A M University
College Station, TX 77843-3128, USA
Tel: +1 (979) 845-5285; E-mail: shihhsun@ece.tamu.edu
21ntelligent Automation, Inc.
Rockville, MD 20855, USA
Abstract
A new dual-frequency dual-polarized array antenna for airborne applications is presented in this paper. Two planar arrays
with thin substrates (RIT Ouroid 5880 substrate, with
e;
= 2.2 and a thickness of 0.13 mm) are integrated to provide
simultaneous operation at S band (3 GHz) and X band (10 GHz). Each 3 GHz antenna element is a large rectangular ring
resonator antenna, and has a 9.5 dBi gain that is about 3 dB higher than the gain of an ordinary ring antenna. The 10 GHz
antenna elements are circular patches. They are combined to form the array with a gain of 18.3 dBi, using a series-fed
structure to save the space of the feeding line network. The ultra-thin array can be easily placed on an aircraft's fuselage, due
to its lightweight and conformal structure. It will be useful for wireless communication, radar, remote sensing, and surveillance
applications.
Keywords: Antenna arrays; microstrip arrays; microwave antenna arrays; aircraft antennas; planar arrays; polarization
1. Introduction
M
odern wireless communication systems demand low-profile,
lightweight, and relatively inexpensive antennas [1]. There
has been increasing interest in the development of dual-polarized
antennas/arrays, due to many advantages in performance improve
ment for wireless communication and radar systems. Dual
polarization avoids the requirement
of
precise alignment needed in
single-polarized systems. Dual-polarized operation can also pro
vide more information for radar systems, can increase the isolation
between the transmitting and receiving signals
of
transceivers and
transponders, and can double the capacity of communication sys
tems by means of frequency reuse [2, 3]. In simultaneous transmit
ting-receiving applications, dual-polarized antennas with two-port
connections offer an alternative to the commonly used bulky
circulator, or separate transmitting and receiving antennas. Further
more, dual-polarized antennas can provide polarization diversity,
which prevents the system's performance degradation due to multi
path fading in complex propagation environments [4]. Compared
with space diversity, the polarization-diversity technique has the
advantage of reducing the number and size of the antenna elements
in the system.
To create dual polarization, the antenna element has to be fed
at two orthogonal points or edges, such that two degenerate reso
nant modes can be excited for the orthogonal polarizations, i.e.,
vertical and horizontal polarizations. Design techniques of dual
polarized antennas can be classified into three main categories. The
first is to make an orthogonal arrangement
of
the radiation ele-
ments for two polarizations [5, 6]. The second is to properly
choose stacked structures of the radiating elements [7, 8]. The third
is to use special feeding techniques [9, 10]. In general, symmetric
structures tend to give better dual-polarization performance. At
higher frequencies, a dielectric-resonator antenna can be used to
reduce the metallic loss of the patch [11].
The dual-polarized antenna elements can be assembled to
construct a high-gain array. Although many dual-polarized anten
nas have been proposed, not all of them are good candidates for
array design, due to their complex structures and feeding-line net
works. Many of them are bulky and heavy, and not suitable for air
borne applications. On the other hand, isolation is one
of
the
important parameters to be considered in dual-polarized array
design. Most reported dual-polarized arrays achieve at least 20 dB
isolation [12-15]. To minimize the coupling between the feed-line
networks, a proximity/aperture-coupling structure can be applied to
prevent radiation due to the microstrip line and radiator from
degrading the polarization performance. The multilayer structure
can also be used to enhance the isolation, in which isolations of
better than 20-30 dB can be obtained with more-complicated struc
tures. However, most dual-polarized antenna designs will result in
a bulky array that is not suitable for use in aircraft, airships, or
unmanned aerial vehicles (DAVs).
In many airborne applications, an array antenna should have
good isolation, high efficiency, and ease of integration with the
aerial vehicle. A simple feeding-line network with lower loss and
high isolation is generally desired. Microstrip series-fed arrays
70
ISSN
1045-9243120091 25
©2009IEEE
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have been shown to have a structure that enhances the antenna's
efficiency [1]. This is because the array feeding-line length is
significantly reduced, compared to the conventional corporate feed
ing-line network. Such arrays can be either traveling-wave or reso
nant arrays. A planar structure with a thin and flexible substrate is
a good choice, because it will not disturb the appearance
of
the air
craft, and can be easily integrated with electronic devices for signal
processing.
In this paper, a dual-frequency dual-polarized array antenna is
presented for airborne antenna applications. A multilayer structure
is adopted for dual-band operation. The antenna arrays for the two
frequencies are separated on different layers. To reduce the array's
volume and weight, a series-fed network is used. An ultra-thin sub
strate is chosen in order to make the array conformal, and the array
can be easily placed on an aircraft's fuselage, or inside the aircraft.
The parameters affecting the array's characteristics are discussed,
and the measured return losses, radiation gains, and array patterns
are presented.
Two RTlDuroid 5880 substrates (61 =63 =2.2 and a foam
layer 6 2 = 1.06)
form
the multilayer structure. The thicknesses
of
the substrates and h
2
) are both only 0.13 rom (5 mil). These
ultra-thin and flexible substrates make it possible for the array to
be easily attached onto the aircraf t' s fuselage, or installed inside
the a ircraft . The foam layer has a th ickness of
h
2
=1.6mm. The
dimensions of the array were opt imized by using the full-wave
electromagnetic simulator, I [.
2 1 X Band Antenna and Subarray
The X-band array uses the circular patch as its unit antenna
element. The patch radius, R, for the dominan t TM)) mode at the
resonant frequency (Ir in GHz) can be calculated from [17]
2 Array Design
(1)
Unlike most other elements, the electrical parameters
of
the
substrate will be affected by the temperature and moisture varia
tions occurring in airborne applications, and these affect the
performance
of the antennas. The magnitude of the transmitting
power may also generate a large amount of heat, which results in a
significantly increased temperature. The choice of the substrate is
therefore an important factor in airborne-antenna design . The
configuration
of
the antenna element determines the complexity
of
the array feeding-line network, which controls the size and mass of
the array. The configuration of the feeding-line networks may also
incur different levels of port isolation and pattern polarization.
Important design considerations of a dual-polarized airborne array
antenna are summarized in Table
I.
Our goal is to design a low
mass conformal array antenna for airborne applications. An ultra
thin substrate is used to achieve the
conformal
antenna.
D
The multi layer array structure for dual-band (S band and X
band) operation is shown in Figure
I.
The S-band antenna elements
sit on the top layer , and the X-band antennas are on the bottom
layer. A foam layer
2
) serves as the spacer, and is sandwiched
between the two subst rate layers . One of the important design
considerations for this multilayer dual-band array is that the S-band
antenna element should be nearly transparent to the X-band
antenna elements. Otherwise, the S-band element may degrade the
performance of
the X-band antenna.
y.
x
Figure 1. The multilayer structure of th e dual- band dual
polarized array
antenna.
Table 1. Design considerations
for
the airborne array antenna.
Factor
Impact
Temperature
This is determined by the aerial conditions and delivered power, which will affect the electrical
parameters
ofthe
substrate.
Moisture
The effects due to moisture are similar to the temperature effects.
Electrical parameters
This controls the antenna's nerformance and is mainly changed by large temperature variations.
Feeding-line network
This determines the complexityof the array. Unsuitable antenna elements could result in a bulky
and heavy array.
Isolation
Port-to-port isolation and cross-polarization level can be enhanced by using well-designed
feeing-line networks, such as proximity/aperture coupling and multilayer configurations.
Coupling
For dual-frequency operation, the interference between antennas operating at different
frequencies may affect radiation patterns and gain.
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where O and h (in em) are the relative dielectric constant and
thickness of
the substrate, respectively. At the operating frequency,
fr
=10GHz, an initial value of R of 5.82 mm, calculated from
Equation
I ,
was used. The optimized value
of
5.95
mm
was
obtainedwith the aid
of
IE3D.
The circular patches are fed with microstrip lines at the
circumferential edge, as shown in Figure2a. For a single circular
patch, two microstrip feeding lines are used to feed the circular
patch to generate two orthogonally radiating TM
t
1
modes for dual
polarized operation. Two feed points are located at the edge
of
the
patch, 90
0
away from each other, so that the coupling between
these two ports can be minimized. The port isolation also depends
on the quality factor of the patch. Increasing the substrate's thick
ness decreases the isolation [18]. Therefore, using thin substrates
could improve the quality
of
isolation.
with
F =8.79I
F
,
(2)
, 2R , L H-port
<
l
L- -- --- ::::_ :t------------------- -------------------
Figure 2a. The layout of the X-band antenna, where
L =
106,
W =183, R =
5
.82,
LxI =
22,
and W
xt
=
0.17 (all dimensions
are in mm),
..-.._. _._.._.
L,5
----------------------------------------------------
1
,
,
.
H-port :
Figure 2b.
The
layout of the S-band antenna, where
L
=
106,
W=183
, L
st
=53.89 , L
s2
=44.6 , L
s3
=0.94 , L
s4
=19.31 ,
L
ss
=
34
.17,
W
st
=
4
.9 , and W
s2
=
7.8 (all dimensions
are
in
mm).
Figure2a shows a 4 x 8 dual-polarized X-band array. The V
port and the H port are the input ports for the two orthogonal
polarizations (vertical and horizontal). The array is composed
of
two 4
x
4 subarrays. The corporate-fed power-divider lines split the
input power at eachport to the subarrays.Within each subarray, the
circular patches are configured into four 4
x
1 series-fed resonant
type arrays, which make the total array compact and have less
microstrip line losses than would a purely corporate-fed type of
array [19]. An open circuit is placed after the last patch
of
each
4 x 1 array.The spacing between adjacent circular-patch centers is
about one guided wavelength
g
=
21.5mm at 10 GHz). This is
equivalent to a 360
0
phase shift between patches, such that the
main beam points to the broadside. The power coupled to each
patch can also be controlled by adjusting the size
of
the individual
patch to achieve a tapered amplitude distribution for a lower
sidelobe design. Another advantage of using the series-fed array
configuration is that the array can be easily converted into a travel
ing-wave array, with a matched termination at the end
of
the last
elements, if a steered-beamarray is needed.
2 2 S Band Antenna and Subarray
Figure 3. The geometry of the dual-band dual-polarized array
antenna.
As shown in Figure 2b, the S-band antenna elements are
printed on the top substrate, and are separated from the X-band ele
ments by the foam layer. To reduce the blocking of the radiation
from the X-band elements at the bottom layer, the shape of the S
band elements has to be carefully selected. A ring configuration
was a good candidate, since it uses less metallization than an
equivalent patch element. Here, a square-ringmicrostrip antenna is
used as the unit element
of
the S-band array. Because antenna ele
ments at both frequency bands share the same aperture, it is also
preferred that the number of elements on the top layer be as small
as possible, to minimize the blocking effects.
The stacked X-band and S-band array antennas are shown in
Figure 3. As can be seen in the figure, the four sides of the square
ring element are laid out in such a way that they only cover part of
the feeding lines on the bottom layer, but none
of
the radiating ele
ments. Unlike an ordinary microstrip-ring antenna that has a mean
circumference equal to a guided wavelength, the antenna proposed
here has a mean circumference
of
about 2
g
g
=
82.44mm at
V-port .
(S-band) •
H-port
(X-band)
H-port
(S-band)
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• Mean radius of the ring - 2
9
GHz
• Select two operating frequencies I r I I
• Use very thin substrates
10 GHz
• Use Eq. (1)
to
calculate R
• Select I
X1
- 1 '9' w
x1
= 0.17 mm
• Based on (l
s2
- Is1)< (lx1 - 2R)
and (l
s2
+ I
s1)
2 - 2 '9
select
I
s1
and I
s2
• Select I
s3
and W
s3
(50 ohm)
No
• Design the power divider
• Interconnect the patch elements
No
Yes
Yes
Overlay the 3 GHz elements on the 10 GHz elements with
minimum blockage of the circular patches
Figure 4. The design flowchart of the proposed
antenna array.
3 GHz). Although the size of the proposed unit element is larger
than an ordinary ring antenna, its gain is about twice as high,
because
of
its larger radiation-aperture area. The ring is loaded by
two gaps at two of its parallel sides, and these make it possible to
achieve a 50 n input match at the edge
of
the third side without
using a small value
of
Ls
LsI
as mentioned in [20]. For an edge-
fed microstrip ring, if a second feed line is added to the orthogonal
edge, the coupling between the two feeding ports will be high. The
V-port and H-port feeds are therefore placed at two individual ele
ments, so that the coupling between the two ports can be signifi
cantly reduced. Using separate elements seems to increase the
number of antenna elements within a given aperture. However, this
harmful effect could be minimized by reducing the number
of
ele
ments with the use of larger-sized microstrip rings. A design flow
chart of the proposed antenna array is provided in Figure 4.
3. Measurement and Discussion
A dual-band array prototype was fabricated and tested in the
anechoic chamber of the Texas A&M University. The array was
tested for one polarization at one frequency at a time, while the
other three ports were terminated with 50n loads. Detailed simu
lated and measured results are given and discussed in the follow
ing.
3 X-band Array
top of it. The center frequency of both polarizations was at around
9.95 GHz, and the return losses 8
11
and 8
22
) were better than
22 dB. 8
11
was the return loss for the V port, and 8
22
was for the
H port. The isolation
21
or 8
12
)
between the V and the H
polarizations was better than 30 dB at the resonant frequency, and
better than 25 dB over a wide frequency band. These results were
considered excellent for a dual-polarized array for which the feed
ing lines for both polarizations were present at the same layer.
These results were similar to those reported in [4], and the physical
size of the array was reduced, due to the hybrid use
of
the corpo
rate-fed and series-fed configurations.
Normalized measured radiation patterns are shown in Fig
ure 6. Well-defined patterns were observed. Cross-polarization lev
els in the E plane and H plane were 17dB below the co-polarized
beam peaks. It was noted that the dimensions
of
the ground plane
used for the array were about 18.3 em x 10.6 em, which were close
to those of the array's aperture. This could create strong edge
diffraction, and might account for the relatively higher cross
polarization levels. The peak sidelobe levels (SLL) were
-10
to
-13dB, which were normal for the arrays with a uniform ampli
tude distribution. The asymmetric sidelobes of the H port were
caused by its feeding-line network asymmetry with respect to the
array's center. The maximum measured gain was
18.3
dBi, and the
averaged radiation efficiency was 31%. The half-power beamwidth
(HPBW)
of
the x-z plane was 9°, and that
of
the y-z plane was 17°.
This difference was due to the asymmetry of the 4
x
8 array
arrangement.
Figure 5 shows the return loss and the isolation
of
the X-band
array. The measurements were carried out with the S-band layer on
Theoretically, a microstrip antenna has a very good front-to
back ratio (FBR), due to its infinite - or relatively large - ground
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plane . This is similar to the case when an airborne antenna is
mounted on an airframe. Here, with a finite ground plane , the
front-to-back radiation ratios of both the E and H planes were bet
ter than 30 dB. The back radiation was very small , and hence not
shown. A ground plane with a larger dimension can provide a
higher front-to-back radiation ratio, since the coupling and diffrac
tion
of
electromagnetic waves are reduced. Detailed specifications
of the X-band array are summarized in Table 2.
3 2 S Band Array
- - Mea sured E-planeco-pol
--- -- Measured E-plane x-pol
_
_
<
_ Simulated E-plane co-pol
Measured H-plane co-pol
-----
Measured H-plane x-pol
Simulated H-plane co-pol
Figure 6a. The radiation patterns of the X-band array: V-pert
feed, E plane.
he return loss of the S-band array is shown in Figure 7. The
measurements were carried out with the X-band layer under the
S-band layer. About 19 dB return loss was obtained at the resonant
frequency of 2.96 GHz, for both polarizations. Since two separate
elements were used for two polarizations, the results show that the
port isolation was close to 30 dB. Normalized measured radiation
patterns for both polarizations are shown in Figure 8. A typical sin
gle main beam with a wide 3 dB beamwidth was observed . The
cross-polarization levels were better than 20 dB. The maximum
gain of the antenna was measured to be 9.5 dBi, and the averaged
radiation efficiency was 81%. The HPBW of the S-band array was
about 56° on each plane. The front-to-back radiation ratios on both
planes were better than 34 dB for the V port and 25 dB for the H
port, which indicated that the dimensions of the ground plane were
acceptable for the S-band antenna.
(dB) -30 -20 -10
Measured E-plane co-pol
Measured E-plane x-pol
Simulated E-plane co-pol
Figure 6b. The radiation patterns of the X-band array: V-port
feed, H plane.
-20
- Measured S11
........Simulated S11
> < Measured S12
-10 ...
o
·30 .
-20 -10
Measured H-plane co-pol
0° - - - - - Measured H-plane x-pol
Simulated H-plane co-pol
Figure 6c. The radiation patterns of the X-band array: H-port
feed, E plane.
-30 .
-20 .
-40
9.5 9.6 9.7 9.8 9.9 10 10.1 10.2 10.3 10.4 10.5
Frequency (GHz)
Figure Sa. The S parameters of the X-band array, where
port
1
is the V-port feed for vertical polarization.
o
Figure Sb. The S parameters ofthe X-band array, where port 2
is the H-port feed for horizontal polarization.
Figure 6d. The radiation patterns of the X-band array: H-port
feed, H plane.
74
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Table
2. A
summary of the measured and simulated results for the X-band
and
S-band array antennas.
X
-Band S-Band
Polarization
V
Port
H
Port
V
Port
HPort
Frequency (GHz)
Measurement 9.8-9.98 9.81-10.0 2.94-2.96 2.94-2.96
Simulation 9.91-9.99 9.9-10.0 2.94-2.96
2.94-2.96
Bandwidth (%)
Measurement
1.8
2.3
1.03 1.03
Simulation
1.0 1.0 1.03 1.03
Gain (dBi)
Measurement
18.3
17.3 9.5
7.9
Simulation 17.6 17.6 8.72
8.73
Efficiency (%) Measurement 32.4 28.8 96.8 66.8
Simulation 27.5 27.5 80.9
80.9
HPBW (degrees)
E plane
17°
8.9° 58° 59°
Hplane
9.5°
7 4°
53° 52°
Peak SLL (dB)
Measurement
-12
.3 -1 0.0
None
None
Simulation -13 .0 -13 .0
FBR (dB) -30
-30 -34 .8
-25.5
Isolation (dB)
> 3I.l > 25.3 >36.4
>33 .8
X to S band X to S band S to X band S to X band
3.3 Mutual Coupling Effects of Two Layers
Figure
7.
The
S parameters
of the S-band
array, where port 1
is the V-port feed for vertical polarization, and port 2 is the H
port
feed for horizontal polarization.
- - Measured [- plane co-pol
Measured [- planes -pot
Measured If-plane co-pol
Measur ed ..plane x..pol
• SimulatedE-plane co-pol
Simulated H-plane co-pol
- - Measured E-plane co-pol
Measured [ -plane r-pel
Measured H-plane co-pol
- Measu red H-plan e s-pot
• S imulat ed [ - planeco-pol
Simulated If-planeco-pol
-90
Figure
8a. Radiation
patterns
ofthe
S-band array
:
V-port
feed.
Figure 8b. Radiation patterns of the S-band array: H-port feed.
ments for the S-band array, with or without the presence of the X
band array, including the S parameters and the radiation patterns.
When the X band was presented, the isolation of S-to-X (from the
ports
of
the S-band antenna to the ports
of
the X-band) was
between 33 dB and 42 dB; the isolation of X-to-S was better than
25 dB.
It
was observed that increasing the spacing between the S
band and the X-band antennas did not significantly improve the
isolation. Instead, it raised the center frequency
of
the S-band
antenna, and vice versa. Consider the case where the foam-layer
thickness
z
)
is changed within
±O.5 mm
, If h
z
is increased by
3.2.15.1
- - - - Measured511
Measured 522
- Measreud 521
-e-o-e-Simulated S
- Simulated S22
2.95 3 3.05
Frequency (GHz)
2.9
.85
-20 - - - - - -- - - - -- - - - - - -- - - - -- - - - - - - -- -- - - - - - - - - - -
-30
·10
-40
-50
2.8
CD
Simulated results
of
the return losses and radiation patterns
from IE3D are also shown for comparison. Good agreement was
observed for both frequency bands. Small discrepancies in the
resonant frequencies may be attributed to the accuracy of the
permittivity and the thickness dimension ofthe foam layer given by
the manufacturer. The former value played a more important role.
The resonant frequency shifts from 3 GHz to 2.95 GHz when the
dielectric constant of the foam layer
6Z)
changes from 1.06 to
I.l2 , a 5.7% change within the inaccuracy
of
the fabrication proc
ess. The effects due to the thickness ofthe foam layer are described
in the following section. The specifications of the S-band array are
summarized in Table 2.
The results presented here for both the X and S bands were
measured with the concurrent presence of
both antenna layers
(dual-band operation). However, measurements of the single layer
without the presence of the other layer,
i.e.,
single-band operation,
were also conducted, to investigate the mutual-coupling effects
between the elements of different frequency bands.
It
was found
that there were no distinct variations between the two measure-
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Dual layer
- - - X-ba nd la yer only
-90
Figure
9a.
Comparisons
of
the X-band
radiation patterns of
the If-port feed for the E plane.
5 Acknowledgment
The authors would like to thank the Rogers Corporation for
donating the high-frequency laminates, and Mr. Ming-Yi
Li of
Texas A&M University for his technical assistance and helpful
suggestions.
6 References
1. A. Vallecchi and G. Gentili, Design of Dual-Polarized Series
Fed Microstrip Arrays with Low Losses and High Polarization
Purity, IEEE Transactions on Antennas and Propagation, AP-S3,
5, May 2005, pp. 1791-1798.
3. X. Qu, S. Zhong, Y. Zhang, and W. Wang, Design of an SIX
Dual-Band Dual-Polarised Microstrip Antenna Array for SAR
Applications,
lET
Microwave, Antennas, and Propagation, 1, 2,
April 2007, pp. 513-517.
5. G. Chattopadhyay and J. Zmuidzinas, A Dual-Polarized Slot
Antenna for Millimeter Waves, IEEE Transactions on Antennas
and Propagation, AP-46, 5, May 1998, pp. 736-737.
2. S. Gao and A. Sambell, Dual-Polarized Broad-Band Microstrip
Antennas Fed by Proximity Coupling, IEEE Transactions on
Antennas and Propagation, AP-S3, 1, January 2005, pp. 526-530.
4. S. Zhong, X. Yang, S. Gao, and J. Cui, Comer-Fed Microstrip
Antenna Element and Arrays
of
Dual-Polarization Operation,
IEEE Transactions on Antennas and Propagation, AP SO 10,
October 2002, pp. 1473-1480.
-30dB)
Dual layer
- - - X-band layer only
Figure
9b. Comparisons of the X-band radiation patterns of
the
H-port feed for
the
H plane.
0.1 mm, the frequency is raised by 40 MHz; while if
h
2
is
decreased by 0.1 mm, the frequency is reduced by about 60 MHz.
6. K. Mak, H. Wong, and M. Luk, A Shorted Bowtie Patch
Antenna with a Cross Dipole for Dual Polarization, IEEE Anten
nas and WirelessPropagation Letters, 6, 2007, pp. 126-129.
For the X-band array, only small variations in the sidelobe
levels were observed, and the measured peak gain dropped about
0.5 dB with the presence of the top S-band layer. Comparisons of
the radiation patterns
of
the H port are shown in Figure 9. The S
band layer presented little effect on the performance of the X-band
layer. The statement that one antenna layer was transparent to the
other one was therefore confirmed.
7. S. Gao and A. Sambell, Dual-Polarized Broad-Band Microstrip
Antennas Fed by Proximity Coupling, IEEE Transactions on
Antennas and Propagation, AP-S3, 1, January 2005, pp. 526-530.
8. K. Ghorbani and R. Waterhouse, Dual Polarized Wide-Band
Aperture Stacked Patch Antennas, IEEE Transactions on Anten
nas and Propagation, AP-S2, 8, August 2004, pp. 2171-2174.
4 Conclusions
A dual-frequency (S-band and X-band) dual-polarization
array antenna has been developed. An ultra-thin structure was
adopted for the purpose of use with aircraft. The conformal array
can be installed on the airframe or inside the aircraft, due to its
small size and light weight. The X-band array used a series-fed
configuration to save the space
of
the feeding-line network. The S
band array adopted a larger radiation aperture to decrease the num
ber of elements, to reduce the blockage, and to enhance the radia
tion gain. The V and H ports were put on two separate elements to
achieve high isolation. More subarrays could be assembled to
obtain higher gain with a narrower beamwidth. The newly devel
oped dual-frequency dual-polarization array antenna should be use
ful for future wireless communications, remote sensing, surveil
lance, radar systems, and UAV applications.
9. D. Krishna, M. Gopikrishna, C. Aanandan, P. Mohanan, and K.
Vasudevan, Compact Dual-Polarized Square Microstrip Antenna
with Triangular Slots for Wireless Communication, lEE Electron
ics Letters, 42,16, August 2006, pp. 894-895.
10. K. Wong and T. Chiou, Design
of
Dual Polarized Patch
Antennas Fed by Hybrid Feeds, Proceedings of the IEEE Fifth
International Symposium on Antennas, Propagation, and EM The
ory, Beijing, China, August 2000, pp. 22-25.
11. Y. Guo and K. Luk, Dual-Polarized Dielectric Resonator,
IEEE Transactions on Antennas and Propagation, AP-51, 5, May
2003,pp.1120-1123.
12. B. Lindmark, S. Lundgren, J. Sanford, and C. Beckman, Dual
Polarized Array for Signal-Processing Applications in Wireless
Communications,
IEEE Transactions on Antennas and Propaga
tion, AP-46, 6, June 1998, pp. 758-763.
76
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13. A Parfitt and N. Nikolic, A Dual-Polarised Wideband Planar
Array for X-Band Synthetic Aperture Radar, IEEE International
Symposium on Antennas and Propagation Digest, 2, Boston, MA,
July 2001, pp. 464-467.
14. H. Wong, K. Lau, K. Luk, Design
of
Dual-Polarized L-Probe
Patch Antenna Arrays with High Isolation, IEEE Transactions on
Antennas and Propagation, AP-52, 1, January 2004, pp. 45-52.
17. C. Balanis, Antenna Theory, Second Edition, New York, John
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Antennas, Volume 1, London, Peter Peregrinus, 1989.
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1.
Bahl, and A. Itt ipiboon, Microstrip
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1633-1639.
Introducing the Feature Article Authors
Texas -A&M University, he conducted research on wireless power
transmission, power combining, ultra-wideband antennas, and
phased arrays. In 2007, he joined Intelligent Automation Inc.,
Rockville, MD, where he is involved in research and development
of
advanced antennas and RP/microwave systems, including
conformal antennas, metamaterial antennas, VAV collision avoid
ance radar, airborne weather radar, through-the-wall noise radar,
and RPID sensors for biological-warfare agent detection and struc
ture health monitoring.
Kai
Chang
received the BSEE degree from the National Tai
wan University, Taipei, Taiwan; the MS degree from the State
University
of
New York at Stony Brook; and the PhD degree from
the University of Michigan, Ann Arbor; in 1970, 1972, and 1976,
respectively.
From 1972 to 1976, he worked for the Microwave Solid-State
Circuits Group, Cooley Electronics Laboratory
of
the University
of
Michigan as a Research Assistant. From 1976 to 1978, he was
employed by Shared Applicat ions , Inc., Ann Arbor, where he
worked in computer simulation
of microwave circuits and micro
wave tubes . From 1978 to 1981, he worked for the Electron
Dynamics Division, Hughes Aircraft Company, Torrance, CA,
Yu-Jiun
Ren
received the BSEE from Nat ional Chung
Hsing University, Taiwan; the MS degree in Communication Engi
neering from National Chiao-Tung University, Taiwan; and the
PhD degree from Texas A&M Univers ity at Col lege Stat ion; in
2000,2002,
and 2007, respectively. From 2002 to 2003, he was a
research assistant with the Radio Wave Propagation and Scattering
Laboratory, National Chiao-Tung University, and involved in
mobile-radio propagation, channel modeling, and cell planning. At
Shih-Hsu
Hsu received the BSEE degree from National
Cheng-Kung University, Taiwan; the MS degree in Electrical Engi
neering from the University of Wisconsin-Madison; and the PhD
degree from Texas A&M University at College Station; in 2000,
2004, and 2008 , respectively. At Texas A&M University, his
research activities involved microstrip reflectarrays, reconfigurable
antennas, and wideband antennas. In October 2008, he joined
Applied Optoelectronics, Inc., where he is involve in high-speed
laser design.
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where he was involved in the research and development of
millimeter-wave solid-state devices and circuits, power combiners,
oscillators and transmitters. From 1981 to 1985, he worked for the
TRW Electronics and Defense, Redondo Beach, CA, as a Section
Head, developing state-of-the-art millimeter-wave integrated cir
cuits and subsystems including mixers, VCOs, transmitters,
amplifiers, modulators, up-converters, switches, multipliers,
receivers, and transceivers. He joined the Electrical Engineering
Department
of
Texas A&M University in August 1985 as an
Associate Professor, and was promoted to a Professor in 1988. In
January 1990, he was appointed Raytheon E-Systems Endowed
Professor
of
Electrical Engineering. His current interests are in
microwave and millimeter-wave devices and circuits, microwave
integrated circuits, integrated antennas, wideband and active anten
nas, phased arrays, microwavepower transmission, andmicrowave
optical interactions.
Dr. Chang has authored and coauthored several books. He
served as the editor of the four-volume Handbook ofMicrowave
and Optical Components, published by John Wiley in 1989 and
1990 (second edition 2003), and the editor for the Wiley
Encyclopedia
of
RF and Microwave Engineering (six volumes,
2005). He is the editor of Microwave and Optical Technology Let
ters, and the Wiley book series in Microwave and Optical
Engineering (over 70 books published). He has published over 450
papers, and many book chapters in the areas of microwave and
millimeter-wave devices, circuits, and antennas. He has graduated
over 25 PhD students and over 35MS students.
Dr. Chang has served as technical committee member and
session chair for IEEE MTT-S, AP-S, and many international
conferences. He was the Vice General Chair for the 2002 IEEE
International Symposium on Antennas and Propagation. He
received the Special Achievement Award from TRW in 1984, the
Halliburton Professor Award in 1988, the Distinguished Teaching
Award in 1989, the Distinguished Research Award in 1992, and
the TEES Fellow Award in 1996 from the Texas A&M University.
Dr. Chang is a Fellow of the IEEE.
78
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